Printed Quasi-Yagi Antennas Using Double Dipoles and Stub-Loaded Technique for Multi-Band and Broadband Applications

© 2013 IEEE. Double dipoles on a single-layer substrate are utilized to construct a triple-mode printed quasi-Yagi antenna for the multi-band and broadband antenna applications. A stub-loaded dipole generating two resonant modes (i.e., lower dual-mode dipole) is allocated on the underside of a simple dipole (i.e., upper single-mode dipole) introducing the third resonant mode. Using these three resonant modes, three compact printed quasi-Yagi antennas, i.e., tri-band, dual-band, and broadband printed quasi-Yagi antennas, are designed with the same antenna prototype but different parameter values. Seen from the measured results, all of these three antennas have good unidirectional radiations, high radiation efficiencies, and low cross-polarization levels at the operating frequencies within the impedance bandwidths

Analysis of Miniaturized, Circularly Polarized Antennas for Bidirectional Propagation

Size reduction is necessary to fit the recent demand for small sized communication systems in consumer electronics. Wireless communication systems rely on antennas for long range transmission of signals, so size reduced antennas have been sought after in recent years. Also, not many antennas are designed for use in bidirectional scenarios like subways, tunnels, bridges, etc. Three sized reduced antennas with circular polarization are presented for use in bidirectional communication systems. An electrically small pattern reconfigurable array, an electrically small two-sided printed cross dipole, and a size reduced printed wideband antenna are introduced within this thesis. All antennas’ results are obtained from simulation, with two of the antenna designs being measured to verify their results

Analysis of Dual-Element Antenna Configurations

Dual-dipole antennas have been extensively researched previously for their bandwidth enhancing effect on Yagi type antennas. In this thesis, dual-dipole antennas are fabricated and measured. These experimental measurement results are verified with simulated values. First, a standard dual-dipole antenna is investigated and found that the high magnitude, opposite current directions on the dipole arms are the reasoning behind the creation of a high gain mode. This pattern is similar to a Yagi antenna. Next, a dual-band implementation of the dual-dipole antenna is shown, with two distinct resonances in a lower band and an upper band. Both bandwidths exhibit a dipole like mode, as well as a high gain mode. Finally, a dual-element cross-dipole antenna application is investigated. The antenna exhibits multiple dipole like modes within the bandwidth, high gain points, and CP generation at the center frequency

A comprehensive survey on 'circular polarized antennas' for existing and emerging wireless communication technologies

Circular polarized (CP) antennas are well suited for long-distance transmission attainment. In order to be adaptable for beyond 5G communication, a detailed and systematic investigation of their important conventional features is required for expected enhancements. The existing designs employing millimeter wave, microwave, and ultra-wideband (UWB) frequencies form the elementary platform for future studies. The 3.4-3.8 GHz frequency band has been identified as a worthy candidate for 5G communications because of spectrum availability. This band comes under UWB frequencies (3.1-10.6 GHz). In this survey, a review of CP antennas in the selected areas to improve the understanding of early-stage researchers specially experienced antenna designers has presented for the first time as best of our knowledge. Design implementations involving size, axial ratio, efficiency, and gain improvements are covered in detail. Besides that, various design approaches to realize CP antennas including (a) printed CP antennas based on parasitic or slotted elements, (b) dielectric resonator CP antennas, (c) reconfigurable CP antennas, (d) substrate integrated waveguide CP antennas, (e) fractal CP antennas, (f) hybrid techniques CP antennas, and (g) 3D printing CP antennas with single and multiple feeding structures have investigated and analyzed. The aim of this work is to provide necessary guidance for the selection of CP antenna geometries in terms of the required dimensions, available bandwidth, gain, and useful materials for the integration and realization in future communication systems

프린티드 다이폴 배열 내 스캔블라인드니스와 유한한 크기를 갖는 기판 효과에 대한 연구

학위논문 (박사) -- 서울대학교 대학원 : 공과대학 전기·정보공학부, 2021. 2. 남상욱.In this thesis, a study on the mechanism of a printed dipole array mainly used in the millimeter wave band is conducted, which has an important role on the performance of it. Two types of printed dipoles are mainly dealt with, and the first of them describes the scan blindness effect and causes that may occur in T-printed dipoles. Second, the operation principle of bow-tie element printed on a substrate with high dielectric is explained through Physical Optics and diffraction theory. First, the scan blindness in a T-printed dipole in 1D and 2D array is analyzed, and an elimination strategy is proposed. The scan characteristics are obtained using an active element pattern (AEP) with an infinite rectangular lattice arrangement. Based on the propagation of a guided wave along the antenna row and the electric-field distribution observed during simulations, an equivalent circuit model for a unit cell of the T-printed dipole is obtained. A quasi-transverse electromagnetic (TEM) guided wave is predicted using the dispersion relation curve obtained from the equivalent circuit, and it is proved that the calculated curve is in good agreement with the eigen mode simulations and measured trajectory of the scan blind angle, for different frequencies. In addition, the Q value and scan blindness of the 1D array and the 2D array is considered, and it is verified that the mutual coupling of a 1D array decreases more than that of a 2D array due to the radiation loss. Next, slits and stubs are introduced as parasitic structures, to eliminate the scan blindness and improve the antenna scan range. To confirm the effects of these parasitic elements, a linear array simulation is performed, which confirms the suppression of a quasi-TEM guided wave. Finally, the printed dipole array was fabricated, and an AEP was measured for the 11×1(1D array) and 11 × 3 sub arrays(2D array). Their measurements validate the scan blindness prediction and confirm the proposed mechanism of scan blindness and its improvements. In the second type, the principle of a printed bow-tie antenna on a high dielectric substrate is presented. For the sake of simplicity, it is assumed that the infinitesimal horizontal-electric dipole (HED) with unit current is located on the dielectric slab. And the theory of Physical Optics (PO) is applied. Through this study, it is theoretically concluded that the components of the antenna far-field are composed of geometric optics in which the direct ray radiated directly from the HED and the reflected wave by the dielectric are combined, and the diffracted ray generated by the space wave, surface wave, and leaky wave. In order to verify the validity of the theory, the electromagnetic wave analysis programs CST MWS and FEKO are used to compare and verify the theoretically obtained closed form. According to the results of the study, in the case of a high dielectric substrate with dielectric constant of 10 or more, the main component that constitutes the radiation pattern is TE0 surface-wave(SW) diffracted ray generated from the edge of the dielectric slab. In addition, the directivity of the antenna can obtain a high gain of 10 dBi or more, which is advantageous for single antenna design. Finally, based on the previously derived theory, a ku-band printed Bow-tie array antenna with low mutual coupling is proposed. The diffracted ray induced at the edge of the truncated dielectric slab not only generates high gain, but also can reduce mutual coupling due to the cancellation effect of direct ray and reflected ray. In addition, an ohmic sheet is added on the dielectric slab of the single element to attenuate the surface waves traveling to the side and it leads to minimize the mutual coupling. In particular, there is an advantage of obtaining high gain and low mutual coupling without adding additional structures. The single element is designed and fabricated as a 1D sub-array in the H-plane direction. It is confirmed that the gain is higher than 6.5 dBi, and the mutual coupling is less than -20 dB in the band after 12.8 GHz. In conclusion, this thesis propose that the electromagnetic mechanism of the two printed type antenna. For the T-printed dipole, it is revealed that the cause of the scan blindness is TEM guided mode. And It is theoretically revealed that the main cause of the printed bow tie antenna with high permittivity is the diffracted ray of TE0 SW generated from the antenna.본 논문에는 밀리미터파 대역에서 주로 사용되는 프린티드 다이폴 안테나에서 성능에 중요한 영향을 미치는 전자기적 현상에 대해 집중적으로 연구를 진행하였다. 크게 두 가지 형태의 프린티드 다이폴에 대해 다루었는데, 그 중 첫번째는 T-형태를 가지는 프린티드 다이폴에서 발생할 수 있는 스캔블라인드니스 현상과 원인에 대해 기술하였다. 두 번째는 고 유전율을 가지는 기판에 프린티드 된 Bow-tie 안테나를 Physical Optics(PO)와 회절파 이론을 바탕으로 동작 원리를 설명하였다. 첫 번째 타입으로, Ka-대역 프린티드 다이폴이 일차원과 이차원으로 배열되어 있을 때, 스캔 블라인드니스(scan blindness)를 분석하고 제거할 수 있는 방안을 제안하였다. 먼저 일차원과 이차원 사각배열에서 능동소자패턴의 E-면 방향으로 스캔 블라인드니스가 발생함을 확인하였다. 그리고 분산 관계와 스캔 블라인드니스와의 관계를 통해 quasi-TEM 공진 모드의 존재를 확인하였다. 공진모드에서 전기장의 분포를 분석하여 T-형태를 가지는 프린티드 다이폴의 단위 셀에 대한 등가회로를 구현하였고, 이로부터 분산 관계식을 도출하였다. 또한 일차원 배열과 이차원 배열일 때 Q 값 및 프린티드 다이폴의 스캔블라인드니스를 고찰하였고, 일차원 배열인 경우 방사 손실로 인해 상호 결합의 양이 이차원 배열보다 감소함을 확인하였다. 그리고나서, 스캔블라인드니스를 없애고 안테나의 빔 조향 범위를 향상시키기 위해, 슬릿과 스터프를 추가하였다. 이러한 기생 소자들의 효과를 확인하기 위해, 선형 배열 시뮬레이션을 수행하였고 quasi-TEM 모드가 억제되는 것을 확인 하였다. 마지막으로 일차원 배열에 대해서 11×1 부배열을 제작, 자기 벽 효과를 주기 위해 이차원 배열에 대해서는 11×3 부배열을 제작하였다. 측정을 통해 스캔 블라인드 예측을 검증하고 제안 된 스캔 블라인드 메커니즘과 그 개선 사항을 확인하였다. 두 번째 타입으로, 고 유전율을 가지는 기판에 프린티드 된 Bow-tie 안테나의 동작 원리를 고찰하였다. 구조를 단순화 하기 위해 잘려진 유전체 기판위에 미소 다이폴 전류를 가정하였고, Physical Optics 이론을 적용하였다. 본 연구를 통해 안테나 방사패턴의 성분은 미소 다이폴에서 직접 방사하는 직접파와 유전체에 의한 반사파가 합쳐진 Geometric Optics(GO), 그리고 공간파(Space wave), 표면파(Surface wave), 누설파(Leaky wave)에 의해 발생되는 회절파(Diffracted Ray)로 구성되어 있음을 이론적으로 밝혔다. 이론의 타당성을 검증하기 위해 전자파 해석 프로그램인 CST MWS 및 FEKO를 사용하여 이론적으로 구한 closed form과 비교 검증하였다. 연구 결과에 따르면 유전율 10 이상을 가지는 고 유전체 기판의 경우 방사패턴을 구성하는 성분 중 대부분을 차지하는 주요성분은 기판의 모서리에서 발생된 TE0 모드 표면파에 의해 생성된 회절파 때문이다. 또한 이론적으로 안테나의 지향성이 6.5 dBi 이상 고이득을 얻을 수 있어 단일 안테나 설계에 유리하다. 앞서 도출한 이론을 바탕으로 낮은 상호결합을 갖는 ku-대역 프린티드 Bow-tie 배열 안테나를 제안 하였다. 잘려진 유전체의 모서리에서 유기된 회절파는 높은 이득을 생성할 뿐만 아니라, 직접파, 반사파의 상쇄효과로 인해 상호결합을 감소시킬 수 있다. 또한 단일 소자의 측면 하단에 Ohmic sheet를 추가하여 측면으로 진행하는 표면파를 감쇠 시켜 상호결합현상을 최소화 하였다. 특히 별도의 장치를 추가하지 않아도 높은 이득과 낮은 상호 결합을 얻을 수 있는 장점이 있다. 단일 안테나를 H-면 방향으로 일차원 부배열로 설계 및 제작하여 하여, 방사패턴의 이득은 6.5 dBi 이상 고 이득을 얻었고, 상호결합현상은 12.8 GHz 이후의 대역에서 -20 dB 미만을 얻어 설계의 타당성을 검증하였다. 결론적으로, 본 논문에서는 두 가지 형태의 프린티드 다이폴의 주요 전자기적 현상을 검증하였다. T-프린티드 다이폴에 대해 스캔블라인드니스의 원인이 TEM guided mode임을 밝혔고, 고 유전율을 가지는 프린티드 bow tie 안테나의 주요 원인은 antenna에서 generated된 TE0 SW의 diffracted ray 임을 이론적으로 밝혔다.Table of Contents Abstract i Table of Contents v List of Figures ix List of Tables v Chapter 1. Introduction 1 1.1. Trend of T-Printed Dipoles 1 1.2. Scan blindness and its elimination methods. 3 1.3. High Dielectric Antennas and Physical Optics (PO) 7 1.4. References 9 Chapter 2. Analysis of Scan Blindness in a T-Printed Dipole Array 12 2.1. Motivation 12 2.2. Study on Single Element Design 13 2.3. Active Element Pattern in Infinite Array 16 2.4. E-field Distribution in Infinite Array 18 2.5. Dispersion Diagram 20 2.6. Guided quasi-TEM mode. 22 2.7. Equivalent Circuit 24 2.8. Parasitic Elements for Eliminating Scan Blindness 25 2.8.1. Comparison of Active Reflection Coefficients. 27 2.8.2. Guiding Wave Suppression 27 2.8.3. Comparison of Dispersion Relations. 30 2.9. 1D Array Analysis 31 2.10. Finite Array and Measurements 37 2.10.1. Physical Explanation for Finite Array. 37 2.10.2. Scanning Performance 38 2.10.3. Simulations and Measurements 41 2.11. Conclusion 46 2.12. References 48 Chapter 3. Analysis of an HED on a Truncated Dielectric Slab 50 3.1. Motivation 50 3.2. Basic Formulation for PO 52 3.3. Polarization Current in Multilayered Structures 52 3.3.1. TEz Mode Solution in Spectral Domain. 56 3.3.2. TMz Mode Solution in Spectral Domain 60 3.3.3. Complete Polarization Current Solution 65 3.4. Dispersion Equation Solutions 67 3.4.1. Surface Wave Solutions 67 3.4.2. Complex Guided Wave Solutions 71 3.5. The Saddle-Point Method of Integration 74 3.6. Geometrical Optics 81 3.7. Diffracted Rays 84 3.7.1. Surface wave diffracted wave 84 3.7.2. Leaky wave diffracted wave 85 3.7.1. Space wave diffracted wave 86 3.8. Numerical Results 87 3.8.1. Analysis of both sides truncated substrate 85 3.8.2. Analysis of a substrate standing perpendicular to the ground plane 94 3.8.3. Limitation of the Physical Optics 94 3.9. Implementation of Bow-tie printed Dipole Array 100 3.9.1. Motivation 100 3.9.2. Single Antenna Design 101 3.9.3. Return Loss Characteristics 101 3.9.4. Mutual Coupling 101 3.9.5. Measurements 111 3.9.5.1 Single Element 111 3.9.5.2 H-plane 1-D Array 113 3.10. Conclusions 116 3.11. References 118 Chapter 4. Conclusions 119 Appendix 121 A.1. Derivation of Dispersion Relations. 121 A.2. Detailed Derivation of (3.43) 128   List of Figures Fig. 1.1. Evolution of a Printed Dipole Antenna. (a) Conventional dipole with a coaxial balun. (b) Printed dipole with integrated balun. (c) Recent printed dipole. 2 Fig. 1.2 Scan Blindness and its elimination methods 5 Fig. 1.3 Printed Dipole Type Antenna with High Dielectric.(a) WideBand Modified Printed Bow Tie Antenna. (b) Surface Wave Enhanced Yagi-Uda Antenna. (c) Horizontal Electric Dipole on ㅇdd High Dielectric Substrate.. 7 Fig. 1.4 Measured ambient RF power at ourdoors [23-25]. (a) London. (b) Colorado, USA. (c) Seoul, South Korea. 8 Fig. 2.1 Geometry of an element: (a) Basic T-printed dipole. (b) Tprinted dipole with slits and stubs.. 12 Fig. 2.2 Active element pattern in an infinite array. (Comparison of E-plane and H-plane): (a) Basic T-printed dipole. (b) Tprinted dipole with slits and stubs . 14 Fig. 2.3 E-field distribution between two dipoles in infinite array when Floquet excitation is performed so that the incident angle is (a) blind angle (36°) and (b) boresight angle (0°) ... 16 Fig. 2.4. Simulated dispersion diagrams of eigen mode, trajectory of scan blindness, and calculated dispersion relations obtained from Appendix (A.12). 19 Fig. 2.5. Electric field distribution of coupling between linear infinite dipole arrays placed 10 unit cells apart at 35 GHz for basic Tprinted dipole array 20 Fig. 2.6. (a) Equivalent topology of a T-printed dipole unit cell. (b) Transformed equivalent circuit 20 Fig. 2.7. Comparison of four types of two-dimensional active reflection coefficients: (a) basic T-printed dipole, (b) with stubs, (c) with slits, and (d) with slits and stubs 23 Fig. 2.8. Electric field distribution of coupling between linear infinite dipole arrays placed 10 unit cells apart at 35 GHz for basic Tprinted dipole array) 23 Fig. 2.9. Electric field distribution of coupling between linear infinite dipole arrays placed 10 unit cells apart at 35 GHz for basic Tprinted dipole array.. 25 Fig. 2.10. Simulation and measurement results of (a) Input reflection coefficient. (b) Maximum realized gain and total radiation efficiency. 26 Fig. 2.11. Simulated dispersion diagram of eigen mode for different length of the slit (slith) and stub (stubh) 26 Fig. 2.12. 1D Active element pattern 28 Fig. 2.13. Dispersion relations comparison between eigen mode simulation and scan blindness. 31 Fig. 2.14. Q factor comparison between 1 D and 2 D. . 31 Fig. 2.15. Electric field simulation of coupling between linear dipole arrays placed 9 unit cells apart for open boundary. (a) Open boundary (b) PMC boundary 32 Fig. 2.16. S-parameter between linear dipole arrays placed 9 unit cells apart 32 Fig. 2.17. Geometry of 11 × 3 basic T-printed dipole array for a finite active element pattern 33 Fig. 2.18. Structure of an active element pattern of the 11 × 3 array (a) Basic printed dipole. (b) Proposed 33 Fig. 2.19. Structure of fully excited 8 × 1 arrays in E-plane with 41° scan angle. (a) Basic printed dipole. (b) Proposed. 33 Fig. 2.20. Simulated scanning performance in the E-plane for the 11 × 3 arrays with an excited 8-element linear array. (a) Basic printed dipole (b) Proposed 36 Fig. 2.21. Four types of center row substrate arrays fabricated: (a) Active element pattern (AEP) of the 11 × 3 arrays for the basic T-printed dipole. (b) AEP of the 11 × 3 arrays for the proposed T-printed dipole. (c) Fully excited 8 × 1 arrays for the basic T-printed dipole. (d) Fully excited 8 × 1 arrays for the proposed T-printed dipole. (e) Array of printed dipoles with slits and stubs mounted on the antenna bracket 61 Fig. 2.22. . E-plane co-polarization active element pattern of the 11 × 3 arrays. (a) Basic printed dipole. (b) Proposed 56 Fig. 2.23. Fully excited 8 × 1 arrays in E-plane co-polarization at 41° scan angle. (a) Basic printed dipole. (b) Proposed. 59 Fig. 2.24. Printed Dipole AEP in the E-plane for 1D array. (a) 11×1 sub array for printed dipole. (b) 11×1 sub array for printed dipole with slit and stub 43 Fig. 2.25. Comparison of scan blindness occurrences when the printed dipole array antenna is steering from broadside to 50° in the Eplane. Eigen mode (simulated) vs. dispersion relations (calculated) vs. scan blindness (measured) 44 Fig. 3.1 Geometry of an HED over a truncated dielectric slab on the ground plane 52 Fig. 3.2. Equivalence theorem. (a)current on dielectric slab. (b)Equivalent current source. 55 Fig. 3.3 Spectral equivalent circuit for the TEz mode. 57 Fig. 3.4. Spectral equivalent circuit for the TMz mode 62 Fig. 3.5. Surface-wave pole solutions. (a)TE mode. (b)TM mode. 70 Fig. 3.6. Graphical solution for complex pole solutions. Solid lines are for real part solution; broken lines are imaginary part solution. (a)TE mode. (b)TM mode. 74 Fig. 3.7. Topology of the proper Rieman sheet of the complex ky plane 76 Fig. 3.8. Topology of the top Rieman sheet of the complex s plane. 77 Fig. 3.9. Radiation Pattern for infinite dielectric slab. (a) 3D simulation results (b) comparison 83 Fig. 3.10. Truncated dielectric slab modeling 88 Fig. 3.11. Radiation pattern at H-plane for truncated dielectric slab(ϕ component, = 4). 88 Fig. 3.12. Radiation pattern at Diagonal plane for truncated dielectric slab. (a) ϕ component (b) θ component 90 Fig. 3.13. Surface-wave diffraction contribution. 91 Fig. 3.14. Space-wave diffraction contribution. 91 Fig. 3.15. Geometric Optics contribution. 92 Fig. 3.16. Fig. 3.16. Diffracted ray and GO contributions for = 12.2. 93 Fig. 3.17 (a)Truncated dielectric slab over the ground plane. (b) Image theory model 1 (c) Image theory model 2.. 95 Fig. 3.18. Far-field pattern for = 12.2, = 0.2 0... 96 Fig. 3.19. Fabricated 15x1 H-plane Array... 99 Fig. 3.19 Design model (a)Front view (b) Side view 97 Fig. 3.20. Directivity for = 12, = 0.2 0. : (a) Radiation pattern and E-field distribution, and (b) comparison PO calculations with CST simulations. 99 Fig. 3.21. (a) An HED on the finite dielectric substrate over the ground, and its (b) directivity pattern depend on cutting angle.. 102 Fig. 3.22. (3D directivity pattern for Fig.3.21: (a) θ_cut=0° , and (b) θ_cut=60°. 102 Fig. 3.23. An HED on the substrate with trapezoidal type over the ground: (a)Perspective view, (b)Front view,(c) Side view, and (d) Top view. 105 Fig. 3.24. E and H-plane single element pattern. 106 Fig. 3.25. Single element model: (a)model 1, (b)model 2, and (c) model 3.. 107 Fig. 3.26. Return loss characteristics for Fig. 3.25. 108 Fig. 3.27. Two element array: (a) geometrical model, and (b) mutual coupling on dielectric constant changes. 110 Fig. 3.28. Fabricated single element. 111 Fig. 3.29. Far-field pattern in H-plane: (a) Measurements VS CST Simulations, and (b) Measurements VS PO Calculations. 112 Fig. 3.30. Fabricated 15x1 H-plane Array.. 113 Fig. 3.31. S-parameter results. (a) Retrun loss (b) Mutual coupling between the center element and the adjacent element.. 114 Fig. 3.32. E and H-plane co-polarization active element pattern of 15x1... 115 Fig. A.1.1 Extraction of TL parameters. (a) Simulation setup for TL parameters. (b) Characteristic Impedance ZTL and phase constant βTL..... 125 Fig. A.1.2. Extraction of gap capacitance parameters. (a) Simulation setup for gap capacitance parameters. (b) Gap capacitance for series (Cgs) and gap capacitance for parallel (Cgp). 126 Fig. A.1.3. Extraction of transformer parameters. (a) Simulation setup for transformer parameters. (b) Self-inductance (L) and mutual inductance (Lm). 27 List of Tables TABLE 2.1. Design parameters of T-printed dipole element 13 TABLE 3.1. Special equivalent circuit for electric current 56 TABLE 3.2. Dispersion equation 65 TABLE 3.3. Dispersion equation for complex wave 71 TABLE 3.4. Diffraction Coefficient 81Docto

Wireless Applications of Radio Frequency Micro-Electro-Mechanical Systems

With mass proliferation of wireless communication technologies, there is a continuous demand on fast data transmission rate and efficient use of frequency spectrum. As a result, reconfigurable systems are of significant importance and research is being conducted in numerous universities. The purpose of this research is to develop novel RF MEMS based reconfigurable wireless systems. By utilizing the RF MEMS switches as a basic building block, this thesis focus on developing a unique design technique for the design and development of RF MEMS delay line phase shifter, frequency reconfigurable antennas and pattern reconfigurable antennas. This thesis work is divided into four parts: 1. Investigation and development of nano-electro-mechanical systems (NEMS) based 3-bit phase shifter. Analyzing the slow wave structure to further reduce the size of delay line phase shifter. 2. Development of frequency reconfigurable antennas to compete with broadband and multi-band antennas. Two novel MEMS-loaded frequency reconfigurable antennas were designed with spectrum switchable between WPAN band (57 to 66 GHz) and the whole E-band (71 to 86 GHz). 3. Investigation of microstrip-to-coplanar striplines (CPS) transition balun used for antennas to explain the inherent phase delay of this type of structure. Based on the discovery, a pattern reconfigurable quasi-Yagi antenna was designed. The antenna exhibits excellent RF performance, compact size and switchable end-fire radiation pattern with the goal to replacing existing phased array antennas. It has the full functionality of a multi-antenna phased array plus phase shifting network while its size is same as a fixed single Yagi antenna. 4. Development of full seven masks all metal fabrication process of the RF MEMS integrated reconfigurable antennas. The fabrication processes are optimized based on Australian National Fabrication Facility (ANFF) New South Wales node’s equipment