The experimental results from simple ultra

wideband link are presented. The UWB link consisting of typical

broadband microwave circuits built of commercially available

components is able to send and detect unmodulated broadband

electrical pulses with 20 MHz pulse repetition frequency. The

system operates with approximately 60% of fractional

bandwidth in 4GHz band with spectral density of -140dBW/Hz

Le Kernec, Julien

Rulikowski, Pawel

Sokol, Vratislav

Waliwander, Tomasz

English

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Waliwander, T., Rulikowski, P., Le Kernec, J., and Sokol, V. (2005) The 
Experimental UWB Link. In: 15th International Traveling Summer School on 
Microwaves and Lighywaves, L'Aquila, Italy, 9-15 July 2005. 
 
 
There may be differences between this version and the published version. You are 
advised to consult the publisher’s version if you wish to cite from it. 
 
 
http://eprints.gla.ac.uk/121067/ 
     
 
 
 
 
 
 
Deposited on: 13 July 2016 
 
 
 
 
 
 
 
 
 
 
 
 
 
 
 
 
 
 
 
 
 
 
Enlighten – Research publications by members of the University of Glasgow 
http://eprints.gla.ac.uk 
 Abstract—The experimental results from simple ultra 
wideband link are presented. The UWB link consisting of typical 
broadband microwave circuits built of commercially available 
components is able to send and detect unmodulated broadband 
electrical pulses  with 20 MHz pulse repetition frequency. The 
system operates with approximately 60% of fractional 
bandwidth in 4GHz band with spectral density of -140dBW/Hz.  
 
Index Terms—Pulse generation, pulse shaping circuits, 
transient response, ultra wideband (UWB) 
 
I. INTRODUCTION 
ECENTLY, ultra wideband technology (UWB) has 
attracted many  worldwide interest in many areas of 
research and commercial applications. Undisputable benefits 
of this emerging technology create new challenges in many 
aspects. Many traditional approaches to system and hardware 
design often fail when applied to UWB, therefore there is a 
need for modified or even new techniques.  
 The most widespread definition for UWB system seems to 
be that proposed by US Federal Communication Commission. 
This defines the UWB communication as any kind of 
communication that occupies more than 25% of fractional 
bandwidth or at least 500MHz as a nice gesture to industrial 
community. Even though this definition causes some 
controversy and misunderstanding it is the only one at the 
moment consistent. The Europe is still behind in terms of 
legislation but proposed by ETSI regulation quite well reflects 
American regulations. The FCC (and proposed ETSI) 
approved UWB mask is presented in Fig.1. The allowed 
frequency band spreads from 3.1 to 10.6 GHz giving 
enormous, never before accessible 7.5GHz of constant 
bandwidth. An allowed power spectral density is -41.3 
dBm/MHz of Effective Isotropically Radiated Power (EIRP).  
The UWB technology has some unique features that 
promise its utilization in many applications that hard or even it 
is not possible to realize in “traditional” narrowband 
technologies. Some of the unique features and applications 
include:  
• UWB system is highly immune to multipath 
propagation phenomenon 
• Able to provide 3D tracking capability with 
resolutions down to centimetres  
• Power aware and efficient from power consumption 
point of view 
• Extremely hard to eavesdrop 
• Achievable high data rates (up to 500Mb/s) in short 
range communication with low power consumption 
(i.e. Bluetooth quality of service with 100 times less 
average power) 
• Able to see through obstacles (often used as ground 
penetrating radars) 
 
Fig.1. FCC approved UWB indoor mask.  
The UWB systems are potentially interesting mainly for short 
range high data rate communication, precise location, power 
aware systems as sensor networks.  
II. THE ULTRA WIDEBAND TRANSCEIVER  
A. General structure 
An UWB system structure is illustrated in Fig.2. 20 MHz 
square wave crystal stabilized generator provides trigerring 
signal for a baseband pulse generator. The system exploits 
heterodyning principle for placing baseband pulse around the 
desired center frequency in our case 4GHz. Finally, the 
shaped pulse is supplied to the broadband antenna and 
transmitted.  
Our receiver uses one of the simplest possible 
implementation of the UWB receiver – threshold detector. 
Such approach is not optimal but relatively simple in 
implementation. The received signal from antenna is amplified 
in a two stage amplifier – the first stage is LNA based on 
discrete pHEMT transistor and the second one uses a MMIC 
The Experimental UWB Link  
Pawel Rulikowski, Student Member, IEEE, Vratislav Sokol, Member, IEEE, 
Tomasz Waliwander, Julien Le Kernec  
Centre of Adaptive Wireless Systems, Cork Institute of Technology, Department of Electronics,  
Rossa Avenue, Cork, Ireland, Email: prulikowski@cit.ie, sokolv@fel.cvut.cz, 
tomasz.waliwander@cit.ie, julienlekernec@gmail.com  
R 
 broadband amplifier. In the next stage amplified signal is 
detected by a broadband threshold pulse detector.  
Vivaldi 
AntennaCLOCK20MHz
Pulse 
Generator
Fo
TransmitterA)
LNA 
ATF36163
ERA21SM
Threshold 
detector
Vivaldi 
Antenna
Receiver
B)
 
Fig.2. The experimental UWB link structure. 
B. Transmitter 
1) Pulse generator  
The pulse generator used in our experiment uses as a step 
generator step recovery diode MA44769, that excited from the 
digital driver creates two balanced steps that are further 
shaped in distributed pulse shaping network to create short 
electrical pulse [4]. The detailed schematic is shown in Fig.3.  
 
Fig. 3. The circuit diagram of balanced subnanosecond pulse generator.  
A balanced “input generator – driver” was created using  
a CMOS buffer with a Schmitt type input (74ACT14) U1 in 
order to get balanced output from an unbalanced generator. 
The second function of the generator is to shorten rise time of 
square wave connected to input U2 from 50 ns down to 1 ns 
and provide source or sink current of ± 50 mA. When one 
group of gates U1-A, U1-B goes high, the other one U1-F, 
U1-E and U1-D drops it logic state down. Such connection 
results in changing polarization of the diode from forward to 
reverse state. The balanced driver combines two features in 
one – it both biases and triggers SRD generator. The pulse 
forming concept [1] is based on the division of a propagated 
wave over the transmission line between short circuited stub – 
T6 (T3), with impedance being half that of the main line, and 
the “main line” – T1, T2 (T4, T5) and use of occurring 
reflections to shape the output pulse. The generated pulse 
(parameters in Table I and time waveform in Fig.4) has a 
duration equal to twice that needed for propagation over the 
shorted stub and occurs at the output after time needed for 
propagation over T1 and T2 (T4, T5). The step generated by 
the SRD is fed into the pulse forming network through 
capacitor C1 (C2), which prevents current flowing to load 
during the digital driver steady state. The aim of the 
terminating resistors R1 and R2 is to match the transmission 
line junction point to 50 Ω and prevent further reflections of 
pulse thereby reducing ringing. The pulse generator consists 
of two identical sections of the pulse forming network used to 
achieve a balanced output as it is shown in Fig. 3. They were 
connected to the anode and cathode of the SRD. In order to 
obtain unbalanced to balanced transition the ground planes of 
the digital and microwave part of the pulse generator were 
separated. It is worth noting that the whole circuit uses only a 
single positive 5V bias voltage. This feature is extremely 
useful in portable and handheld applications, where negative 
biasing is often simply not accessible or requires use of DC-
DC converters.  
TABLE I  
PULSES’ PARAMETERS 
Parameter Distributed pulse forming network  
Rise/fall time (10-90%) 168 ps 
Pulse width (50-50%) 335ps 
Pulse amplitude ±750 mV (10 dBm –peak power) 
 
 
Fig.4. Time waveform of the positive pulse created with pulse shaping 
network.  
2) Mixer 
A mixer used for up-conversion of the baseband pulse was  
 double balanced type with broadband frequency characteristic 
from 2-8GHz. It is commercial LTCC mixer Mini-Circuits 
type MCA1-85. The layout created on Taconic Cer-10 
(εr=9.5) 0.63mm thick organic substrate with 35µm double 
side copper cladding is presented in Fig. 5.  
 
Fig.5. MCA1-85 mixer layout.  
Block capacitors on the LO (left) and RF (right) were used to 
remove DC components from those signals. The IF port was 
fed by positive pulse from the pulse generator. The LO signal 
was supplied from HPE4433B signal generator. The RF port 
was used as the output port. 
3) Vivaldi antenna  
One critical part of the UWB communication link is the 
effect of the antenna on the transmitted pulse [2]. Thus the 
antenna should have minimal distortion on the transmitted 
pulse. The designed UWB antenna consists of three main parts 
that are microstrip impedance transformer, microstrip to 
slotline transition and Vivaldi antenna, see Fig.6. The 
designed UWB antenna has small dimensions 110x120mm.  
 
Fig.5. UWB Vivaldi antenna.  
The Vivaldi antenna is excited via the microstrip to slotline 
transition, see Fig. 5. The transition construction exploits 
wideband features of a microstrip radial stub used as a virtual 
wideband short. Thus the microstrip is virtually shunted to the 
second half of the slotline metallization while the first half 
serves as a ground metallization for the microstrip line. Since 
the slot line is terminated by an open end the energy can 
propagate through transition without any reflection and 
insertion loss in an ideal case. Unfortunately, the real 
transition has insertion loss about 1.8 dB at 10.6 GHz, which 
is caused by radiation from the transition. The nominal line 
impedance 100 ohm was chosen as a trade off between 
radiation and slotline gap dimension at the transition. The 
angle of the radial stub is 60 degrees and the length was 
adjusted using Microwave Office® in order to obtain the 
reflection coefficient as close as possible to a short over the 
frequency band 3.1-10.6 GHz.  
The antenna was designed on Taconic TLX-8 substrate  
(εr =2.55, h=0.76 mm, td=0.0019 at 10GHz). Results of the 
modeling of the antenna using CST MicroWave Studio® are 
shown in Fig. 7 and Fig.8.  
 
Fig.7. Simulation of the return loss of the designed Vivaldi antenna.  
 
Fig.8. An UWB pulse distortion caused by antenna.  
C. Receiver 
1) Receiving antenna 
The antenna used at the reception side is of Vivaldi type the 
same type as that used in the transmitter.  
2) Low noise amplifier 
 Low noise amplifier is one of the most important 
elements of UWB receiver. Its aim is to amplify very weak                  
LO 
RF 
IF 
 (-100dBm/MHz of power spectral density) and broadband 
incoming signal, which appears at antenna's aperture, to the 
appropriate level which ensures non-problematic detection 
and retrieving the data. Moreover, since the power spectral 
density of UWB pulse just detected by the antenna is very 
low, broadband amplifier has to contribute as low noise 
energy to the amplified signal as possible. Therefore LNA for 
the UWB receiver has to meet some particular requirements, 
for instance: low-noise figure, high gain, good input and 
output matching and constant group delay in order to give 
amplified non-distorted signal. Those parameters have serious 
influence in the overall sensitivity of the receiver. Since UWB 
transceiver is tend to be employed in the remote devices, it has 
to have low power consumption. 
Low noise amplifier employed in our system was design to 
reach the lowest possible NF. The Agilent ATF-36163 
transistor was chosen mainly because provides NF less than    
1 dB with associated high gain (16 – 12 dB) in the frequency 
range of 3-8 GHz. The amplifier was designed with the aid of 
Microwave Office® CAD tool, on Taconic CER-10 substrate 
with thickness 0.635 mm and 35 µm for dielectric substrate 
and copper metallization layers, respectively.  The final 
amplifier layout is shown in Fig.9. 
 
Fig.9. Layout of the low noise amplifier. 
Drain and gate biasing networks were designed to reduce 
leakage of the RF signal to the DC power supply circuit and to 
give negligible effect on the overall gain of the amplifier. 
They were designed to provide good isolation in the whole 
UWB frequency range regulated by FCC. In Fig.10 frequency 
response of drain biasing circuit is shown. 
0.5 5.5 10.5 15.5 18
Frequency (GHz)
S11 and S21 of the drain biasing network
-100
-80
-60
-40
-20
0
DB(|S(1,1)|)
drain biasing circuit5
DB(|S(2,1)|)
drain biasing circuit5
 
Fig.10. Frequency response of the drain biasing network. 
Both biasing networks consist of the narrow 75 Ω 
characteristic impedance microstrip line followed by double 
radial stub and 210 Ω (drain) or 50 Ω (gate) resistors.  
The 1 nF capacitor shunts the RF signal to the ground. 
Utilization of double radial stubs provides better isolation in 
the frequency band of interest.   
Input matching network was designed to provide the lowest 
possible NF in the range of 3-8 GHz, while output network 
gives best power matching. Simulated gain and NF of the 
amplifier are given in Fig.11. 
2 4 6 8 10
Frequency (GHz)
Gain and NF of the amplifier
-20
-10
0
10
20
6 GHz
1.181 dB
6 GHz
13.6 dB
3 GHz
7.513 dB
8 GHz
7.447 dB
8 GHz
1.573 dB
3 GHz
4.573 dB
DB(|S(2,1)|)
amplifier
DB(NF())
amplifier
 
Fig.11. Simulated gain and noise figure of the amplifier. 
The measurements taken with Rhode & Schwarz spectrum 
analyzer FSU 26 are shown in Fig.12. 
Fig.
0
0
0
 NF12 
12. 
1 2
Measured gain and NF of the amplifier. 1Gain
 3) The second amplifier 
The second stage of the amplifier was realized by MMIC 
commercially available, surface mount broadband amplifier 
ERA21SM. The measured frequency response in range from 
1-11GHz of the amplifier mounted on Taconic Cer-10 
substrate is shown in Fig.13. 
 
 
Fig.13. Measured ERA21SM gain and noise figure.  
4) Pulse detector  
The concept of the detector is based on fast monostable 
flip-flop circuit that provides strong positive feedback [3]. 
Therefore, a fast time response to the input pulse can be 
expected. A schematic diagram of the detector model is 
depicted in Fig.14.   
The first stage of the detector, containing transistor Q1, 
shapes an arbitrary input pulse to one negative trigger peak. 
This is illustrated in Fig.15 . This negative peak triggers the 
following monostable flip-flop circuit consisting of transistors 
Q2 and Q3. In order to minimize the influence of load 
impedance on the monostable flip-flop circuit the buffer 
transistor Q4 is placed at the end. The measured response of 
the detector is presented in Fig. 16. 
 
Fig.14. Spice model of the pulse detector. Subcircuit Q1-Q4 contains 
Gummel-Poon model of bipolar transistor BFG 410 W including parasitic 
elements of its package SOT343R. 
 
Fig. 15. Simulated waveforms of the pulse detector. 
 
 
Fig.16. Measured waveform at the detector output for the PRF=20 MHz and 
the attenuation 6 dB (positive peak value 329 mV) of the excitation pulse.  
III. THE UWB LINK MEASUREMENTS 
The measurements of the UWB link were carried out with 
the setup presented in Fig.2. At the beginning performance of 
the mixer as a fast multiplier has been checked. Pulse of the 
parameters presented in Table I and waveform in Fig.4 was 
used as a baseband switching signal and provided to IF mixer 
input. The RF frequency was supplied from the signal 
generator set to 3.9 GHz and power of 10d Bm. The output 
signal was supplied directly to Vivaldi antenna.  
Fig.17 presents waveforms of the signals in time and 
frequency domain. In Fig.17a) spectrum of the baseband 
signal is shown – useful -10dB bandwidth spreads from DC 
up to above  
2 GHz with the power spectral density -122 dB/Hz around  
0.5 GHz. The up-converted pulse and its spectrum can be seen 
in Fig. 17b) and c) respectively. The RF burst lasts for around 
0.7ns and has the center frequency of the 3.9 GHz as it is 
presented in Fig.17c). The maximum PSD is -137 dB/Hz 
(FCC allows -131dB/Hz), -10 dB bandwidth covers 
frequencies from 2.8-5.2GHz giving 61% of fractional 
bandwidth (FCC requires at least 25%). Peak to peak values 
of the received signal over short distances, mainly because of 
the limitations of the low gain of the used amplifiers and the 
length of the connecting cables, are gathered in Table II.  
Gain 
NF 
0 
10 
20 
 Fig.1
UWB
V. CONCLUSION 
 a) 
The principles of UWB link has been tested with the 
designed in house setup. Simple arrangement proves low 
complexity of the UWB transceiver. Even though the 
experiment does not take into account such issues as multiple 
access to the medium, synchronization etc. it shows that 
hardware arrangement promises low cost, low complex radio 
for future applications with respectable data rates.  
ACKNOWLEDGMENT 
This work, part of the M-Zones Project (www.m-zones.org) 
has been supported under the Programme for Research in 
Third Level Institutions (PRTLI) by Irish Higher Education 
Authority (HEA). 
 ) b 
REFERENCES 
[1] G. F. Ross “The Synthetic Generation of Phase-coherent Microwave 
signals for Transient Behaviour Measurements”, IEEE MTT 
Transactions,1965, 704-705  
[2] P.Piksa, V.Sokol, “Small Vivaldi Antenna for UWB”, internal report, 
CIT 2005 
[3] V. Sokol, P. Rulikowski and J. Barrett, “Low Power Consuming Pulse 
Detector”, submitted for publication at 2005 IEEE International 
Conference on Ultra-Wideband, Zurich 2005  
[4] P. Rulikowski and J. Barrett, “Truly Balanced Step Recovery Diode 
Pulse Generator with Single Power Supply,” in IEEE RAWCON 
Conference Proc., Atlanta 2004 
     
 
 c) 
7. a) Baseband pulse spectrum, b) UWB pulse waveform after mixer, c) 
 signal spectrum.  
TABLE II  
RECEIVED SIGNAL AMPLITUDE VS DISTANCE 
Distance 
[cm] 
Peak to peak voltage at the 
input of the detector 
[mV] 
33 457 
27 562 
22 670 
19 721 


The Experimental UWB Link

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The Experimental UWB Link

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