Direct Sequence Spread Spectrum (DS-SS) signals exhibit cyclostationary properties which imply a redundancy between frequency components separated by multiples of the symbol rate. In this paper we present
a Multiple Access Interference Canceller that explores this property and applies to UMTS-TDD. This frequency domain canceller operates in the spreaded signal in such way that the interference and noise at its output is
minimized (Minimum Mean Squared Error Criterium). The performance is evaluated in two con gurations: one including the Frequency Shift Canceller (FSC) and the other concatenated with Parallel Interference Canceller(PIC). The results are benchmarked against the performance of the conventional RAKE detector and the conventional PIC detector

Gameiro, A.

Gonçalves, L.

Gonçalves, Luís

Gameiro, Atílio

Repositório Institucional da Universidade de Aveiro

FREQUENCY SHIFT BASED MULTIPLE ACCESS INTERFERENCECANCELLER FOR MULTIRATE UMTS-TDD SYSTEMSLuis Gonc‚alves, Atilio GameiroInstituto de Telecomunicac¸o˜es, Campus Universita´rio de Santiago, 3810-193 Aveiro, PortugalDepartamento de Electro´nica e Telecomunicac¸o˜es, Universidade de Aveiro, Portugalemail: lgoncalves@av.it.pt, amg@det.ua.ptAbstract— Direct Sequence Spread Spectrum (DS-SS) signals exhibit cy-clostationary properties which imply a redundancy between frequency com-ponents separated by multiples of the symbol rate. In this paper we presenta Multiple Access Interference Canceller that explores this property andapplies to UMTS-TDD. This frequency domain canceller operates in thespreaded signal in such way that the interference and noise at its output isminimized (Minimum Mean Squared Error Criterium). The performanceis evaluated in two configurations: one including the Frequency Shift Can-celler (FSC) and the other concatenated with Parallel Interference Canceller(PIC). The results are benchmarked against the performance of the conven-tional RAKE detector and the conventional PIC detector.Keywords—cyclostationary, redundant , frequency shift, PIC, DS-CDMAI. INTRODUCTIONDirect sequence spread-spectrum (DS-SS) code division mul-tiple access (CDMA) has emerged as one of the most promisingtechniques to implement various radio communication systems.It presents significant advantages over Time Division MultipleAccess (TDMA), namely frequency diversity, multipath diversityand more spectrum efficiency on multicell systems [1], which ledto its choice as the technology for third generation cellular sys-tems. The second generation CDMA systems were based on theconventional RAKE receiver, which is known to be limited bythe multiple access interference (MAI) and require a very pre-cise power control. To overcome these limitations and thereforeenhance the capacity of CDMA systems, joint detection of thereceived DS-SS signals has been proposed to be used at the basestation (BS) or at the user equipment. The optimum joint de-tector [2] although well known requires however a prohibitivelyhigh computational complexity, and consequently effort has beenmade to devise suboptimum algorithms with good compromisebetween performance and complexity that can be implementedwithout prohibitive costs in near future CDMA systems. Thiscommunication fits in this approach, and aims at presenting amoderate complexity MAI canceller operating on the broadbandDS signal. The Frequency Shift Detector can be used either asstandalone unit or it can be used prior to a PIC multiuser detectorwhere it is intended to produce signals clean enough so that thefirst decisions of the PIC can be considered reliable enough to beused by the subsequent stages.The DS-SS signal is a particular case of a stationary randompulse amplitude modulation of pulses (symbols). This kind ofsignals are known to have cyclostationary properties [3], [4].Those properties imply redundancy between frequency compo-nents separated by multiples of the symbol rate. It is this charac-teristic that we explore to propose a new MAI canceller.In previous work [5] was defined and implemented the FSCfor fixed spreading factor. The results show that considerableThis work is being supported by a PhD Grant from PRAXIS XXI, Fundaca˜opara a Cieˆncia e Tecnologia and the projects ASILUM and VISEFimprovement is achieved against the RAKE and the FSC evenoutperforms the PIC for moderate to high values of Eb/N0. TheFSC was also tested as a precanceller to be used prior to a PICand the results have shown that nearly complete interference can-cellation in a UMTS-TDD scenario with up to eight simultaneoususers in the same slot.In the present paper we extend the work of [5] to includethe general situation of UMTS-TDD where users with differentspreading factors can coexist in the same time slot. This enablesthe transmission of services with different rates by varying thespreading factor, but the structure of the spreading (channeliza-tion times scrambling) in UMTS-TDD lead to codes that extendover several symbols.The paper is outlined as follows. In section two we show thatin a DS signal non overlapping frequency bands separated by amultiple of the baud rate are linearly related. This result is usedto present in section three the architecture of a MAI canceller thatexplores this redundancy. In section four we present simulationresults that illustrate the performance provided by the new can-celler. Finally in section five the main conclusions of this workare outlined.II. THEORETICAL BACKGROUNDThe UMTS-TDD standard was designed to accommodate sev-eral simultaneous transmission rates. The different rates are ac-complished by varying the spreading factor. The spreading codeis composed by the product between the channelization code andthe scrambling code. The channelization code lasts for one sym-bol and its number of chips is equal to the spreading factor. Thescrambling code lasts for sixteen chips (the maximum spreadingfactor Qmax) or during QmaxQ symbols being Q the spreading fac-tor. Then to construct the spreading code the channelization codemust be repeated QmaxQ and multiplied by the scrambling code.The spreading code extends for more than one symbol in the caseof Q 6= Qmax. In the case of UMTS-TDD, Qmax = 16.This leads to the following representation of the single userDS-SS signal:s(t) =∑kQmaxQ−1∑l=0alkgl(t− lT −QmaxQ kT ) (1)where {alk} are the sequences of information symbols,1T thesymbol rate and gl(t) are the components of signature waveform.To obtain these components the spreading code (length equal toQmax) is divided in QmaxQ sequences (each equal to one compo-nent) of length Q and then each one is pulse shaped and affectedby channel estimates.The Fourier Transform of this signal is:S(f) =∑kQmaxQ−1∑l=0alkGl(f)e−j2piflT e−j2pifQmaxkT/Q=QmaxQ−1∑l=0Gl(f)e−j2piflT∑kalke−j2pifQmaxkT/Q(2)Notice that ifAl(f) =∑kalke−j2pifQmaxkT/Q (3)thenAl(f +iQQmaxT) = Al(f) ∀i ∈ Z (4)Defining the vectors:a(f ) =[A0(f) ... AQmax/Q−1(f)](5)g(f ) =[G0(f) ... GQmax/Q−1(f)e−j(Qmax/Q−1)2pifT](6)we can write (2) in a matrix form:S(f) = g(f )aT(f ) (7)where T means transpose.In the same way if we define:s(f ) =[S′(f) S′(f + QTQmax ) ... S′(f + 1T −QTQmax)](8)whereS′(f) =S(f) se f ∈[0, QTQmax[0 se f /∈[0, QTQmax[ (9)S(f) is defined in (2) and (7).Defining:gi(f ) =[G′0 (f +iQTQmax) ...G′Qmax/Q−1(f + iQTQmax )e−j(Qmax/Q−1)2piT (f+iQT Qmax)] (10)where i ∈ {0, 1, ..., Qmax/Q− 1} and:G′l(f) =Gl(f) se f ∈[0, QTQmax[0 se f /∈[0, QTQmax[ (11)Gl(f) is defined in (2).Then:sT(f ) = g0(f )..gQmax/Q−1(f ) aT(f ) = G(f )aT(f ) (12)sT(f + mT) = G(f + mT)aT(f ) (13)and then:sT(f ) = G(f )G−1(f + mT)sT(f + mT) (14)This means that we can relate the signal information in frequencybands spaced by a multiple of the baud rate by a linear transfor-mation.The conversion method between redundant bands expressed by(14) is the base of the FSC and represents a generalization of thetechnique developed in [5] where the use of a fixed spreadingfactor resulted a scalar transformation.III. PRINCIPLES OF THE CANCELLERLet the representation of signal in the time domain at the inputof the canceller be r(t) =U∑u=1s(u)(t) + n(t) where U is thenumber of users and n(t) is stationary noise with power spectraldensity ηin(f). We consider without loss of generality that userone (u = 1) is the user of interest. The canceller operates inthe frequency domain and in a practical implementation or in asimulation system the time to frequency domain conversion isperformed digitally through a FFT. In the following derivationwe adopt a continuous signal notation.The objective and design criteria for the frequency domain can-celler is to minimize the overall disturbance (MAI+noise) subjectto the condition that s(1)(t) must not be distorted.Let R(f) and V (f) be the fourier transform of r(t) and v(t)(representation on the time domain of the output signal of theFSC) respectively, Q(1) the spreading factor of the user one andG(u)(f) the matrix given by (12) (G(f)) where we have addedthe subscript (u) to indicate that it refers to user u.We define the vectors r(f ) and v(f ) as:r(f ) =[R′(f) R′(f +Q(1)TQmax) ... R′(f + 1T −Q(1)TQmax)](15)v(f ) =[V′(f) V′(f +Q(1)TQmax) ... V′(f + 1T −Q(1)TQmax)](16)where:R′(f) =R(f) se f ∈[0,Q(1)T(1)Qmax[0 se f /∈[0,Q(1)T(1)Qmax[ (17)V′(f) have a identical definition as R′(f) where R(f) is re-placed by V (f).The FSC uses the information of bands i to reconstruct bandsm for the user of interest according to (14) while minimizing theoverall distortion. In the frequency domain the output signal isrelated with the input one through the following relation:vT(f + mT(1) )=∑iαi,m (G(1)(f +mT(1))G−1(1)(f +iT(1))rT(f + iT(1)))(18)where αi,m is a set of complex vectors with size QmaxQ(1) whoseselection is intended to minimize the Multiple Access Interfer-ence and noise at the output.  designates an element by elementmultiplication between matrices or vectors.Let sT(u)(f) be the matrix given by (12) where we have addedthe subscript (u) to indicate that it refers to user u. Let a(u)(f)be the matrix given by (5) where we have added the subscript (u)to indicate that it refers to user u.After some processing we obtain from (18):vT(f + mT(1)) = sT(1)(f +mT(1))∑iαi,m+U∑u=2∑iαi,m (G(1)(f +mT(1))G−1(1)(f + iT(1))hT(u)(f +iT(1)))+ n′T(f + iT(1))(19)ChannelParametersFE G HCBA PathAlignmentFFTFrequencyShiftCancellerDownSampling IFFTMatchedFilterDownSamplingDFig. 1. Receiver with FSCwherehT(u)(f) =∑zW(u),z(f)aT(u),z(f) (20)• If the spreading factor of the interfering signal u is equal tothe signal of interest the sum in (20) has only one term andW(u),1(f) = G(u)(f), a(u),1(f) = a(u)(f) and h(u)(f) =s(u)(f).• If the spreading factor of the interfering user is not equal to thespreading factor of the user of interest then h(u)(f ) 6= s(u)(f ) be-cause of incompatibility between dimensions of matrices in (19).– If the spreading factor of a interfering user u is less thanthe user of interest then we must decompose sT(u)(f) =G(u)(f )aT(u)(f ) in a sum of several products of matrix timesvector (W(u),z(f)aT(u),z(f)) to obtain hT(u)(f ) taking care toget a good representation of the signal in question.– If the spreading factor of the particular user u is greaterthan the user of interest then we must divide the signaturewaveform in the same number of components of the userof interest (see Section II) generating G(u)(f ) of the samedimension of G(1)(f ). In that situation h(u)(f ) is equal tos(u)(f).Let x(u),i,m,z(f) be the vector line which elements are of thefirst line of G(1)(f + mT(1))G−1(1)(f + iT(1))W(u),z(f + iT(1))(see (19) and (20)).Let α0i,m be the first element of each αi,m and N′0(f) the firstelement of n′(f).The first line of the vector vT(f − mT(1)) in (19) is:V′(f + mT(1))= S′(1)(f + mT(1)) ∑iα0i,m+U∑u=2∑iα0i,m∑zx(u),i,m,z(f)aT(u),z(f) + N′0(f + mT(1))(21)Let ηin(f) be the power spectral density of the noise at theinput of the canceller. Defining the vector line:ηin(f ) =[η′in(f ) η′in(f +Q(1)T(1)Qmax) ...η′in(f +1T(1)−Q(1)T(1)Qmax)] (22)where:η′in(f) =ηin(f) se f ∈[0,Q(1)T(1)Qmax[0 se f /∈[0,Q(1)T(1)Qmax[ (23)The power spectral density of the noise N′0(f + mT(1))in (21)is:η′out(f + mT(1))=∑i∣∣α0i,m∣∣2 M(f )ηTin (f + iT ) (24)where M(f ) is the vector line with elements |w1j |2 which w1j isthe (1, j) element of G(1)(f + mT(1))G−1(1)(f + iT(1)).Let Xe(u),i,m,z(f) be a element of x(u),i,m,z(f) (see (21))withe ∈{1, 2, ..., QmaxQ(1)}.The design criteria implies that the weights α0i,m are dimen-sioned so that:CU∑u=2∑z∑e∫f∣∣∣∣∣∑iα0i,mXe(u),i,m,z(f)∣∣∣∣∣2df+TBt∫fη′out(f + mT(1))df(25)is minimized to the condition that∑iα0i,m = 1. C is equal tothe number of symbols of the user of interest in each burst timesQ(1)Qmax. In (25) TBt is the burst duration time. Notice that thepower of each symbol in UMTS-TDD is one. This process mustbe repeated for all the lines of vT(f + mT(1)) and for all bands m.In the previous approach we consider that f ∈[0,Q(1)TQmax[(see(17)). We can divide this interval in several intervals and applythe procedure presented in this section for each of the intervalsincreasing the performance. The shorter each interval the betterthe performance is.IV. APPLICATION OF THE CANCELLER TO UMTS-TDDIn this section we present some numerical results illustratingthe performance of the proposed detector configurations withUMTS-TDD signals. The proposed configurations are more suit-able to be implemented in the uplink because the detectors re-quire knowledge of the spreading codes of the active users. Toevaluate the canceller performance a simulation chain was imple-mented. Basically this simulation chain is composed by a trans-mitter, a transmission channel and a receiver.A. TransmittersThe transmitters are compliant with the 3GPP specificationsfor UMTS-TDD.B. Channel ModelThe channel model used in this work was the GeometricalBased Single Bounce Elliptical Model (GBSBEM) proposed byLiberti [6]. This model was developed for microcell and pic-ocell environments. The propagation channel is characterizedby L paths for each user, one in line of sight plus L − 1 ar-riving from remote reflectors located randomly within an ellip-sis where the base station and the mobile unit are at the foci.Each path is characterized by complex constant and a delay. Thedelay is uniformly distributed between zero and the maximumdelay spread. The phase of the complex constant is uniformly   H Channel Parameters J A  PIC Frequency Shift Canceller   Fig. 2. Receiver including FSC plus PIC0 4 8 12 16 2010−410−310−210−1 Eb/No (dB) Average BER RAKE       FSC        PIC        FSC−PIC    Single UserFig. 3. BER for users with spreading factor of sixteen, in the first scenariodistributed in [0, 2pi[. The amplitude of the complex constantfollows a Rayleigh distribution. The channel parameters are as-sumed to be constant within each burst.C. ReceiversFigure 1 depicts the basic configuration for the detector thatincludes the Frequency Shift Canceller. If we remove the Fre-quency Shift Canceller block the detector is a conventionalRAKE. The Path Alignment and Downsampling are blockswhose operations are done in frequency domain despite the factthat the names reflect the correspondent time domain operations.The path alignment includes delay alignment and maximum ratiocombining of the spreaded signal. The signal in A (figure 1) hasa resolution of four samples per chip and the first downsamplinghas the same factor. The second downsampling takes a factorequal to the spreading factor. In H we have the signal beforesymbol decision. Notice that the Discrete Fourier Transform ismade only once in each slot (and burst).The second detector configuration to be evaluated is the de-tector composed by FSC plus PIC (figure 2). The FSC in figure2 corresponds to the whole receiver chain of figure 1. The FSCprovides the signal for the symbol decision inside the PIC.These two configurations are benchmarked with the conven-tional RAKE and conventional single stage PIC.D. ResultsThe simulations were made with the parameters shown in Ta-ble I. The channel parameters estimates were assumed to be per-fect.0 4 8 12 16 2010−410−310−210−1 Eb/No (dB) Average BER RAKE       FSC        PIC        FSC−PIC    Single UserFig. 4. BER for users with spreading factor of eight, in the first scenario0 4 8 12 16 2010−410−310−210−1 Eb/No (dB) Average BER RAKE       FSC        PIC        FSC−PIC    Single UserFig. 5. BER for users with spreading factor of four, in the first scenarioFollowing we present some results for two scenarios.• First Scenario:– Eight users: Four with spreading factor of sixteen, twowith spreading factor of eight and two with spreading factorof four (Full load system).– The users with spreading factor eight and four have powersabove the users of spreading factors of sixteen of 3 and 6dBrespectively.• Second Scenario:– Four users: Two with spreading factor of sixteen, one withspreading factor of eight and one with spreading factor offour (Half load system).– The users with spreading factor eight and four have powersabove the users of spreading factors of sixteen of 3 and 6dBrespectively.Plots of the BER achieved are shown in Figures 3-8. Figures 3,4 and 5 refer to the first scenario and apply to users with spreadingfactors of 16, 8 and 4 respectively.In each plot we compare the following configurations: RAKE,0 4 8 12 16 2010−410−310−210−1 Eb/No (dB) Average BER RAKE       FSC        PIC        FSC−PIC    Single UserFig. 6. BER for users with spreading factor of sixteen, in the second scenarioTABLE ISIMULATION PARAMETERS SETTINGSSpreading Factor 16, 8, 4Number of Taps 2Channel GBSBEMVelocity 50 Km/hPath Loss 3.7BurstType 1Maximum Delay Spread 2.0 µsDegrees of Freedom of FSC 16, 8, 4Number of samples per chip 4Line of Sight Distance 300mPIC, FSC, FSC+PIC and Single User. The BER shown for agiven spreading factor is the average of the various users withthe same spreading factor.The same kind of comparisons are presented in Figures 6-8 forthe second scenario.The results show that significant gains are obtained by the FSCand FSC+PIC over the RAKE receivers. For the case of halfloaded system the FSC+PIC nearly achieves the single user per-formance limit.It was noticed that the results presents some sensivity in rela-tion with the specific scrambling code used. This issue is cur-rently under study.V. CONCLUSIONSIn this communication we presented a new linear canceller op-erating in the frequency domain that takes advantage of frequencyredundancy of the DS-SS signals. This Multirate Canceller wasapplied in a UMTS-TDD simulation chain in two scenarios: onein a full loaded system and one in a half loaded system. In bothscenarios significant performance gains were obtained. When0 4 8 12 16 2010−410−310−210−1 Eb/No (dB) Average BER RAKE       FSC        PIC        FSC−PIC    Single UserFig. 7. BER for users with spreading factor of eight, in the second scenario0 4 8 12 16 2010−410−310−210−1 Eb/No (dB) Average BER RAKE       FSC        PIC        FSC−PIC    Single UserFig. 8. BER for users with spreading factor of four, in the second scenariooperating in concatenation with a PIC the FSC nearly achievessingle user performance in the case of a half loaded system.REFERENCES[1] Ryuji Kohno, Reuven Meidan and Laurence B. Milstein, Spread SpectrumAccess Methods for Wireless Communications, IEEE Communications Mag-azine, January 1995[2] Sergio Verdu, Minimum Probability of Error for Asynchronous GaussianMultiple-Access Channels, IEEE Transactions of Information Theory, Jan-uary 1986[3] Gardner, William A., Cyclostationarity in Communications and Signal Pro-cessing, IEEE PRESS, 1994[4] Gardner, William A., Exploitation of Spectral Redundancy in Cyclostation-ary Signals, IEEE Signal Processing Magazine, April 1991[5] Luis Gonc¸alves, Ada˜o Silva, Atilio Gameiro, Frequency Shift Based MultipleAccess Interference Canceller for DS-CDMA Systems , Vehicular Technol-ogy Conference 2001 Fall, October 2001, Atlantic City, USA.[6] J. C. Liberti, T. S. Rappaport, Smart Antennas for Wireless Communications:IS-95 and Third Generation CDMA Applications, Prentice Hall, 1999.

Frequency shift based multiple access interference canceller for multirate UMTS-TDD systems

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Frequency shift based multiple access interference canceller for multirate UMTS-TDD systems

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