In a software-defined radio (SDR) receiver it is desirable to minimize RF band-filtering for flexibility, size and cost reasons, but this leads to increased out-of-band interference (OBI). Besides harmonic and intermodulation distortion (HD/IMD), OBI can also lead to blocking and harmonic mixing. A wideband LNA [1, 2] amplifies signal and interference with equal gain. Even a low gain of 6dB can clip 0dBm OBI to a 1.2V supply, blocking the receiver. Hard-switching mixers not only translate the wanted signal to baseband but also the interference around LO harmonics. Harmonic rejection (HR) mixers have been used [3, 1, 4], but are sensitive to phase and gain mismatch. Indeed the HR in [4] shows a large spread, whereas other work only shows results from one chip [3, 1]. This paper describes techniques to relax blocking and HD/IMD, and make HR robust to mismatch

Klumperink, E.A.M.

Nauta, B.

Ru, Z.

Wienk, G.J.M.

English

Klumperink, Eric A.M.

Wienk, Gerhardus J.M.

Nauta, Bram

University of Twente Research Information

230 •  2009 IEEE International Solid-State Circuits ConferenceISSCC 2009 / SESSION 12 / RF BUILDING BLOCKS / 12.812.8 A Software-Defined Radio Receiver Architecture Robust to Out-of-Band InterferenceZ. Ru, E. A. M. Klumperink, G. J. M. Wienk, B. NautaUniversity of Twente, Enschede, NetherlandsIn a software-defined radio (SDR) receiver it is desirable to minimize RF band-filtering for flexibility, size and cost reasons, but this leads to increased out-of-band interference (OBI). Besides harmonic and intermodulation distortion(HD/IMD), OBI can also lead to blocking and harmonic mixing. A widebandLNA [1, 2] amplifies signal and interference with equal gain. Even a low gainof 6dB can clip 0dBm OBI to a 1.2V supply, blocking the receiver. Hard-switch-ing mixers not only translate the wanted signal to baseband but also the inter-ference around LO harmonics. Harmonic rejection (HR) mixers have beenused [3, 1, 4], but are sensitive to phase and gain mismatch. Indeed the HRin [4] shows a large spread, whereas other work only shows results from onechip [3, 1]. This paper describes techniques to relax blocking and HD/IMD,and make HR robust to mismatch.The zero-IF architecture described targets the 0.4-to-6GHz band, where mostmobile applications reside. The baseband can be shared if the low-pass filter(LPF) is made flexible. As shown in Fig. 12.8.1 (bottom), only two externalfilters for 0.4 to 2.5GHz and 2.5 to 6GHz are used. The 0.9 to 6GHz signals aredownconverted by the lower quadrature mixer using a 4-phase LO, while the0.4-to-0.9GHz signal needs an HR-mixer with 8-phase LO to reject 2nd-to-6thharmonics (0.4GHz·7 and 0.9GHz·3 out of filter band). With only one 0.4-to-6GHz band filter, an 8-phase LO up to 2GHz would be needed, leading to highpower consumption and low phase accuracy. This work focuses on a 0.4-to-0.9GHz receiver to demonstrate techniques to counter OBI.Figure 12.8.1 (top) shows the implemented receiver, consisting of low-noisetransconductance amplifiers (LNTA) with input matching added, mixers driv-en by a divide-by-8 and two cascaded IF-amplifier stages with LPF. ThreeLNTAs have a gain ratio of 2:3:2 to approximate 1:√2:1 for HR. The LNTAdelivers current to the switching mixers. The voltage amplification only occursafter mixing to IF simultaneously with a first-order RC LPF to suppress OBI.However, such a filter cannot remove OBI already folded over the wanted sig-nal due to harmonic mixing. Since harmonics can be as strong as blockers,HR is wanted before the first voltage amplification. The mixers driven by 8-phase clocks provide 2nd-to-6th-order HR.To reject 3rd and 5th harmonics we can employ gain factors 1:√2:1 and 45°phase shift [3]. Traditionally zero-IF HR mixers deliver quadrature outputs.Instead, we use a 2-stage polyphase HR idea exploiting 8-phase mixer outputs(Fig. 12.8.1). The 2nd-stage HR employs a gain-ratio of 5:7:5 (via resistor ratios7:5:7), and the 45° phase shift comes from the 8-phase baseband signal. The2-stage HR provides two benefits: a) it uses integer ratios which can be real-ized accurately on chip to approximate √2 closely; b) it reduces the total gainerror, e.g. due to mismatch or parasitics, to a product of gain errors (1%becomes 0.01%). Figure 12.8.2 (top) shows a calculation of how √2 isapproximated to 0.03% error as 41:29 by simple integer 2:3:2 and 5:7:5ratios. This 41:29 ratio in the effective LO amplitude is constructed via threesignal paths, each with a weighing factor of stage 1 (time-dependent factor 0,2 or 3 in the array) and stage 2 (constant 5 or 7). Figure 12.8.2 (bottom)shows how, for the desired signal, polyphase contributions from three pathsadd up, while for the 3rd and 5th harmonics, they cancel nominally. As twostages are cascaded, the product of the gains determines the result. Thismeans that the total relative error (αβ/4) is the product of the relative errors(1st stage: α/2, 2nd stage: β/2). If the 2nd stage has 1% error (β), ideally thisimproves HR by (β/2)-1, i.e. 46dB.With such a large reduction of gain errors, phase errors are likely to dominate.We use a power-efficient divider with minimum phase-error accumulation.The divide-by-8 is a loop with eight dynamic transmission gate (TG) flip-flops(FF), one of which is shown in Fig. 12.8.3 (top). The same master clock (CLK),with 8-times the LO-frequency, clocks all FFs. Only one inverter (INV2) is usedas a buffer to minimize the path from CLK to mixer, to minimize phase mis-match. A preset data pattern is required to deliver the wanted 1/8 duty cycle.The simulated 3σ phase error is only 0.34° at 0.8GHz LO including contribu-tions from mixer switches.Figure 12.8.3 (bottom) shows the LNTA. M1 provides input matching whilethe input is also connected to the AC-coupled inverter (M2, M3). The sourceof M1 is biased to GND via an external inductor. Since the LNTA outputs seea low TIA input impedance, the output swing is small and the output distor-tion is low. The distortion caused by in-band interference is suppressed by thenegative feedback (Fig. 12.8.1). The loop gain degrades at high frequency, butthe LPF suppresses OBI, improving out-of-band linearity. Out-of-bandHD/IMD is then dominated by the V-to-I function of the LNTA. The TIA is basedon a two-stage OTA with a Class-AB output for high swing [5]. A receiver withLNTA and TIA combination has been presented in [6], however using a nar-rowband LNTA and no HR. To the authors’ knowledge this work proposes thefirst wideband receiver exploiting the baseband LPF to suppress OBI. In [7], afeedforward path, using mixers and first-order high-pass filter, cancels OBI atthe LNA output, but significantly degrades noise.The chip occupies a total area of 1mm2 (excluding pads) in 65nm CMOS (Fig.12.8.7). Capacitors take a large portion of area in the TIA, and also the OTAinput pair is big to achieve a low 1/f noise corner (measured at 30kHz). Toprove the receiver is robust to OBI, all measurements are performed on PCBwithout external filter. Figure 12.8.4 (top) shows the measured gain, NF andin-band IIP2/IIP3. The NF degrades at 0.4GHz due to 1/f noise of the LNTAsince minimum-length transistors are used for wideband low S11 to show validHR at high frequency (measured S11<-10dB up to 5.5GHz). The frequencyrange (<1GHz) is limited by clock speed, but the signal path has >5GHz BW(simulation). Figure 12.8.4 (bottom) shows signal gain versus blocker power.The measurement was carried out to imitate DVB-H signals (470MHz and862MHz) with a GSM blocker at 935MHz. The signal gain is reduced by 1dBat a blocker power of -5dBm (470MHz) and -12dBm (862MHz) respectively,well above the measured 1dB compression point of -22dBm, showing that theblocker filtering is effective. Smaller LPF BW or higher filter order can providemore filtering.Figure 12.8.5 (top) plots the HR ratio (HRR) from 1-stage only as well as thetotal 2-stage HR of a typical sample. A dramatic improvement of 30dB for both3rd and 5th HR is found, thanks to the 2-stage HR technique. In general, theimprovement is in the range of 20 to 40dB for all samples. Figure 12.8.5 (bot-tom) shows the 2-stage HRR of 40 randomly-selected samples measured at0.8GHz LO. The minimum 3rd HRR is 60dB and 5th HRR is 64dB. All even-orderHRR are >60dB. Without extra RF filtering, the 3rd and 5th HRR in the literatureare around 40dB [1, 4]. This work achieves a minimum of 60dB without anytrimming or calibration. Figure 12.8.6 summarizes the measured parameters.The +18dBm out-of-band IIP3 shows the good linearity of the LNTA.Acknowledgements:The work is funded by Freeband. We thank H. Brekelmans, D. Leenaerts, H. de Vries, andNXP for chip fabrication.References:[1] R. Bagheri, A. Mirzaei, S. Chehrazi et al, “An 800MHz-to-5GHz Software-DefinedRadio Receiver in 90nm CMOS”, ISSCC Dig. Tech. Papers, pp. 480-481, Feb. 2006.[2] J. Craninckx, M. Liu, D. Hauspie et al, “A Fully Reconfigurable Software-DefinedRadio Transceiver in 0.13µm CMOS”, ISSCC Dig. Tech. Papers, pp. 346-347, Feb. 2007. [3] J. Weldon, J. Rudell, L. Lin et al, “A 1.75GHz Highly Integrated Narrowband CMOSTransmitter with Harmonic-Rejection Mixers”, ISSCC Dig. Tech. Papers, pp. 160-161,Feb. 2001.[4] Z. Ru, E. Klumperink and B. Nauta, “A Discrete-Time Mixing Receiver Architecturewith Wideband Harmonic Rejection”, ISSCC Dig. Tech. Papers, pp. 322-323, Feb. 2008.[5] S. Rabii and B. Wooley, “A 1.8V Digital-Audio ΔΣ Modulator in 0.8µm CMOS”, IEEEJ. Solid-State Circuits, vol. 32, no. 6, pp. 783-796, June 1997.[6] M. Valla, G. Montagna, R. Castello et al, “A 72mW CMOS 802.11a Direct-ConversionFront-End with 3.5dB NF and 200kHz 1/f Noise Corner”, IEEE J. Solid-State Circuits, vol.40, no. 4, pp. 970-977, April 2005.[7] H. Darabi, “A Blocker Filtering Technique for Wireless Receivers”, ISSCC Dig. Tech.Papers, pp. 84-85, Feb. 2007.978-1-4244-3457-2/09/$25.00 ©2009 IEEE231DIGEST OF TECHNICAL PAPERS  •ISSCC 2009 / February 10, 2009 / 11:15 AMFigure 12.8.1: Architecture of the interference-robust software-defined radio receiver.Figure 12.8.2: Principle of the 2-stage polyphase harmonic rejection. (α and β areerrors in √2 of the 1st and 2nd stages respectively, but it also works for errors in 1:1 mis-match.)Figure 12.8.3: Dynamic transmission-gate flip-flop (top) and low-noise transconduc-tance amplifier (bottom).Figure 12.8.5: Measured harmonic-rejection (HR) ratio. Figure 12.8.6: Summary of measured key parameters.Figure 12.8.4: Measured gain, NF, in-band IIP2/IIP3 over RF (top); measured signalgain vs. blocker power (bottom).ε===⋅+ ⋅+ ⋅12•  2009 IEEE International Solid-State Circuits Conference 978-1-4244-3457-2/09/$25.00 ©2009 IEEEISSCC 2009 PAPER CONTINUATIONSFigure 12.8.7: Micrograph of the chip fabricated in a 65nm baseline CMOS.

A Software-Defined Radio Receiver Architecture Robust to Out-of-Band Interference

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A Software-Defined Radio Receiver Architecture Robust to Out-of-Band Interference

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