A description is presented of novel transconductor and integrator circuits which can be used in integrated video frequency filters in bipolar technology. The transconductor consists of a parallel connection of a passive nominal transconductance and an active variable transconductance, resulting in good high-frequency performance up to 70 MHz and less than 1% linearity error for input signals up to 2V pp. The integrator incorporates an operation transconductance amplifier circuit which provides a tunable integrator phase. Simulation results for all circuits and for a fifth-order elliptic low-pass filter with a nominal cutoff frequency of 5 MHz are presente

Heij, W.J.A. de

Hoen, K.

Seevinck, Evert

English

de Heij, W.J.A.

University of Twente Research Information

TRANSCONOUCTOR AND INTEGRATOR CIRCUITS FOR INTEGRATED BIPOLAR VIDEO FREQUENCY FILTERS r W.J.A. de Heij E. Seevinck K. Hoen ‘ot &++ I -1 I 7 -ql! /GT - University of Twente, P.O. Box 217, Enschede, the Netherlands. Also with Philips Res. Labs., P.O. Box 80000, Eindhoven, the Netherlands. f ; . I  I I ’ Abstract This paper describes novel transconductor and an integrator circuits which can be used in integrated video frequency filters in bipolar technology. The transconductor consists of a parallel connection of a passive nominal transconductance and an active variable transconductance, resulting in good high frequency performance up to 70 MHz and less than 1% linearity error for input signals up to 2Vpp. In the integrator an operational transconductance amplifier circuit is used which provides tunable integrator phase. Simulation results of all circuits and of a fifth order elliptic lowpass filter with a nominal cutoff frequency of 5 MHz are presented. Introduction Bipolar technology has been used recently to realize continuous-time filters at video frequencies operating from a single 5 V supply [1],[2]. ‘Modern combined bipolar-CMOS processing allows the combination of analog bipolar filters with CMOS digital logic on one chip, thus extending the range of application of bipolar filters. Until now most integrated filters at video frequencies are of the transconductor-C class [l], [3]-[5]. In this class of filters it is difficult to design bipolar circuits having the same signal handling capability as the CMOS circuits. Linearization techniques such as emitter degeneration force the designer to trade off linearity against tunability [l] unless voltage tuned junction capacitors are used [5]. The high transconductance of the bipolar transistor can not be fully used in transconductor-C filters due to its exponential dependence on the base-emitter voltage. In active-RC filters however the high transconduc- tance of the bipolar transistor can be used effectively in the design of amplifiers with high GB-product and low output resistance. In this paper novel transconductor and amplifier circuits are presented which extend the operating range of active-RC filters beyond video frequencies. The transconductor circuit combines both good linearity and tunability. The circuits are balanced, resulting in better PSRR, less crosstalk through the substrate, and cancellation of some parasitic effects and nonlinearities. The authors wish to thank B. Nauta and R.F. Wassenaar for their comments and fruitful discussions. Tunable transconductor design In a balanced activeRC integrator the input branch consists of two matched resistors with a nominal value R, following from filter synthesis. If oxide capacitors are used the resistors have to be made variable to tune the filter poles and zeros to their correct positions against production spread (tm% for a modem process) and temperature variations. A tunable trans- conductor can be realized by connecting a variable transconductance %, realized by an active circuit and having positive or negative values (nominally zero) in parallel with the resistors R, as shown in fig. 1. Fig. 1 : Tunable transconductor principle. The variable transconductance is defined by: (1) io+-io- vi  G ,  = -For the complete transconductor we find: i -i So we have created a tunable transconductance G which is varied around a nominal design value l / R l  since the nominal value of @ is zero. The parallel connection shown in fig. 1 has significant advantages in integrated filter applications where a deviation from the unity gain frequency %=l/RIC has to be corrected by tuning. In a well controlled process <0.2/R, is a realistic value [SI. Therefore 6 will be small, resulting in the following advantages: 114 ISCAS ’89 CH2692-2/89/0000-0114 $1.00 0 1989 IEEE mpartmnt ot ~ecmcxu and mmputer Engmeenng and The coordinated Science Laboratory 1) The high frequency performance is dominated by the resistors R,, instead of the active circuit, resulting in a very small phase shift at video frequencies. 2) The negative effect of non-linearity of the active circuit on the linearity of the complete transconductor is relatively small. 3) The active circuit can operate at low bias currents thus adding little noise to the thermal noise of the resistors R,. The influence of non-idealities of the active circuit is limited by the amount of tuning needed in the integrated filter. Fig. 2 shows a circuit realization of the transconductor principle. The variable transconductance .G, consists of the resistors A, and the npn-transistors Q, to Q,. Using the translinear principle and eq. 2 we can find for the transconduc- tance G: 0.03 i -i (31 G - t+ t -  - 1 I 1 - * 2  1 v ,  7+7,*71; - Fig. 3 shows a simulation of the transconductance G as a function of the input voltage. We have chosen R,=6kn, R2=4W, I,=1!50pAA, and l,+l,=5OpA. Results are shown for +=, +lo%, o%, -10% and -20% tuning with respect to 1 P l .  6 3 0.01 -0.01 -0.03 - ..... - . - t20% +lo% 0% -10% -20% 1 ; ; - The maximum differential input signal amplitude is mainly determined by 1, and R,: For a given value of l3 the amount of tuning depends on 1,4, and R,. For small values of I1-l2 most of the noise produced by Q, Q, and the resistors R, is eliminated within the circuit by the cross-coupled pairs Q,,Q, and Q?Q4. In that case Q,-Q, produce most of the output noise. Therefore the sum current I,+l, should not be made larger than is necessary for the expected values of 1,4, needed for tuning. The circuit has good high-frequency performance. The feed-forward currents through the base-collector capa- citances of Q, to Q4 cancel each other if balanced signals are applied to the circuit. It is possible to obtain a flat phase characteristic almost independent of tuning by including capacitors C, and C,. C, compensates for the phase of R,, while C, compensates for the phase of R, and the active circuit. + ;i ground virtual ground it- f- I .  I I A- - Fig. 2: Circuit realization of complete transconductor. -1.5 -1.0 -0.5 0.0 0.5 1.0 1.5 Vi [VI Fig. 3: Transconductance G versus input voltage. The input voltages where 1% deviation from the trans- conductance at Vi=O is found are 1.02V, 1.13V, l . lOV and 0.95V for +20%, +lo%, -10% and -20% tuning. In case of +lo% tuning the 1% error is caused by a 10.7% error of the variable transconductance %, which shows the reduced effect of nonlinearity of the active circuit. In fig. 4 phase characteristics are shown for the same parameters used in fig. 3 and C,=14.3fF and C2=48fF, using a frequency dependent model for base diffused resistors [7]. The phase characteristics stay veq close to 0' for video frequencies. In practice there will be small deviations due to process tolerances. In high-Q applications they can be corrected by phase tuning, which will discussed in the next section. - ...... - _  - -10% -20% t20% +lo% 0% -0.05  1- 1.C€tO6 ID307  t m  t w  Fig. 4: Phase characteristics for different tuning. It can be concluded that the transconductor principle resutts in good performance with respect to both linea- r i i  and high frequency performance. Its performance has an optimum for the nominal case (%=O). 115 The acthreRC Integrator Fig. 5 shows the concept for the tunable activeRC integrator. The transconductor described in the previous section is used to tune the unity gain frequency U, while variable resistors Ro in the feed- back path are used to tune the phase of the integrator. Instead of an operational voltage amplifier a transcon- ductance amplifier (OTA) having transconductance GA is used. If the OTA is assumed to have input resistance ri and output resistance r, the following transfer function is found: The OTA circuit Fig. 5: Active-RC integrator concept. From eq. 5 we find the following parameters of the integrator: ( 6 )  D.C.-gain: lADc l  = Rl+rl r 1  (l+RIGT) GAr0  GA(l+GTRl) l+GTR1 Un i t y  ga in  frequency: oo = + =7qc-  (7 )  (' G A R 1 ) C  Eqs. 6 and 7 show that o, and A, are both equally tuned by the variable transconductance +. A more detailed analysis shows that this is caused by the current source output of the variable transconductance q. At DC the output current of @ is forced through the resistors R, and the input resistance ri of the OTA resulting in an additional voltage gain of 1 +R,+ in the transfer function from the integrator input vol- tage Vi to the input voltage of the OTA. For sufficient nominal DC-gain and I R&l <0.2 this has little effect. Eq. 5 also indicates that the integrator has a high frequency zero. For Ro=O the zero is located in the right-half-plane. If Ro=l/GA the zero is placed at in- finity. If Ro>l/GA the zero is moved into the left-half plane where it can be used to compensate for the excess phase caused by parasitic poles and zeros of the circuit. For high4 applications Ro can be made variable so that the phase of the integrator can be tuned. Automatic phase tuning circuits have already been reported in literature on high frequency integrated filters [4],[8]. !DO Fig. 6: OTA with feedback path and output buffers. Fig. 6 shows the OTA with the feedback path to the right of the dotted line in fig. 5. The OTA consists of Q1 and Q,, biased by l;=21s. Q, and Q, act as the variable resistors R,. If Io=O the collector current of Q, and Q4 is given by: Then Rel/GA where GA is the transconductance of Q, and Q2 and the zero is placed at infinity. If lo>O la and I, become smaller than I, and I, resulting in Ro>l/GA and the zero is placed in the left-half-plane. The use of diodes Q, and Q4 has two more important advantages besides tunability: 1) If Io is not too large cornpara to Is the diodes also provide large signal tracking of R, with l/GA and the variation of the base-emitter voltage U,, (U,,,) compensates the variation of Ubl (Ub2) thus reducing harmonic distortion. 2) At high frequencies the impedance of the diodes has a zero at almost the same position as the zero in 1/GA (both caused by the base resistance) extending the frequency range where the compensation is effective. The current l,4, through Q, and Q4 is established by a common-mode feedback circuit, which senses the common mode output voltage with a resistive divider with R3=R, driven by Q, and Q,. At DC the output impedance at the emitters of Q, and Q, is high so that output buffers 0, and 0, (not shown in fig. 5) are necessary to maintain sufficient DC-gain since the outputs are loaded by transconductors shown in fig. 2. The total bias current of these transconductors flows also through Q, and Q, adding up to the bias current I, in fig. 6. Fig. 7 shows integrator simulations for -20%. 0% and +20% tuning of the unity gain frequency. For the OTA I,'=I,=5OOpA. R,=6161 and C=S.OpF. In case of 0% tuning a unity gain frequency 0, of 5.OMHz is found, which is O.lMHz lower than the value found from eq. 5 or eq. 7 due to signal loss in the emitter followers Q, and Q,. The DC-gain is 50dB for 0% tuning and it varies by the same amount as w, as expected from eq. 6. The phase characteristic in fig. 7 is obtained with lg=52pA and is independent of tuning, because the dominant pole is independent of 6, see eq. 5. 116 Department ot ELecmcal and Computer Enpeenng and The coordinated Science Laboratory . . . . . . . . . 0% -20% 20 1 4 -88.5 -20 '-90.5 1 E 4 6  1 E 4 7  f [ k l  Fig. 7: Integrator simulations with different tuning. Filter simulations A fifth order elliptic lowpass filter with 5.OMHz cutoff frequency, 1.25dB passband ripple, 50.3dB stopband damping and transition width n /n,= 1.39 was designed and simulated using the circuits Jescribed. The active-RC filter was implemented directly from the signal flow graph of the passive LCR prototype filter. All transconductors were chosen equal for optimum matching, with R,=6kn. First a nominal design was made, assuming the unity gain frequency of the integrators to be o,=l/R,C,. Without tuning the cutoff frequency of this design was 12.6% below specifcation because of neglection of eq. 7 and signal loss in the emitter followers. All feedback capacitors can be scaled down by 12.6% to obtain the specified cutoff frequency and notch frequencies without frequency tuning. This scaled design can be implemented on chip, taking full advantage of the transconductor properties. In fig. 8 and 9 its response is compared with the ideal response. A Q-control current of 95pA is used to obtain the active filter response. There is additional insertion loss in the passband due to the low DC-gain of the integrators and some peaking near the passband edge. The latter is caused by poor output buffering by Q, and Q,. ideal ------ adive 10 I 1 0 -10 1:;; 2 1;:d0 -70 -80 1 .we5 1 .E* 1 . E a 7  f M  Fig. 8: Simulation of scaled design and ideal response. -5.0 d 1 . E 4 5  1 .E* 1.0€+07 f M  Fig. 9: Passband detail from fig. 8. Conclusions In this paper novel transconductor and integrator cir- cuits are presented for bipolar integrated video frequency filters. The tunable transconductor principle is shown to result in good linearity up to 2Vpp input signal and good frequency response up to 70MHz. In the nominal case (GT=O) the performance is even much better.The integrator circuit is based on a simple OTA with capacitive feedback. The right-half-plane zero caused by the finite transconductance of the OTA is moved to the left-half-plane by insertion of a variable resistance in the feedback path. Thus phase tuning is obtained besides frequency tuning. Simulations of a fifth order elliptic lowpass filter at 5.OMHz indicate that- the circuits can be used for filter realization. Futu!e research aims at improvement of the output buffers, the design of alternative transconductor circuits using the described principle, and on-chip tuning. References (11 S. Takagi et. a/.,'A Differential Input/Output Integrator Without PNP Transistors and its Application to Leapfrog Filter Synthesis', Proc. ISCAS 1988, pp. 2855-2858. [2] H. Hagiwara et &.,'A Monolithic Video Frequency Filter Using NIC-Based Gyrators', IEEE J. Solid-state Circ., vol. 23, pp 175-182, Feb. 1988. [3] F. Krummenacher and N. Joehl,'A 4-MHz CMOS Con- tinuous-Time Filter with On-Chip Automatic Tuning', IEEE J. Solid-state Circ., vol. 23, pp 750-758, June 1988. [4] C.S. Park and R. Schaumann,'Design of a 4-MHz Analog Integrated CMOS Transconductance-C Bandpass Filter' IEEE J. Solid-state Circ., vol. 23, pp 987-996, August 1988. [5] K.W. Moulding er. &.,'Gyrator Video Filter IC with Automatic Tuning', IEEE J. Solid-state Circ., vol. [6] 'BIMOS Design Manual', Internal publication of Philips Elcoma, Nijmegen, the Netherlands. [7] 'PHIWAC Processblock Manual', Internal publication of Philips Elcoma, Nijmegen, the Netherlands. [E]  C.F. Chiou and R. Schaumann, 'Design and Performan- ce of a Fully Integrated Bipolar 10.7 MHz Analog Bandpass Filter', IEEE J. Solid-state Circ., vol. SC-21, pp 6-14, Feb. 1986. SC-15, pp. 963-+, Dec. 1980. 117 

Transconductor and integrator circuits for integrated bipolar video frequency filters

NARCIS 

4-MHz CMOS Continuous-Time Filter with On-Chip Automatic Tuning',

Design and Performance of a Fully Integrated Bipolar 10.7 MHz Analog Bandpass Filter',

Gyrator Video Filter IC with Automatic Tuning',

Hagiwara et &.,'A Monolithic Video Frequency Filter Using NIC-Based Gyrators',

Manual', Internal publication of Philips Elcoma,

PHIWAC Processblock Manual', Internal publication of Philips Elcoma,

Schaumann,'Design of a 4-MHz Analog Integrated CMOS Transconductance-C Bandpass Filter'

Takagi et. a/.,'A Differential Input/Output Integrator Without PNP Transistors and its Application to Leapfrog Filter Synthesis',

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Transconductor and integrator circuits for integrated bipolar video frequency filters

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