A novel approach for highly linear automatic gain control (AGC) in small-signal ampli¯ers is presented in this paper. A HEMT based topology was implemented, biasing the transistor in the transition between the saturated and linear operation regions. Gain control with low distortion is achieved by simultaneous adjustments of the gate to source (Vgs) and drain to source (Vds) voltages, along the line where the second derivative of the transconductance (Gm3) has a null. Comparatively to the traditional approach, with the transistor biased in the saturated region, this ampli¯er has better intermodulation behavior and e±ciency, without important reduction in the gain control range

Garcia, José Angel

Malaver, Emigdio

Mediavilla, Angel

Tazon, Antonio

English

AMS Acta - Alm@DL - Università di Bologna

A NOVEL APPROACH FOR HIGHLY LINEAR AUTOMATIC GAIN
CONTROL OF A HEMT SMALL-SIGNAL AMPLIFIER.
Emigdio Malaver ?, Jose´ Angel Garc´ıa ??, Antonio Tazo´n ? ? ?, Angel Mediavilla ? ? ?
? Universidad de Los Andes, Departamento de Electro´nica y Comunicaciones,
Av. Tu´lio Febres Cordero, Me´rida 5101, Venezuela, emalaver@ing.ula.ve
?? TTI Norte S.A, Av Los Castros No 1, Santander 39005, Spain, email: jangel@ieee.org
? ? ? Universidad de Cantabria, Departamento de Ingenieri´a de Comunicaciones,
Av Los Castros, Santander 39005, Spain, tazon@dicom.unican.es
ABSTRACT
A novel approach for highly linear automatic gain control (AGC) in small-signal amplifiers is presented in this
paper. A HEMT based topology was implemented, biasing the transistor in the transition between the saturated
and linear operation regions. Gain control with low distortion is achieved by simultaneous adjustments of
the gate to source (Vgs) and drain to source (Vds) voltages, along the line where the second derivative of the
transconductance (Gm3) has a null. Comparatively to the traditional approach, with the transistor biased in the
saturated region, this amplifier has better intermodulation behavior and efficiency, without important reduction
in the gain control range.
INTRODUCTION
Nowadays, there are many communications systems using digital modulation techniques. These techniques ge-
nerally result in time-varying envelope signals and demand devices, such as amplifiers and mixers, with very high
linearity to avoid unwanted frequency components in the adjacent channels [1]. The intermodulation distortion
is tightly specified by the communication standard being used, therefore should be minimized in transmitter
and receiver circuits. The three traditional methods used to improve the intermodulation performance of RF
amplifiers are the predistortion, feedback and feedforward techniques [2]. The predistortion uses a network with
a transference function equal to the inverse transference function of the distorting amplifier. The feedback uses
negative feedback to compensate the non-linearity, while the feedfoward extracts a fraction of the third order
intermodulation product and recombine it out of fase with the output from the amplifier. All of these techniques
are complex and expensive. Thus tremendous effort are being placed on optimizing the linearity capabilities of
the amplifiers in order to avoid the need of external circuitry or to reduce the requirements to be implement by
the external block.
Output power control is also becoming a must in current and future wireless standards [3]. However, the
classical way to do that in amplifiers and attenuators do not guarantee a high linearity in the whole control
range. Therefore a great interest is appearing on developing highly linear control applications.
PROPOSAL
Normally, an AGC amplifier is designed with the transistor working in the saturated region. The gain control is
achieved by adjusting the gate to source voltage Vgs from pinch-off to the point of maximum transconductance
(Gm1), see Fig. 1.
Gm1 and Gm3 represent the first and third order of Ids derivatives with respect to Vgs for the Taylor-series
expansion given in equation (1).
Ids(Vgs, Vds) = Ids(VGS , VDS) + Gm1 ∗ Vgs + Gds ∗ Vds + Gm2 ∗ V
2
gs + Gmd ∗ Vgs ∗ Vds + Gd2 ∗ V
2
ds+
Gm3 ∗ V 3gs + Gm2d ∗ V
2
gs ∗ Vds + Gmd2 ∗ Vgs ∗ V
2
ds + Gd3 ∗ V
3
ds + · · ·
(1)
The gain control path drives over points of maximum Gm3, being impossible to assure good linearity figures
along the whole range.
The authors in [4] proved the existence of (Vgs, Vds) pairs along the transition from the lineal to saturated region
where Gm3 has a null while Gm1 experiments an important variation. We then propose to taking advantage
of biasing the transistor in this region in order to guaranteeing gain control with low intermodulation levels.
−1 −0.8 −0.6 −0.4 −0.2 0
−150
−100
−50
0
50
100
150
200
Gate to source voltage Vgs(V)
* 
Gm
1(m
S) &
 + G
m3
(mS
/V2
)
GAIN CONTROL PATH 
Figure 1: Gain control path when the transistor is biased in saturation.
CHARACTERIZATION AND EXPERIMENT
Following the procedures described in [4] and [5], Gm3 and the points where Gm3 is null were found for a
typical NE3210s01 HEMT device from NEC, see Fig. 2, Fig. 3 and Fig. 4
−1
−0.8
−0.6
−0.4
−0.2
0
0.2 0.1
0.2
0.3
0.4
0.5
0.6
0.7
0.8
−200
−150
−100
−50
0
50
100
150
Vds (V)
Vgs (V)
Gm
3 
(m
S/V
2 )
Figure 2: Gm3 evolution surface
−1 −0.5 0
0.1
0.2
0.3
0.4
0.5
0.6
0.7
0.8
Vgs (V)
Vd
s (
V)
X 
X 
X 
X 
X 
X 
X 
X 
X 
X 
X 
X 
X 
X 
Figure 3: Gm3 proyected in the Vgs-Vds plane. The marks
correspond to the first and second Gm3 null
0 0.5 1 1.5 2
0
10
20
30
40
50
60
70
Drain to source voltage Vds(V)
Dr
ain
 cu
rre
nt 
Ids
(mA
)
Figure 4: ¦ Gm3 second null evolution along the I/V DC characteristic
Two amplifiers were implemented using the characterized device, one operating in the transition region and the
other biased in saturation. The two-tones experiment was carried out with tones at 1800 MHz and 1801 MHz
as excitation frequencies. The input power levels were selected as high as possible, while assuring operation in
small-signal regime.
−1 −0.9 −0.8 −0.7 −0.6 −0.5 −0.4 −0.3 −0.2
−20
−15
−10
−5
0
5
10
15
Gate to source voltage Vgs(v)
Tr
an
sd
uc
er
 P
ow
er
 G
ain
 (d
B)
Figure 5: Measured Amplifiers Transducer Gain. ¦ Pro-
posal. . Conventional.
−30 −20 −10 0 10 20
−35
−30
−25
−20
−15
−10
−5
0
5
10
15
Transducer Power Gain (dB)
3 
dB
 In
te
rc
ep
t P
oin
t
Figure 6: Measured Amplifiers 3dB Intercept Point. ¦
Proposal. . Conventional
Fig. 5 shows the gain achieved in both amplifiers, being the maximum gain and the control range of the new
AGC approach only 8dB worse than the conventional solution.
The output IP3 evolution is shown in Fig 6. It can be noticed that the amplifier operating in saturation has one
optimum point associated with the Gm3 null defining the pinch-off point. However, for the rest of the control
region the proposed approach shows a better performance.
Due to the relevance of assuring high linearity and also preserving good efficiency [6], the RF output power and
the output IP3 per DC power consumption were computed and shown in Fig 8 and Fig.7.
10−3 10−2 10−1 100 101 102
−55
−50
−45
−40
−35
−30
−25
−20
−15
−10
DC Power Consuption (mW)
RF
 O
ut
pu
t P
ow
er
 (d
Bm
)
Figure 7: RF output power versus DC power consumption.
¦ Proposal. . Conventional with same RF input power. ∗
Conventional with same RF output power.
10−3 10−2 10−1 100 101 102
−35
−30
−25
−20
−15
−10
−5
0
5
10
15
DC Power Consuption
3 
dB
 In
te
rc
ep
t P
oin
t (d
Bm
)
Figure 8: IP3 versus DC power consumption. ¦ Proposal.
. Conventional.
As it can be appreciated, the new AGC amplifier provides a better linearity/efficiency trade-off, constituting a
promising solution for its use on mobile terminals.
In this work one goal was to measure the evolution of 1 dB compression point along the gain control path, since
it was expected to have a poor behavior when the transistor is biased in the proposed operating region. In Fig.
9, it can be appreciated that the proposal has a similar evolution than the conventional AGC amplifier with
just 5 dB of maximum deviation respect to the conventional 1 dB evolution curve.
Finally, it would be interesting to evaluate the capabilities of this technique when exciting with digital or
multicarrier signals. Using the results in [7], it is possible to extend the two-tones linearity figures to estimate
the adjacent channel power ratio (ACPR).
ACPR =
ω2∫
ω1
C21
No
2 dω
18
ωh+Bω∫
ωh
C23
(
No
2
) (
ω2
2 − (Bω + ωh)ω +
(Bω+ωh)
2
2
)
dω
(2)
−20 −15 −10 −5 0 5 10 15
−35
−30
−25
−20
−15
−10
−5
0
5
Transducer Power Gain dB
1 
dB
 C
om
pr
es
sio
n 
Po
int
Figure 9: 1 dB compression point evolution. ¦ Proposal.
. Conventional.
−20 −15 −10 −5 0 5 10 15
20
25
30
35
40
45
50
55
Transducer Power Gain (dB)
AC
PR
 (d
B)
Figure 10: Output ACPR relation. ¦ Proposal. . Conven-
tional.
The equation (2) was applied to obtain the ACPR evolution of the new AGC approach for a W-CDMA signal
to be employed in the future UMTS wireless standard, see Fig. 10. As it can be seen, the new AGC amplifier
would satisfy an ACPR requirement of 43 dB in the whole gain control range, while the amplifier operated in
saturation only satisfies a 23 dB request.
A quantitative analysis of the novel approach has shown that these results could be extended to large-signal
regime and this will be the objective of a future work.
CONCLUSIONS
A new AGC amplifier technique has benn proposed. The new approach provides good intermodulation perfor-
mance and low power consumption. It may constitute a promising solution for the availability of efficient and
highly linear output-power control in mobile terminals.
ACKNOWLEDGEMENT
This work was supported in part by the European Union under projects ALFA and TIC-1FD1997-1066-C02-01.
J.A. Garc´ıa has participated supported by the project IST-1999-12070 TRUST. The author wants to thank to
all institutions that participate at this project.
REFERENCE
[1] L.E. Larson, RF and Microwave Circuit Design for Wireless Communications, 1996 Artech House, Norwood
Massachusetts.
[2] C. S. Aitchison. “The Current Status of RF and Microwave Amplifier Intermodulation Performance,IEEE
Radio Frequency Integrated Circuits Symposium 2000, pp. 113-116.
[3] ETSI TS 125101 and 125102 (2000-2001), Universal Mobile Telecommunications Systems (UMTS), UE Radio
Transmission and Receptions (FDD and TDD).
[4] J. A. Garc´ıa et. al. “Resistive FET Mixer Conversion Loss and IMD Optimization by Selective Drain Bias”,
IEEE Trans. Microwave Theory and Techniques, vol. 47, No. 12, pp. 2382-2392, Dec. 1999.
[5] J. Pedro and J. Pe´rez. “Accurate simulation of GaAs MESFET’s intermodulation distortion using a new
drain-to-source current model”, IEEE Trans. Microwave Theory and Techniques, vol. 42, pp. 25-33, Jan. 1994.
[6] D. B. Kenington, High Linearity RF Amplifier Design, 2000 Artech House, Norwood Massachusetts.
[7] J.C. Pedro and N.B. Carvalho. “On the Use of Multitone Techniques for Assessing RF Components’ Inter-
modulation Distortion”, IEEE Trans. Microwave Theory and Techniques, vol. 47, No. 12, pp. 2393-2402, Dec.
1999.


A novel approach for highly linear automatic gain control of a hemt small-signal amplifier

AMS Acta

A NOVEL APPROACH FOR HIGHLY LINEAR AUTOMATIC GAINCONTROL OF A HEMT SMALL-SIGNAL AMPLIFIER.Emigdio Malaver ?, Jose´ Angel Garc´ıa ??, Antonio Tazo´n ? ? ?, Angel Mediavilla ? ? ?? Universidad de Los Andes, Departamento de Electro´nica y Comunicaciones,Av. Tu´lio Febres Cordero, Me´rida 5101, Venezuela, emalaver@ing.ula.ve?? TTI Norte S.A, Av Los Castros No 1, Santander 39005, Spain, email: jangel@ieee.org? ? ? Universidad de Cantabria, Departamento de Ingenieri´a de Comunicaciones,Av Los Castros, Santander 39005, Spain, tazon@dicom.unican.esABSTRACTA novel approach for highly linear automatic gain control (AGC) in small-signal amplifiers is presented in thispaper. A HEMT based topology was implemented, biasing the transistor in the transition between the saturatedand linear operation regions. Gain control with low distortion is achieved by simultaneous adjustments ofthe gate to source (Vgs) and drain to source (Vds) voltages, along the line where the second derivative of thetransconductance (Gm3) has a null. Comparatively to the traditional approach, with the transistor biased in thesaturated region, this amplifier has better intermodulation behavior and efficiency, without important reductionin the gain control range.INTRODUCTIONNowadays, there are many communications systems using digital modulation techniques. These techniques ge-nerally result in time-varying envelope signals and demand devices, such as amplifiers and mixers, with very highlinearity to avoid unwanted frequency components in the adjacent channels [1]. The intermodulation distortionis tightly specified by the communication standard being used, therefore should be minimized in transmitterand receiver circuits. The three traditional methods used to improve the intermodulation performance of RFamplifiers are the predistortion, feedback and feedforward techniques [2]. The predistortion uses a network witha transference function equal to the inverse transference function of the distorting amplifier. The feedback usesnegative feedback to compensate the non-linearity, while the feedfoward extracts a fraction of the third orderintermodulation product and recombine it out of fase with the output from the amplifier. All of these techniquesare complex and expensive. Thus tremendous effort are being placed on optimizing the linearity capabilities ofthe amplifiers in order to avoid the need of external circuitry or to reduce the requirements to be implement bythe external block.Output power control is also becoming a must in current and future wireless standards [3]. However, theclassical way to do that in amplifiers and attenuators do not guarantee a high linearity in the whole controlrange. Therefore a great interest is appearing on developing highly linear control applications.PROPOSALNormally, an AGC amplifier is designed with the transistor working in the saturated region. The gain control isachieved by adjusting the gate to source voltage Vgs from pinch-off to the point of maximum transconductance(Gm1), see Fig. 1.Gm1 and Gm3 represent the first and third order of Ids derivatives with respect to Vgs for the Taylor-seriesexpansion given in equation (1).Ids(Vgs, Vds) = Ids(VGS , VDS) + Gm1 ∗ Vgs + Gds ∗ Vds + Gm2 ∗ V2gs + Gmd ∗ Vgs ∗ Vds + Gd2 ∗ V2ds+Gm3 ∗ V 3gs + Gm2d ∗ V2gs ∗ Vds + Gmd2 ∗ Vgs ∗ V2ds + Gd3 ∗ V3ds + · · ·(1)The gain control path drives over points of maximum Gm3, being impossible to assure good linearity figuresalong the whole range.The authors in [4] proved the existence of (Vgs, Vds) pairs along the transition from the lineal to saturated regionwhere Gm3 has a null while Gm1 experiments an important variation. We then propose to taking advantageof biasing the transistor in this region in order to guaranteeing gain control with low intermodulation levels.−1 −0.8 −0.6 −0.4 −0.2 0−150−100−50050100150200Gate to source voltage Vgs(V)* Gm1(mS) & + Gm3(mS/V2)GAIN CONTROL PATH Figure 1: Gain control path when the transistor is biased in saturation.CHARACTERIZATION AND EXPERIMENTFollowing the procedures described in [4] and [5], Gm3 and the points where Gm3 is null were found for atypical NE3210s01 HEMT device from NEC, see Fig. 2, Fig. 3 and Fig. 4−1−0.8−0.6−0.4−0.200.2 0.10.20.30.40.50.60.70.8−200−150−100−50050100150Vds (V)Vgs (V)Gm3 (mS/V2 )Figure 2: Gm3 evolution surface−1 −0.5 00.10.20.30.40.50.60.70.8Vgs (V)Vds (V)X X X X X X X X X X X X X X Figure 3: Gm3 proyected in the Vgs-Vds plane. The markscorrespond to the first and second Gm3 null0 0.5 1 1.5 2010203040506070Drain to source voltage Vds(V)Drain current Ids(mA)Figure 4: ¦ Gm3 second null evolution along the I/V DC characteristicTwo amplifiers were implemented using the characterized device, one operating in the transition region and theother biased in saturation. The two-tones experiment was carried out with tones at 1800 MHz and 1801 MHzas excitation frequencies. The input power levels were selected as high as possible, while assuring operation insmall-signal regime.−1 −0.9 −0.8 −0.7 −0.6 −0.5 −0.4 −0.3 −0.2−20−15−10−5051015Gate to source voltage Vgs(v)Transducer Power Gain (dB)Figure 5: Measured Amplifiers Transducer Gain. ¦ Pro-posal. . Conventional.−30 −20 −10 0 10 20−35−30−25−20−15−10−5051015Transducer Power Gain (dB)3 dB Intercept PointFigure 6: Measured Amplifiers 3dB Intercept Point. ¦Proposal. . ConventionalFig. 5 shows the gain achieved in both amplifiers, being the maximum gain and the control range of the newAGC approach only 8dB worse than the conventional solution.The output IP3 evolution is shown in Fig 6. It can be noticed that the amplifier operating in saturation has oneoptimum point associated with the Gm3 null defining the pinch-off point. However, for the rest of the controlregion the proposed approach shows a better performance.Due to the relevance of assuring high linearity and also preserving good efficiency [6], the RF output power andthe output IP3 per DC power consumption were computed and shown in Fig 8 and Fig.7.10−3 10−2 10−1 100 101 102−55−50−45−40−35−30−25−20−15−10DC Power Consuption (mW)RF Output Power (dBm)Figure 7: RF output power versus DC power consumption.¦ Proposal. . Conventional with same RF input power. ∗Conventional with same RF output power.10−3 10−2 10−1 100 101 102−35−30−25−20−15−10−5051015DC Power Consuption3 dB Intercept Point (dBm)Figure 8: IP3 versus DC power consumption. ¦ Proposal.. Conventional.As it can be appreciated, the new AGC amplifier provides a better linearity/efficiency trade-off, constituting apromising solution for its use on mobile terminals.In this work one goal was to measure the evolution of 1 dB compression point along the gain control path, sinceit was expected to have a poor behavior when the transistor is biased in the proposed operating region. In Fig.9, it can be appreciated that the proposal has a similar evolution than the conventional AGC amplifier withjust 5 dB of maximum deviation respect to the conventional 1 dB evolution curve.Finally, it would be interesting to evaluate the capabilities of this technique when exciting with digital ormulticarrier signals. Using the results in [7], it is possible to extend the two-tones linearity figures to estimatethe adjacent channel power ratio (ACPR).ACPR =ω2∫ω1C21No2 dω18ωh+Bω∫ωhC23(No2) (ω22 − (Bω + ωh)ω +(Bω+ωh)22)dω(2)−20 −15 −10 −5 0 5 10 15−35−30−25−20−15−10−505Transducer Power Gain dB1 dB Compression PointFigure 9: 1 dB compression point evolution. ¦ Proposal.. Conventional.−20 −15 −10 −5 0 5 10 152025303540455055Transducer Power Gain (dB)ACPR (dB)Figure 10: Output ACPR relation. ¦ Proposal. . Conven-tional.The equation (2) was applied to obtain the ACPR evolution of the new AGC approach for a W-CDMA signalto be employed in the future UMTS wireless standard, see Fig. 10. As it can be seen, the new AGC amplifierwould satisfy an ACPR requirement of 43 dB in the whole gain control range, while the amplifier operated insaturation only satisfies a 23 dB request.A quantitative analysis of the novel approach has shown that these results could be extended to large-signalregime and this will be the objective of a future work.CONCLUSIONSA new AGC amplifier technique has benn proposed. The new approach provides good intermodulation perfor-mance and low power consumption. It may constitute a promising solution for the availability of efficient andhighly linear output-power control in mobile terminals.ACKNOWLEDGEMENTThis work was supported in part by the European Union under projects ALFA and TIC-1FD1997-1066-C02-01.J.A. Garc´ıa has participated supported by the project IST-1999-12070 TRUST. The author wants to thank toall institutions that participate at this project.REFERENCE[1] L.E. Larson, RF and Microwave Circuit Design for Wireless Communications, 1996 Artech House, NorwoodMassachusetts.[2] C. S. Aitchison. “The Current Status of RF and Microwave Amplifier Intermodulation Performance,IEEERadio Frequency Integrated Circuits Symposium 2000, pp. 113-116.[3] ETSI TS 125101 and 125102 (2000-2001), Universal Mobile Telecommunications Systems (UMTS), UE RadioTransmission and Receptions (FDD and TDD).[4] J. A. Garc´ıa et. al. “Resistive FET Mixer Conversion Loss and IMD Optimization by Selective Drain Bias”,IEEE Trans. Microwave Theory and Techniques, vol. 47, No. 12, pp. 2382-2392, Dec. 1999.[5] J. Pedro and J. Pe´rez. “Accurate simulation of GaAs MESFET’s intermodulation distortion using a newdrain-to-source current model”, IEEE Trans. Microwave Theory and Techniques, vol. 42, pp. 25-33, Jan. 1994.[6] D. B. Kenington, High Linearity RF Amplifier Design, 2000 Artech House, Norwood Massachusetts.[7] J.C. Pedro and N.B. Carvalho. “On the Use of Multitone Techniques for Assessing RF Components’ Inter-modulation Distortion”, IEEE Trans. Microwave Theory and Techniques, vol. 47, No. 12, pp. 2393-2402, Dec.1999.

A novel approach for highly linear automatic gain control of a hemt small-signal amplifier

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