Digital simulation of the data link for the Saturn-Uranus spacecraft design is summarized. The data link is a 40 watt, 400 MegaHertz, 44 bit-a-second link, a power starved link, and uses convolution coding. Aspects of the link of particular interest are atmospheric scintillation and the Doppler to data rate ratio. The major characteristics of the scintillation are modeled in terms of tapped delay lines. A candidate system design and receiver design is developed. A computer flow diagram depicting a routine for the error rate and one for acquisition is presented. Various coding algorithms are analyzed since the convolution code is sensitive to the distribution of error and the actual pattern of the error

Hinrichs, C.

English

NASA Technical Reports Server

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NT[, 26396
DIGITAL RECEIVER SIMULATION
Mr. Carl Hinrichs - McDonnell-Douglas Corporation
MR. HINRICHS: The simulation is summarized on Figure 7-26
and was for the Saturn-Uranus design that you have heard so much
about in the last day and a half. This design is 40 watt, 400
MegaHertz, 44 bit-a-second link and, as has been pointed out, is a
power starved link and uses convolution coding. As far as the
!_imulation itself goes, parameters such as the power level, the
bit rate, and the range are relatively insignificant. These are
taken into the simulated signal-energy-to-noise-density ratios.
The center frequency is, in the simulation, relatively unimpor-
tant because the simulation is entirely in complex amplitude so
that the center frequency is just a normalization.
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As was pointed out, we were interested in encoding this link
and this is one of the reasons that the simulation became par-
ticularly attractive. For convolutional codes we do not have
to concern ourselves with some typical simulation problems such
as very low symbol error rates. We will be looking primarily
for symbol error rates that are around .05. And if we get
down to .01 or .001, this is very solid for the code. This
makes simulation quite attractive.
Fine, it is attractive but why simulate this particular
link? As we have heard from the previous speakers, this link
has several unique aspects. First of all, atmospheric scintil-
lation. We are in an atmosphere here today, we transmit radio
waves back and forth, why don't we have that problem? Well,
primarily because we are not at a ten or thirty bar level.
We are only in a one bar level here. If the pressure were
higher, we would start seeing scintillation problems.
Secondly, the center frequency certainly enters into this,
our Doppler to data rate ratio is very high. What I mean by
this is that relative to the bandwidth of the data, the
VII-43
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TYPICAL lINK
•40 WATTS
• 400MHz
• 44BPS
• I05km RANGE
•CONVOLUTIONAL CODE
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• ATMOSP'-"----_cHERICS {NTILLATION
TYPICAL
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Figure 7-26
VII-44
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frequency uncertainties due the Doppler are quite wide. So
we have a unique aspect in this sense. Because of the unique-
ness of the link, the unique problems and because we are only
looking for fairly high symbol error rates as opposed to an
uncoded system, simulation appears to be a good technique to
determine the applicability of candidate designs.
Now in the next chart, (Figure 7-27), I would like to review
a little bit about atmospheric scintillation. Sometimes we tend
to say that these problems are non-analytic. Certainly in the
past, there have been a n r of articles, at least that I am
familiar with, that deal with fading. In the bulk of the fad-
ing articles, the amplitude is generally considered Raleigh or
Ricean and the phase is assumed to be uniform. In atmospheric
scintillation, neither of these is necessarily the case.
Atmospheric scintillation arises when one has a blob, as
it is called in the literature, of atmosphere with an index
of refraction slightly different from the remaining atmosphere.
This blob may have been generated in a number of ways but
generally, it is some form of thermal instability that creates
it. The blob is unstable and breaks into smaller blobs. The
smaller blobs continually break until the Reynolds number is
finally sufficient and it can dissipate. So there is a range
of inhomogeneities in the index of refraction.
As an electromagnetic wave passes through this range of
inhomogeneities, the larger inhomogeneities tend to affect
the phase of the signal and the smaller inhomogeneities tend
to affect the amplitude of the signal. Thus, we see the
amplitude in the phase characteristics of the siqnal are in-
dependent.
As Mr. Grant pointed out, for this simulation we have modeled
the scintillation amplitude as some value A, with a 4/3rds foll off
VII-45
• .c ¸
SCINTILLATION FILTERING
AMPLITUDE: A (I/(I+ jf/fA))4,'3
"A = 0; °A " 0.23; fA" 2.
PHASE: B (1/( 1 ,j f/fB)) 4i3
A
-_' _B = 0; °B = 0.47; fB = 0.2
(i•A)EXP j(B, B)
B
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Figure 7-27
VII-46
ORIGINAL PAGE IS
OF POOR QUALITY
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at a corner of two Hertz. This amplitude is modeled, in this
case, as having a zero mean and a root variance of .23.
The phase, the other independent variable, again, has a
4/3rds filter roll off. Four-thirds is basically from the
Russian Tatarski. The phase has, again, a zero mean and a
root variance of .47 radians and rolls off at a much lower
corner, 2/10ths of a Hertz.
Typically, in digital simulations, we like to use Z trans-
forms but as one can fairly readily show, when one has a non-
integer number of poles, the Z transform series doesn't collapse
into a closed form. So we spent a fair amount of effort in
modeling the exact characteristics of the scintillation in
terms of tapped delay lines. We took independent Gaussian
numbers and ran them through the delay lines to form the ampli-
tude and the phase. For this simulation we modeled the ampli-
tude as simply unity plus the Gaussian number. A better simu-
lation might utilize a log normal.
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Given the problem, we need a candidate design. In the
first portion of the Saturn-Uranus study, TRW supported Mc-
Donnell Douglas in defining the hardware impacts of various
candidate system designs. In the latter portion of the study,
they took the resultant system design and performed a detailed
receiver design. That receiver design is shown on Figure 7-28.
In the receiver, the lower loop is the frequency tracking
loop. This loop tracks the tones of the transmitter. It is
a continuous phase, FSK transmitter. The upper loop is the
automatic gain control loop which serves to hold the voltage
for the AFC loop at a constant value. The automatic gain
control loop provides a signal strength indication from the
coherent amplitude detector. If it is not locked to the sig-
nal, it can initiate the sweep circuitry.
VII-47
CANDIDATE DESIGN (TRW)
INPUT
OUTPUI:NO. I
HARD SOFT
DOPPLER
I
BASELINE
CORRECTION
..,, .
Figure 7-28
v'rz-48
SOFT HARD
OUTPUT NO. 2
fThe signalfeed back from the tracking loop filter indi-
cates when it may have gone beyond the specified sweep or
anticipated Doppler range. It will then reverse the sweep
direction.
The bit synchronizer, is a relatively straightforward
in-phase, quadrature phase, bit synchronizer. It has a base-
line correction circuit to correct for "drifts," i.e., long
successive strings of either plus ones or minus ones. ')
Fairly early in the simulation efforts, it appeared that it
would be easy in the simulation, since the bulk of the work in
a digital simulation is in the receiver (relatively little of
the work in terms of computing time takes place in the bit
synchronizer) to look at two different type of detectors: a
sampled filter detector and the in-phase integrator (as a de-
tector). For both of these detectors, we look at both a hard
decision; (that is either a plus or minus one) or soft decision
(the relative level of confidence of a level). This is the
candidate design that we have investigated.
• . z,.. .,
This chart (Figure 7-29) represents an abbreviated com-
puter flow diagram. We actually generated two routines, one
for the error rate and one for the acquisition. Unfortunately,
we never got a set of curves of the acquisition probabilities
as every time we tried to acquire, we did. Perhaps if we go
lower in E/No (we only went down to 7 db) we could start to
define the curve. Above 7 db, the receiver acquired every time.
Basically, in the computer flow after initializing the
problem, we may or may not step the scintillation. We are
taking approximately 40 samples per bit in the simulation.
Because the scintillations are only two Hertz and 2/10ths of
a Hertz compared to 88 symbols per second, it was not necessary
to step the scintillation lines every time that we stepped a
sample for a bit. Thus we saved some time here. The simulation
data is a 63 bit PN sequence.
VII-49
ERROR ROUTINE FLOW CHART
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START
ISTEPSCINTILLATION?}Y
! NEWS,T:]Y
[RECEIVERLOOPS1
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GENERATE NEWBIT]
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BIT SYNCLOOP
I ERROR COUNTER
[ END RUN?
Figure 7-2,9
VII-50
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One of the interesting things in the simulation was that
we simulated to the lowest feasible component in the receiver.
Each filter in the receiver, the band-pass filters, the track-
ing filters, were individual Z transforms, the gain constant
of the VCO's were independently variable; each multiplier oc-
curred (the front end of the receiver) as a complex amplitude
multiplication.
We ran some interesting parametrics, Figure 7-30. We looked
at varying the modulation index and, the old 7/10ths modulation
index still holds good. The initial design was for a 1,000 Hertz
IF. It looks like slightly larger IF's might be more advantag-
eous. In the future we will be looking at 1,500 or 2,000 Hertz.
The IF has to be wide enough so that there won't be any phase
distortion in the receiver; but if it is very wide, it is not
necessary. One curious thing that we discovered was that the
dynamic range of the automatic gain control could be increased
somewhat. By this, the AGC tries to keep the voltage level to
the AFC loop constant. What one normally does in a design is
when the signal hits the threshold, the gain stops. If the gain
were a bit greater, the performance improves.
Finally, looking at the two different types of detectors,
in all of the runs that we made, the integrato r detector - that
is, the in-phase integrator in the bit synchronizer, out per-
formed the sample filter detector. It appears that the inte-
grator detector is the best design.
One of the things that we always like to look at is error
rate. The No scintillation and scintillation data shown here
are compared to the original specification which was an FSC BT=2
receiver. The candidate design is performing well within that
bound.
In conclusion (Figure 7-31), in terms of the mean error rate,
this is an acceptable design. However, when considering convolution
VII-51
PARAMETRICS
I0-1
3
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10-2
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3
Pa
lO-2
WITHOUT SCINTILLATION
WITH SCINTILLATION
E/No = 9.4 dB
- /i/ i0-i_
7
Pe
lo-Z
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0.3 0.5 0.l 1.0
MODULATION INDEX
\
\
\
3 I { [ I
500 100_)1500 2000
INTERMEDIATEFREQUENCY
BANDWIDTH - HZ
lO-I -
,K?
DYNR IIt_..%E -
N0MIN
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DYNAMIC RANGE - (:IB
Pe
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,,,,..,.,,. -.-" ""
- SA_E__
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88 144 2OO
SAMPLE FILTERCORNER - Hz
Figure 7-30
ORIGINAL PAGE L_
OF POOR QUALITY
_;;:,_ V I I - 5 2
CONCLUSIONS
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Pe
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10-2
10-3
- _F" STPyEpClcFAI:AcT01DON'LIFMSI_, :0T5=2
-NOSC,NT,'LAT,ON 
-N O SCINTILLAT_
[ I I I 1
6 1 8 9 10 It
E/N0 , dB
CONCLUSIONS -
• CANDIDATE ACCEPTABLE FROM
THE MEAN ERROR RATE VIEWPOINT
• COMPLETE VERIFICATIONAWAITS
ARC CODING ANALYSIS
RECOMMENDATIONS -
• INTERMEDIATE FREQUENCY BANDWIDTH = 1500Hz
•AUTOMATIC GAIN CONTROL DYNAMIC RANGE_ 3 dB
BELOW " MINIMUMSIGNAL"
•USAGE OF INTEGRATION RATHER THAN SAMPLING
FILTER DETECTOR
• FURTHER EXPLORATION OF SCINTILLATION
MODEL EFFECTS
Figure 7-31
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VII-53
ORIGINAL PAGE IS
OF POOR QUA//T_
codes, the mean error rate is only one of the criteria. The
code is sensitive to not only the distribution of errors but
the actual pattern of the errors. At the conclusion of this
study, we cut magnetic tapes for ARC to analyze for different
coding algorithms. The tape records the different detector
performance via soft decisions.
We recommend an IF frequency a little bit greater than
1,000 Hertz; an AGC something below the usual definable
minimum signal, and integration detector rather than a sample
filter detector, and now that we have the tools available to us,
investigate a variety of scintillation models.
Thank you.
MR. GRANT: Our next speaker is Dr. James Modestino,
Associate Professor in the Systems Engineering Division at
Rensselaer Polytechnic Institute. Dr. Modestino will report
on convolutional code performance in fading channels.
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Digital receiver simulation

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