N A T I O N A L A E R O N A U T I C S A N D S P A C E A D M I N I S T R A T I O N April 15, 1971 https://ntrs.nasa.gov/search.jsp?R=19710013725 2020-03-11T21:06:24+00:00Z N A T I O N A L A E R O N A U T I C S A N D S P A C E A D M I N I S T R A T I O N Y C A L I F O R N I A I N S T I T U T E OF T E C H N O L O G Y P A S A D E N A , C A L I F O R N I A April 15, 1971 Prepared Under Contract No. NAS 7- 100 National Aeronautics and Space Administration This report series presents progress on DSN supporting research and technology, advanced development and engineering, and implementation, and DSN operations which pertain to mission-independent or multiple-mission development as well as to support of flight projects. Each issue presents material in some, but not all, of the following categories in the order indicated. Description of the DSN Mission Support Interplanetary Flight Projects Planetary Flight Projects Manned Space Flight Project Advanced Flight Projects Advanced Engineering Tracking and Navigational Accuracy Analysis Communications Systems Research Communications Elements Research Supporting Research and Technology Development and Implementation Space Flight Operations Facility Development Ground Communications Facility Development Deep Space Instrumentation Facility Development DSN Project and System Development Operations and Facilities DSN Operations Space Flight Operations Facility Operations Ground Communications Facility Operations Deep Space Instrumentation Facility Operations Facility Engineering In each issue, the part entitled “Description of the D S N describes the functions and facilities of the DSN and may report the current configuration of one of the six DSN systems (tracking, telemetry, command, monitoring, simulation, and opera- tions control). The work described in this report series is either performed or managed by the Tracking and Data Acquisition organization of JPL for NASA. ... III acilities . . . . . . . . . . . . . 1 N. A, Renzetti perations Control System . . . . . . . . . . . . 4 J. E. Maclay I S ort . . . . . . . . . . . . . . 6 p o r t . . . . . . . . . . . . . . . 18 A. J. Siegmeth P. S. Goodwin ission Support . . . . . . . . . . . . . . 28 D. J. Mudgway port . . . . . . . . . . . . . . 33 R. B. Hartley SiS . . . . 42 0. H. von Roos er. . . . . . . . . . . . . . . 49 W. Lushbaugh e r s . . . . . . . . . . . . . 56 G. lutes nce . . . . . . . . . 5 9 R. J . McEliece etrie 8 9 . . . . . . . . . . . . 6 2 R. J . McEliece and J . E. Savage o d e ding . . . . . . . . 65 1. S. Reed aser Frequency Sta . . . . 72 A. Sward orithm . . . . . . . . 75 U. Timor icrowave Transmission Through lates . . . 80 T . Y . Otoshi esign of Hydrogen uning Servo . . . . . . . . 86 C. Finnie alibration Techniques: System Noise Temperature Calibrations . . . . . . . . . . . . 89 M. S. Reid nergies 04 a Cathode-Ray Tube System . . . . . . . . 92 J. J . Volkoff Mark lllA Simulation Television System . . . . . . . . . . . . . . . . 100 C. F. Leahey enter Interactive Alphanumeric eanalysis Program for Antenna embersize Changes . . . . . 108 R. levy An~enna Structures: Evaluation of B. Marcus and M. S. Katow istortions . . . . . . . . . . . . . . . 1 13 eve s ontrol . . . . . . . . . . . . . . . 122 R. A. Paine ~agnos~ics for the S . . . . . 125 R. A. Wells vi OldSfQ icrowave ais for 50-kbif ransmission . . . . . . . R. G. Hanselman eve ata System . . R. 5 . Crow . . . . . . . . . . . 1. Butcher jests eporting Subsystem . . . . . . . D. G. Tustin ptimal Frame Sync ronization . . . . . . . . . W. Kizner s SN Traceability an eporting Program . . . . . . . J. A. Miccio s ssions . . . . . . . F. E. Bond, Jr. S e n t . . . . . A. Bryan and G. Osborn rocesse J . M . Urech verse s . . . . . . F. D. Mclaughlin . . . . . . . . . . . . . . . . . . . . . 129 . 133 . 136 . 140 . 141 . 145 . 148 . 165 . 169 . 177 vii N. A. Renzetti Mission Support Office The objectives, functions, and organization of the Deep Space Network are sum- marized. The Deep Space Instrumentation Facility, the Ground Communications Facility, and the Space Flight Operations Facility are described. The Deep Space Network (DSN), established by the NASA Office of Tracking and Data Acquisition under the system management and technical direction of JPL, is designed for two-way communications with unmanned spacecraft traveling approximately 16,000 km (10,000 mi) from earth to planetary distances. It supports, or has sup- ported, the following NASA deep space exploration proj- ects: Ranger, Surveyor, Mariner Venus 1962, Mariner Mars 1964, Mariner Venus 67, Mariner Mars 1969, Mariner Mars 1971 (JPL); Lunar Orbiter and Viking (Langley Re- search Center); Pioneer (Ames Research Center); Helios (West Germany); and Apollo (Manned Spacecraft Cen- ter), to supplement the Manned Space Flight Network (MSFN). The DSN is distinct from other NASA networks such as the MSFN, which has primary responsibility for track- ing the manned spacecraft of the Apollo Project, and the Space Tracking and Data Acquisition Network (STABAN), which tracks earth-orbiting scientific and communications satellites. With no future unmanned lunar spacecraft presently planned, the primary objective of the DSN is to continue its support of planetary and interplanetary flight projects. To support fight projects, the DSN simultaneously per- forms advanced engineering on components and systems, integrates proven equipment and methods into the net- work,l and provides direct support of each project through that project’s Tracking and Data System. This manage- ment element and the project’s Mission Operations per- sonnel are responsible for the design and operation of the data, software, and operations systems required for the conduct of fight operations. The organization and procedures necessary to carry out these activities are described in Ref. 1. 1When a new piece of equipment or new method has been accepted for integration into the network, it is classed as Goldstone duplicate standard (GSDS), thus standardizing the design and operation of identical items throughout the network. By tracking the spacecraft, the DSN is involved in the (1) Radio Metric: generate angles, one- and two-way doppler, and range. (2) Tekmety: receive, record, and retransmit engi- neering and scientific data. (3) Command: send coded signals to the spacecraft to activate equipment to initiate spacecraft functions. following data types: The DSN operation is characterized by six DSN sys- tems: (1) tracking, (2) telemetry, (3) command, (4) moni- toring, (5) simulation, and (6) operations control. The DSN can be characterized as being comprised of three facilities : the Deep Space Instrumentation Facility (DSIF), the Ground Communications Facility (GCF), and the Space Flight Operations Facility (SFOF). rac eilifies A world-wide set of deep space stations (DSSs) with large antennas, low-noise phase-lock receiving systems, and high-power transmitters provide radio communica- tions with spacecraft. The DSSs and the deep space com- SCC Goldstone - Tidbinbilla - Madrid municatioms complexes (BSCCs) they comprise are given in Table 1. Radio contact with a spacecraft usually begins when the spacecraft is on the launch vehicle at Cape and it is maintained throughout the mission. part of the trajectory is covered by selected network sta- ) and Mor- mally, two-way communications are established between the spacecraft and the DSN within 30 min after the space- craft has been injected into lunar, planetary, or interplan- etary flight. A compatibility test station at Cape Kennedy (discussed later) monitors the spacecraft continuously dur- ing the launch phase until it passes over the local horizon. The deep space phase begins with acquisition by either DSS 51, 41, or 42. These and the remaining DSSs given in Table 1 provide radio communications to the end of the flight. the Air Force Eastern Test Range FN of the Goddard Space Flight To enable continuous radio contact with spacecraft, the DSSs are located approximately 120 deg apart in longi- tude; thus, a spacecraft in deep space flight is always T h e 9-m (3043) diam antenna station established by the DSN on Ascension Island during 1965 to act in conjunction with the MSFN orbital support 9-m (30-ft) diam antenna station was transferred to the MSFN in July 1968. location I DSS California Australia Australia South Africa Spain Pioneer Echo (Venus)* Mars Woomerab Weemala (formerly Tidbinbilla)b (formerly Booroomba) Johannesburgb Robledob Cebrerasb Robledo BaIIimab DSS serial designation 11 12 13 14 41 42 43 51 61 62 63 Antenna Diameter, rn (ft) 26 (85) 26 (85) 26 (85) 64 (210) 26 (85) 26 (85) 64 (210) 26 (85) 26 (85) 26 (85) 64(210) Type of maunting Polar Polar Az-El Az-El Polar Polar Az-a Polar Polar Polar Ax-El Year of init ial operation 1958 1962 1962 1966 1960 1965 Under construction 1961 1965 1967 Under conrtruc~ion *A research-and-development facility used to demonstrate the feasibility of new equipment and methods to be integrated into the operational network. Besides the 26-m (85-ft) diam ax-el-mounted antenna, DSS 13 has a 9-m (30-ft) diam az-el-mounted antenna that i s used for testing the design of new equipment and support of ground-based radio science. bNormally staffed and operated by government agencies of the respective countries (except for a temporary staff of the Madrid DSCC), with some asrirt- ance of US. support personnel. within the field-of-view of at least one DSS, and for sev- eral hours each day may be seen by two DSSs. Further- more, since most spacecraft on deep space missions travel within 30 deg of the equatorial plane, the DSSs are located within latitudes of 45 deg north or south of the equator. All DSSs operate at S-band frequencies: 2110-2120 MHz for earth-to-spacecraft transmission and 2290-2300 MHz for spacecraft-to-earth transmission. To provide sufficient tracking capability to enable use- ful data returns from around the planets and from the edge of the solar system, a 64-m (210-ft) diam antenna network will be required. Two additional 64-m (21043) diam antenna DSSs are under construction at Madrid and Canberra, which will operate in conjunction with DSS 14 to provide this capability. These stations are scheduled to be operational by the middle of 1973. In 1959, a mobile L-band compatibility test station was established at Cape Kennedy to verify flight-spacecraft- DSN compatibility prior to the launch of the Ranger and Mariner Venus 1962 spacecraft. Experience revealed the need for a permanent facility at Cape Kennedy for this function. An S-band compatibility test station with a 1.2-m (443) diam antenna became operational in 1965. In addi- tion to supporting the preflight compatibility tests, this station monitors the spacecraft continuously during the launch phase until it passes over the local horizon. Spacecraft telecommunications compatibility in the de- sign and prototype development phases was formerly ver- ified by tests at the Goldstone DSCC. To provide a more economical means for conducting such work and because of the increasing use of multiple-mission telemetry and command equipment by the DSN, a compatibility test area (CTA) was established at JPL in 1968. In all essen- tial characteristics, the configuration of this facility is iden- tical to that of the 26-m (85-ft) and 64-m (210-ft) diam antenna stations. The JPL CTA is used during spacecraft system tests to establish the compatibility with the DSN of the proof test model and development models of spacecraft, and the Cape Kennedy compatibility test station is used for final flight spacecraft compatibility validation testing prior to launch. The GCF provides voice, high-speed data, wideband data, and teletype communications between the SFOF and the DSSs. In providing these capabilities, the GCF uses the facilities of the worldwide NASA Communica- tions Network (NASCOM)3 for all long distance circuits, except those between the SFOF and the Goldstone DSCC. Communications between the Goldstone DSCC and the SFOF are provided by a microwave link directly leased by the DSN from a common carrier. Early missions were supported by voice and teletype circuits only, but increased data rates necessitated the use of high-speed circuits for all DSSs, plus wideband cir- cuits for some stations. Network and mission control functions are performed at the SFOF at JPL. The SFOF receives data from all DSSs and processes that information required by the flight project to conduct mission operations. The follow- ing functions are carried out: (1) real-time processing and display of radio metric data; (2) real-time and non-real- time processing and display of telemetry data; (3) simula- tion of flight operations; (4) near-real-time evaluation of DSN performance; (5) operations control, and status and operational data display; and (6) general support such as internal communications by telephone, intercom, public address, closed-circuit TV, documentation, and reproduc- tion of data packages. Master data records of science data received from spacecraft are generated. Technical areas are provided for flight project personnel who analyze spacecraft performance, trajectories, and generation of commands. 3Managed and directed by the Goddard Space Flight Center. 1. The Deep Space Network, Space Programs Summary 37-50, Vol. IT, pp. 15-17. Jet Propulsion Laboratory, Pasadena, Calif., Mar. 31, 1968. J. E. Maclay D S N Engineering and Operations Office A new network capability for high-speed transfer of operational control infor- mation has been implemented. The increase in speed is attributable to two things: use of high-speed data line instead of teletype yields a 24-fold increase, and source data is input in machine language instead of manually, yielding an additional increase. A new DSN operations control system capability to send operational traffic from the SFOF to the DSSs by high-speed data (HSD) line has been implemented. There are currently three types of traffic that go by this mechan- ization: tracking predicts, DSN sequence of events, and DSN 7-day schedule. The implementation uses the standard GCF HSD sys- tem, but required new designs in SFOF and DSIF. The encoding, formatting, and outputting is done in the SFOF by the operations control output router, a software part of the DSN monitor and operations control real-time processor. Reception and decoding are done at the DSS as part of the DSIF monitor program in the digital instru- mentation subsystem Phase I1 (DIS 11), with printout on the DIS line printer (and also a Magpak recording of tracking predicts). To have a backup, and because not all DSSs have a DIS 11, the output router can also output in Baudot code over teletype (TTY) lines for printout on a TTY machine, although any line of printing over 69 characters long is truncated (as opposed to a 132-character line on the DIS line printer). For the two reasons given, tracking predicts and DSN 7-day schedules are constrained to a maximum of 69 characters per line, even when sent via HSD to the DIS line printer. The DSN sequence of events uses all 132 characters of a line printer, and thus cannot be sent by TTY. This is not considered to be a problem, as voice can be used as backup until reception capability is re- stored. (Prior to Pioneer F support, the remaining DSSs will also have this reception capability.) The Mark I1 DSN had an output capability, but it was limited to transmitting tracking predicts by TTY; pre- dicts generated on the 7094 were converted to Baudot code in the 7044 and sent to the GCF comm processor without personnel handling during the process. The DSN 7-day schedule and the sequence of events were hand- written for keyboard entry into the comm processor. The new output router can accept any type of traffic which is in machine language, does a code conversion (to XDS 6-bit BCD for DIS page prints, XDS floating point for Magpak recordings, and Baudot code for TTY transmis- sion), reformats if required, adds the blanks and carriage functions needed in the page print output format, and meters the output onto GCF lines. Currently, “machine language” means EBCDIC1 coding on either magnetic tape or computer files, but may later be expanded to include the Univac 36-bit coding utilized .in an available RF predicts program. Comparison of the HSD medium and the TTY medium is straightforward. One line of print is put in each HSD block, and since the DIS line printer is capable of print- ing faster than the 4 HSD block per second rate of GCF, the network-wide speed is 4 printed lines per sec- ond via HSD line. TTY requires 6 s per printed line, so IExtended binary-coded decimal interchange code. transmission by HSD is 24 times as fast as by TTY. Thus, predicts for initial acquisition, which require 14 min by TTY, will go in 35 s by HSD. For other traffic, the ratio is much higher when one considers that in the new system, a sequence of events (for example, one 200 items long) file is read and output and printed in less than a minute, whereas in the Mark I1 era it required several man-hours to condense a sequence to TTY page size, and then it was manually put onto TTY lines. This capability clearly will help the DSN to give bet- ter support to missions with high activity, and to reduce station turnaround times, b 5 A. J. Siegmeth Mission Support Office This article describes the design profile of the Pioneer F and G missions. The characteristics of these flights that interface with Deep Space Network tracking and data acquisition support are depicted. A delineation of the mission descrip- tion and a summary of spacecraft systems and subsystems are giuen. Reference 1 contains a summary of the tracking and data system planning activities for the Pioneer F and G missions. This article begins the description of Pioneer F and G missions, with special emphasis on characteristics that interface with the tracking and data acquisition functions. DSN planning efforts during the design phase of the spacecraft hardware were centered around the assurance of a functional compatibility between the spacecraft design and the configuration of the DSN Mark I11 system. In this article, Pioneer F and G mission profiles are presented. Spacecraft systems are briefly described and the following subsystems are outlined : electrical power supply, electrical power conditioning, structure, thermal control, propulsion, and attitude control. Future articles will contain: (1) a description of the telecommunications and antenna subsystems; (2) a summary of launch and space flight trajectories, giving tracking and data acqui- sition constraints; (3) the configuration of the six DSN systems; and (4) the planning activities related to navi- gational requirements. eer issi ile The mission objectives of the Pioneer F and G space- craft are to conduct, during the 1972-73 Jovian oppor- tunities, exploratory investigation beyond the orbit of Mars of the interplanetary medium, the nature of the asteroid belt, and the environmental and atmospheric characteristics of the planet Jupiter. As the earth and Jupiter orbit around the sun, their relative position permits a spacecraft to be launched every 13 months into a Jupiter-bound trajectory with a minimum of launch energy. During optimum conditions, injection velocities of approximately 14 km/s will suffice; a favorable launch period can cover several weeks. Dur- ing the remainder of the 13-month interval, the velocity requirements will rise to prohibitive values. Among the available launch vehicle configurations, the Atlas/ Centaur/TE 364 vehicle was chosen for Pioneer F and G missions. The launch energy generated by this vehicle will make possible the launch of Pioneer F during late February or early March 1972, and Pioneer G during April 1973. These missions will make possible the initial direct exploration of vast regions of the solar system from 1 to 6 AU from the sun, including the vicinity of 6 the planet Jupiter. The asteroid belt is located in this region between 2 and 3.6 AU from the sun. Pioneer F will be the first NASA spacecraft to explore the outer planet environment beyond the orbit of Mars. It will take between 600 and 800 days to travel from the earth to the vicinity of Jupiter, using trajectories compatible with the favorable launch opportunities. The spacecraft are equipped with field-and-particles and optical-type instruments. Nearly all the time during travel between the earth and Jupiter will be spent in the interplanetary solar-wind environment. The influences of the earth's magnetosphere will cease several hours after launch. The spacecraft will fly through the high-density Jovian magnetosphere to explore the trapped radiation particles of the solar system's largest planet. The general relationship of a typical Pioneer F trajectory to the sun, earth asteroid belt, and Jupiter is depicted in Fig. 1. EARTH AT ENCOUNTER A simplified mission profile of Pioneer F is shown in Fig. 2. The telecommunications range at Jupiter encoun- ter will be approximately 5% AU and, after the encounter of Jupiter, Pioneer F will move toward the outer planets and eventually leave the solar system. In contrast, the Pioneer G trajectory profile is such that this spacecraft, after the Jupiter encounter, will stay within the solar system as a solar orbiter. The Pioneer F and G missions will require continuous tracking and data acquisition coverage by the 26-m an- tenna network from liftoff to Jupiter encounter plus 6 months. The flight project also expects that during critical mission events the 64-m antenna stations will be continu- ously available to accomplish mission objectives. In addition, the project plans to obtain from the 64-m an- tenna stations at least one horizon-to-horizon tracking pass per week from liftoff to encounter plus 6 months. EARTH AT H ELlOCENTRlC TRAJECTORY: ESCAPE FROM SOLAR SYSTEM ENCOUNTER INTERPLANETA 7 REWRN TO CRUISE STATUS LIFTOFF AND ENTER ECLIPSE After the 'Jovian encounter plus 6 months, the tracking and data system should furnish, as a minimum, daily coverage for both missions until the limit of the receipt of downlink data signal. It will be a continuous challenge for the DSN to acquire and collect the data stream transmitted by the Jovian missions. Variations of the solar activities will naturally have a continuous impact on the measurements made by the field-and-particles instruments relating to the solar wind and the interplanetary magnetic field. Pioneer F and G spacecraft will reach the asteroid belt approximately 6 months after launch and will stay in this belt for more than 6 months. Some astronomers hypothesize that the asteroid belt might be composed of the remnant of a planet which was circling the sun at 2.8 AU. Seventy-five thousand bodies of the asteroid belt are larger than magnitude 20; the largest is Ceres with a diameter of 768 km. The exploration of the asteroid belt can lead to the understanding of the origin and evolution of the solar system; therefore, the data col- lected by the DSN can have a high scientific significance. To give a better idea of the challenge of a Jupiter flyby, a few of the important characteristics of the largest solar planet are given: Jupiter's average distance from the sun is 5 AU, which is equivalent to approximately 75 X lo7 km. Its mass is equivalent to 318 times the earth's mass and comprises 70% of all planetary mass. Jupiter's diameter (approximately 71,387 km) is 11 times the earth's diameter. It is remarkable that Jupiter's density is only approximately one-fourth of the earth's density and its fastest rotation period is 9 h and 50 min. In other words, one Jupiter day is almost 10 h long. Jupiter has 12 satellites, 4 of them having a magnitude equivalent to the earth's moon. The four Jupiter moons are called IO, Europa, Ganymede, and Callisto. Jupiter has many varying striped surface features. The red spot is about three times the size of earth. It is assumed that Jupiter's upper atmosphere is composed of hydrogen, helium, amonnia, and methane. It is an intense radio noise source at decametric frequencies. Fortunately, this noise drops considerably toward the S-band frequency range. The typical cold sky system noise temperature of the Gold- stone 64-m antenna station is 25°K. If one moves this antenna to the direction of Jupiter, the system noise temperature will increase to approximately 30°K. This increase of 5°K can cause a telemetry signal-to-noise degradation of around 0.5 dB. The dense magnetosphere of this large planet is caused by an extremely high mag- netic field, which can be as high as 5 X le4 T (5 gauss). To place the spacecraft on a trajectory to the desired target point in the vicinity of Jupiter, one or more mid- course corrections have to be made to compensate for launch vehicle injection velocity vector errors. Flight project plans to perform the first maneuver within the first ten days and the second, if necessary, about 30 days after launch. The medium and the 2.75-m-diameter high-gain an- tenna of the Pioneer F and G spacecraft have to con- tinuously point toward the earth during the entire flight to assure effective data return (Fig. 3). Therefore, pre- cession maneuvers will be conducted throughout the mission as required to maintain an earth-pointing cruise attitude. These maneuvers, homing into the uplink signal transmitted by the DSN toward the spacecraft, will take place only once every 2 or 3 days during the early phase of each mission, and will decrease in frequency to once in 1 or 2 weeks by the time of Jupiter encounter. Approximately 315 days after launch, the relative posi- tion of the spacecraft and earth will place the spacecraft in a superior conjunction configuration versus the sun and the earth. Because the spacecraft is somewhat out of the ecliptic plane, the spacecraft/earth line will not intercept the sun but will come within a few solar radii. In this configuration, the radio beam will be intercepted through the high-density part of the solar corona and will be influenced significantly by the plasma. Because of the closeness of the spacecraft to the sun, DSN an- tennas will pick up, together with the spacecraft signal, solar high-frequency noise, which will degrade the 8 received signal-to-noise ratio and can cut down consid- erably the quality and usefulness of the spacecraft/earth telecommunication link. This condition will exist for approximately 1 or 2 weeks. To optimize the scientific value of the data return dur- ing the Jovian encounter, the flight project has selected specific flyby trajectories. The objectives of the encounter trajectory for Pioneer F are: To penetrate the Jupiter radiation belt. To provide good viewing conditions for Jupiter be- fore periapsis. To obtain a short occultation for the spacecraft by Jupiter (less than 1 h). To provide a radius of closest approach to the center of Jupiter between 2 and 3 Jupiter radii. The typical trajectories will approach Jupiter in a counterclockwise direction, as viewed from the celestial North Pole, will meet the planet slightly south of the ecliptic, pass around the planet in a counterclockwise direction to an attitude between 1% and 2 Jupiter radii above the surface of the planet, and will exit slightly north of the planet. The postencounter trajectory will have an inclination to an ecliptic of less than 5 deg and the spacecraft will slowly escape the solar system. An additional 3 years will be required to cross Saturn’s solar orbit. The Pioneer G trajectory and sequence of events up to Jupiter encounter will be similar to that of Pioneer F. During the voyage from earth to Jupiter, the space- craft has to travel through a hostile environment, which can directly or indirectly affect the flight hardware and the operational characteristics of the spacecraft transpondedantenna system and can also constrain data acquisition and collection by the DSN. Special efforts will be made to study and analyze all possible constraints and to plan a configuration that will operate properly with these constraints and assure the best data return. A brief summary of some of the space hazards the space- craft can be exposed to follows: (1) As the spacecraft rushes through the earth’s Van Allen belt, the instruments will detect vast intensity variations of high-energy protons and electrons. During this early deep space phase, the DSN must assist the Mission Operations Team in transmission of numerous commands and must col- lect all telemetry data. As the spacecraft travels through the asteroid belt, onboard sensors will attempt to detect meteoroids. During this phase, there is a very small probability that the spacecraft may collide with one of the meteoroids. This can cause an abrupt change in the spacecraft attitude and a loss of the spacecraft high-gain antenna radio link. In this event, the DSN must attempt to acquire spacecraft signals radiated by the medium-gain or omni-antennas. The electrons, protons, and magnetic field in the Jovian belt have a much higher intensity and flux density than in the Van Allen belt. It is possible that the crystal oscillators of the spacecraft trans- ponder will slightly detune as the spacecraft tra- verses the Jovian belt. The high-intensity Jovian magnetosphere can also change the polarization ellipticity of the spacecraft signal radiated toward earth. In addition, the spacecraft flying in the close vicinity of Jupiter undergoes an abrupt velocity change which causes, within 2 or 3 h, a consider- able change in the doppler shift; this shift has to be tracked by the spacecraft receiver and by the DSN. ecr The Pioneer F and G spacecraft are carrying, as pri- mary electrical power sources, radioisotope thermoelec- tric generators (RTGs) provided by a contractor of the Atomic Energy Commission. The spacecraft has been designed by the TRW Systems Group on contract with the NASA/Ames Research Center. The production and testing of the spacecraft will also be performed by the same vendor. The basic design requirements of the Pioneer F and G spacecraft can be summarized as fol- lows: (1) Be compatible with the Atlas SLVSC/Centaur/ BE-364-4 launch vehicle. (2) Be compatible with the Mark HI1 DSN system. (3) Provide a thermally controlled compartment to house the scientific instruments. 3 vide a telecommunications data system to sample readings from the instrumentation and transmit the information to earth. (5) Provide a system to permit changes in operating modes of onboard equipment on command from earth. (6) Provide a magnetically clean and an electromag- netic interference-free spacecraft. (7) Operate in space for more than 2 years and for distances beyond' the orbit of Jupiter. To make the telecommunications subsystem of the spacecraft compatible with the Mark I11 DSN system, specific capabilities must exist. The spacecraft trans- ponder must operate at predetermined DSN S-band fre- quencies and should have the capability to operate in a coherent two-way mode having a fixed ratio between the uplink and downlink frequencies. This capability is necessary to extract at the deep space stations accurate doppler measurements to determine the accurate space- craft velocity relative to earth. These measurements are necessary for trajectory determination and for the gen- eration of predictions required by the deep space stations to make an effective acquisition of the data link possible. The spacecraft must also be equipped with an auxiliary onboard oscillator to provide a downlink during times when the deep space stations have not established a two- way uplink/downlink telecommunications configuration. The Pioneer Project Office of NASA/Ames Research Center plans to control every element of the mission system by a project documentation system. Pioneer F and G Project Specification PC-210, Spacecraft and Related Requirements, describes the functional requirements that the space vehicles must meet to achieve the primary and secondary mission objectives. This key document de- scribes the expected environment through which the spacecraft will pass, and the required characteristics and performance of the spacecraft systems and subsystems. In addition, it specifies the reliability, quality assurance, and test programs to be conducted at the subsystem and system levels and the requirements for the activities at the launch complex to prepare the spacecraft and scientific instruments for launch. This document references the Pioneer F and G specification PC-222, Spacecraft/DSIF Interface Specification. A subset of this document is PC-222.02, which defines the characteristics of the equipment of the deep space stations pertinent to the spacecraft and the requirements imposed on the spacecraft by the DSN. The contents d this document were collected by the Project office from the Standard Practice, 810-5, Revision A, C h h g e 1, Deep Space Networkmght Project Interface Design Hand- book. DSN developed this handbook for use by all flight projects as a standard source of advanced technical information that describes the DSN interfaces with the Project office in the fields of telecommunications, data processing, and simulation. Since the Pioneer project uses the contents of this interface design handbook, assurance can be made that, on a functional level, the project interfaces will be compatible with the DSN con- figuration. To validate the functional compatibility be- tween spacecraft design and the Mark I11 DSN system, plans are underway to make RF and data system com- patibility tests between Pioneer F and G flight hardware at the Compatibility Test Area located at JPL. A veri- fication of this compatibility will be made a few days prior to launch at DSS 71, located at Cape Kennedy, during the final checkout of the spacecraft and launch vehicle. Based on the contractual functional design require- ments of the spacecraft and its telecommunications sub- system, the spacecraft contractor, under the guidance of the Pioneer Project office, has developed the flight hardware. The basic geometry of the Pioneer F and G spacecraft is shown in Figs. 4 and 5, and a perspective view is given in Fig. 6. The spacecraft essentially has two thermally controlled equipment compartments, one hexagonally shaped and containing spacecraft equipment and the other an appendage containing scientific instruments. Forward of the equipment compartments is a 2.75-m- diameter parabolic reflector, which is a basic component of the spacecraft high-gain antenna. Mounted on three struts forward of the reflector are the medium-gain antenna and the feed for the high-gain parabolic reflector. Electric power is supplied by four RTGs mounted in pairs on two radially deployable trusses. The generators are in a stowed position for launch, next to the equipment compartment and under the reflector of the high-gain antenna. The angle between the support trusses for the generators is 120 deg. In the deployed position, the generators extend well beyond the perimeter of the reflector to reduce the radiation environment within the equipment compartments and to reduce their magnetic influence on the magnetometer. This latter instrument is located on the end of a long folding boom, which, in the deployed condition, extends radially from the instrument side of the equipment compartment. -1 . II 1 H I GH-GAI N PRKESSION AND ANTENNA FEED VELOCITY THRUSTERS PLASMA ANALYZER HIGH-GAIN RTG DEPLOYMENT DAMPER CABLE ELECTRIC POWER CABLE DAMPER CABLE THERMAL CONTROL LOUVERS OMNl-ANTENNA ASTEROI D IMETEORO~ ig. ioneer cecraft (side view) DETECTOR Numerous viewing apertures are provided in the equipment compartments, as required by the scientific instruments. Mounts, external to the equipment com- partment, have been provided for the meteoroid and asteroid instrumentation. The spacecraft is spin-stabilized with a spin rate some- what below 5 rpm. The launch vehicle’s spin-up system brings the spacecraft, together with the third stage of the launch vehicle, up to a rotational speed of around 60 rpm. After the third-stage burnout and separation, the spin rate will be slowed down from 60 to 20 rpm by the use of automatically fired thrusters. Finally, deployment of the RTGs and the magnetometer will further de-spin the spacecraft to a nominal spin rate of 4.8 rpm. The objective of this spin stabilization is to stabilize spacecraft attitude. Spin-axis precession maneu- vers will be applied during the mission to orient the spin axis of the spacecraft to the earth and thus illumi- nate the earth with the directional beams of the medium- gain and high-gain antennas. Because the geocentric longitude of the spacecraft varies over a wide range during the mission, the space- 1 craft has a monopropellant attitude-control system which is capable of precessing the spin axis and maintaining the earth-pointing attitude on command. The initial re- orientation to point the spacecraft spin axis toward the earth will occur within a few hours after launch. The spacecraft is equipped with a complete telemetry and data handling system which will generate a data stream containing the output of the scientific instruments and spacecraft equipment measurements. A system for receiving the modulating and distributing command instructions received from earth will provide flexibility in operation of the scientific instruments and the space- craft. Spacecraft equipment will deliver conditioned power to the scientific instruments and also supply the instruments with appropriate timing and orientation indexing signals for control of measurements and data accumulation. s The primary electrical power source consists of four Snap-19 radioisotope thermoelectric generators (RTGs). MAGN ETOMET ER SENSOR 7 OMNl-ANTENNA MET EOROl D D ETECTORS THERMAL CONTROL LOUVERS ANTENNA REFLECTOR PRECESSION AND VELOCITY THRUSTERS PRECESSION AND VELOCITY THRUSTERS SUN SENSOR ELECTRIC POWER CABLE DEPLOYED RT ASTEROI D/METEOROI DOMPARTM ENT DEPLOYED R T G s PARABOLIC HIGH- GAIN ANTENNA MAGNETOMETER INSTRUMENTS -%J 13 The four units are in pairs in a tandem arrangement on the end of two supporting trusses. Provisions are made for retracting the RTGs to fit within the nose fairing of the launch vehicle for launch and for deploying them radially to the extended position after separation from the launch vehicle. Figure 7 shows the spacecraft mounted in the nose fairing and as assembled for the launch phase of the mission. Each RTG unit will provide almost 40 W of electrical power during the early part of the mission and about 30 W 5 years after launch. It is expected that the power requirements of the spacecraft and scient& instruments will not exceed the capability of three of the RTG units at the time of Jupiter encounter. Therefore, the prob- ability of having adequate power available at encounter is very high. The net weight of each RTG is around 13.6 kg. The objective of the electrical power subsystem is to distribute and condition the power received from the RTGs to the spacecraft equipment and the scientific instruments. To meet the requirements of the power loads, the dc output of each RTG goes into a separate inverter. The 2.5-kHz squarewave output of the four inverters is fed into an ac bus. Most of the ac power is rec&ed and filtered to supply the main dc bus, which is shunt-regulated to 28 V 2 1 % by dumping excess power through an external shunt radiator. The regulation of the 28-V dc bus is reflected back through the ac bus and through the fixed-ratio inverters, fixing the RTGs MAN ACCESS \ wow 1 PIATFOWS /-NOSE FAIRING /- %%TALIATION HOLE ---LAUNCH VEHICLE THIRD- STAGE n operating voltage at 4.2 V. A battery automatically carries any temporary overloads and is recharged auto- matically when excess power is available. The scientific instruments and the traveling-wave-tube power ampli- fier of the transmitter section of the transponder receive power from the main dc bus. Most of the other space- craft loads are supplied from the central transformer- rectifier-filter unit, which receives power from the ac bus and provides various dc output voltages. The hexagonal equipment compartment is the basic structural element of the spacecraft. It supports the high-gain antenna on its forward end, attaches the launch vehicle by the launch-vehicle mating ring at its aft end, supports the scientific instrument compartment, accom- modates the major portion of other subsystem assemblies, and provides for mounting of various external compo- nents, including the RTGs and the magnetometer boom. Rigid tubular truss work attached to the framework of the equipment compartment supports the parabolic reflector of the high-gain antenna, the high-gain antenna feed, the medium-gain antenna, the three thruster clusters, the attachment for the deployable booms for the RTGs, and the launch-vehicle mating ring. The reflector of the high-gain antenna is made from an aluminum honeycomb sandwich. The thermal control subsystem provides the required thermal environment for the spacecraft components and scientific instruments during all phases of the flight. The objective of the thermal control subsystem is to maintain, in the vicinity of the scient& instruments, temperatures between -18 and t 3 8 C and to keep the spacecraft equipment at temperatures required for satisfactory operation. Heating effects caused by the RTGs (stowed within the nose fairing), the third-stage motor casing, and the jet plume must be accommodated. The large variations in solar intensity and relative direction during interplanetary flight, and the loss of heat from the equipment compartments through sensor apertures, are compensated through louvers and special installation. lsio I During the major portion of the flight, earth-pointing attitude is necessary for the narrow beam of the high- gain antenna of the spacecraft to illuminate the earth and to maintain effective communications. Figure 8 depicts T . 88 4 - - - - - - \ 3 270. the relative position of the spaoecraft’s spin and antenna axis versus the earth. It can be seen that the geocentric longitude of the spacecraft varies continuously through- out the mission and, therefore, it is necessary to make numerous attitude adjustments. Since the spacecraft high-gain antenna has a half-power beamwidth of ap- proximately 3.5 deg, it is anticipated that more than 200 spin-axis orientation maneuvers will be necessary to com- pensate for the relative movement of the spacecraft versus earth, and also for the precession caused by solar pressure, which is 0.2 deg per day during the early part of the mission. In addition, to provide the planned encoun- ter trajectory at Jupiter, some adjustments of the velocity vector may be required during the interplanetary aght. To make the velocity vector adjustments possible, thrust must be generated in a particular direction; therefore, with thrustors in a fixed relationship to the spacecraft, reorientations of the spacecraft will also be necessary. Changes in the spin rate will also be required. After injection, the rate will be on the order of 60 rpm and must be reduced to about 20 rpm before the deployment of the magnetometer and RTCs. As a result of the de- ployment, the spin rate is reduced to between 4 and 5 rpm (nominal: 4.8 rpm). In addition, to make possible attitude and velocity changes, small changes in the spin rate have to be corrected to maintain the rate of spin within the required limits. Changes in attitude, velocity, and spin rate during the interplanetary agh t will be accomplished by mmopro- elant hydrazine thrusters. Thrust will be provided by ermic decomposition of the hydrazine in a catalyst bed and extension of the gas through a nozzle. Figure 9 depicts the location of the attitude-control spin control, velocity, and precession thrusters at the edge of the para- bolic high-gain antenna structure. These thrusters can be operated in pairs. Each cluster contains a forward- facing nozzle and a rearward-facing nozzle. Two forward or two rearward nozzles will be used for velocity adjust- ment and opposite-facing nozzles will be fired for attitude changes, causing precession of the spin axis. Spin-rate changes will be accomplished by tangentially aligned nozzles thrusting with and against the spin. A sun or star sensor (Fig. 10) provides reference signals necessary to time the thrust pulses for precession of the spin axis in a desired direction (Fig. 11). Attitude changes can be accomplished “open loop” by ground command, or “closed loop” by homing the spacecraft on the S-band uplink signal radiated by a deep space station toward the spacecraft. The duration of thrust for velocity and spin-rate changes is established by ground calculations. This information is transmitted to the spacecraft, where it is stored for execution on command. Similarly, for an “open loop” reorientation, the direction and amount of precession desired can be transmitted to the spacecraft via the command link and stored for execution on com- mand. This storage information can be combined to per- form a precession-velocity change-precession sequence with suitable time intervals in the sequence and to pro- vide a completely automated velocity vector adjustment with return to a selected spacecraft orientation. A closed-loop decision maneuver will be used regularly for accurate realignment of the spin axis of the space- craft toward earth. A medium-gain and a high-gain spacecraft antenna will be used, respectively, for course and fine homing on the uplink and telecommunications signal. For the closed-loop maneuver, the axis of the medium-gain antenna and the feed of the high-gain an- tenna are offset from the spin axis and provide an amplitude-modulated signal when the spin axis is not aligned with the earth. The so-called CQNSCAN subsys- tem processes this signal and fires the precession thrus- ters to establish the required precession and orient the spin axis toward earth. The hydrazine will be supplied to the thrusters through appropriate lines and valves from a single, spherical pres- surized bladder-tank, which is located in the center of the spacecraft equipment compartment. small radioisotope heaters will be used to keep the plumbing of the hydrazine system above 2”C, which temperature keeps the fuel in a liquid state. SPIN CONTROL THRUSTERS 1 -y /- (DE-SPIN) AND 2 (SPIN) STELLAR REFERENCE ASSEMBLY VELOCITY/PRECESSION THRUSTERS 3 (FORWARD) AND 4 (AFT) -X I AND 2 (AFT) X=0.2m Y = O f = 0.48 m FROM SPACECRAFT SEPARATION PLANE ’ 25 deg I -- I +Y CHANNEL FIELD OF VIEW FROM X-2 PLANE, deg FIELD OF VIEW FROM Z-AXIS, deg - 0.5 0.5 90 deg CHANNEL NNEL2 *12 12 - 90 90 - 170 7 - A Y I C -12 SUN SENSOR ASSEMBLY FIELD OF VIEW, deg 30 xO.2 STELLAR REFERENCE ASSEMBLY 14 r 32- . II CLOSED-LOOP EARTH TRACKING PERFORMANCE PROPELLANT -27.2 kg HYDRAZINE COMMAND O N OPEN-LOOP PRECESS I ON , PHASE I 1 DE-SPIN STORED COMMANDS MAGNITUDE SUN SENSOR GENERATOR CAPABI LlTY 200 rn/s +1250-deg PRKESSION + 14 rprn + DE-SPIN PRESSURE BLOWDOWN: 369 - 93 N/crn2 (535 - 135 psi) THRUST (ONE THRUSTER) 0.52 - 0.24 kg SPECIFIC IMPULSE 215 s (CONTINUOUS) 140 s (PULSED) MAGNl TU DE THRESHOLD VELOCITY/PRECESSI ON (P) THRUSTERS (4) SPIN CONTROL (5) THRUSTERS (2) ELECTRON IC S 1 I 1 SPIN AXIS DE-SPIN SENSORS S P I N CONTROL 1. Siegmeth, A. j., “‘Pioneer Mission Support,” in The Deep Space Network, ograms Summary 37-66, Vol. 11, pp. 411. Jet Propulsion Laboratory, Pasadena, Caw., Nov. 30, 1970. 17 P. S. Goodwin Mission Support Office This article relates the historical factors that led to the establishment of Project Helios; describes the project management relationships between the United States and the Federal Republic of West Germany, and the role of the Deep Space Network; and gives a brief description of the spacecraft and its telecommunications subsystem. 1. The Space Act of 1958 authorized the National Aero- nautics and Space Administration (NASA) to conduct programs of international cooperation with other nations in the peaceful exploration of space. To implement this article in the NASA Charter, the U.S. National Academy of Science, in behalf of NASA, introduced an offer of international space cooperation to the Committee on Space Research of the International Council of Scientific Unions. This, in turn, led to discussions between United States and Federal Republic of West Germany officials on the subject of a possible advanced cooperative project in space. An agreement in principle was made between the German Minister for Scientific Research, Gerhard Stoltenberg, and NASA Administrator James E. Webb in September of 1966 to carry out a cooperative solar probe project-provided that a mission of mutual interest could be d This decision was mad Chancellor d‘s visit with President vember 1966. This project was subsequently named Helios after the ancient Greek goddess of the sun. To implement this agreement, NASA and the West German Minister for Science established a Helios Mis- sion Definition Working Group in July P W . This group’s effort culminated in the publishing of the Helios Program “Mission Definition Group Report,” dated April 1969. This report recommended that two solar probes be launched toward the sun-one each in calendar years 1974 and 1975-to achieve a perihelion distance from the sun of approximately 0.3 AU. Each spacecraft would carry ten experiments to perform fields and particle measurements in the region between the earth and the sun. The target perihelion distance of 0.3 AU was selected because it was a heretofore unexplored region of inter- planetary space and was within the estimated extrapola- tion of the state-of-the-art for the high-temperature solar cells necessary to generate spacecraft power. The size and weight of the proposed Helios spacecraft was recom- mended to be compatible with the Ath/Centaur launch vehicle, with the addition of a TE-364-4 solid propellant third-stage. The publication of the port was followed by a “ ing” between NASA Administrator Thomas German Science Minister Gerhard 1969, and by joint statements by Chancellor Kiesinger in August 1969. These documents ratified the activities to date and established the basis 1 . 1 for the future relationships between the two participat- ing countries. The Helios Project would be managed jointly by both countries, with each co-manager being responsible for those elements of the cooperative project that were assigned to his particular country. In brief, the West German Project Manager would be responsible for the design, development, and fabrication of the space- craft and the mission design and operations, while the U.S. Project Manager would be responsible for the launch vehicle, the launch facilities, and the tracking and data system. Of the ten onboard scientific experiments, seven were to be of German origin, and three of U.S. origin- however, the integration of all ten experiments into the spacecraft would be the responsibility of the West Ger- mans. Each country would be responsible for providing the funding necessary to accomplish its portion of the program. Coordination of the effort was to be achieved via semiannual Joint Working Group Meetings to be held alternately in each country. The first Helios Joint Work- ing Group Meeting was held in September 1969 in Bonn, West Germany. Subsequently, two additional Joint Working Group Meetings have been held: the second in April 1970 at Goddard Space Flight Center, Greenbelt, Maryland; and the third in October 1970 in Bonn. The next Helios Joint Working Group Meeting is scheduled for late April 1971 at Goddard Space Flight Center. In addition to the scientific objectives mentioned above, a second and very important objective of the Helios Project is to develop a broad governmental-educational- industrial technological base within the Federal Republic of Germany to conduct space research. Therefore, the West German participation in the Helios Project is not solely restricted to the development of the spacecraft and the mission design, but also includes the develop- ment of German tracking facilities, a German Control Center, and a full mission operations organization to con- duct the mission. In addition, the international agree- ment provides for the cross-training of a significant number of West German specialists at various NASA installations to learn U. S. techniques pertaining to space exploration. These factors, together with the interna- tional character of the project, account for the Lcommittee-like” structure of the Helios Project rganization (Pig. 1). The upper lefthand quadrant of Fig. 1 depicts the familiar NASA flight project organizational structure wherein NASA eadquarters assigns the project man- agement responsibility to one of its field centers, with functional support in spec& areas being provided by other NASA field centers. The significant difference here is that only a portion of the elements comprising a total &ght project are represented. The missing elements ap- pear on the West German side of the interface in the upper righthand quadrant of Fig. 1-along with some new elements due to the factors mentioned above. The West German Helios Project management is seen to I parallel and complement the U.S. Helios Project organ- ization so that, in total, the top half of Fig. 1 represents the formal international project organization for Project Helios. An important advantage of this formal structure is that it provides a clear and distinct division of re- sponsibility between the two countries in the admin- istration, technical supervision, and financial management of the Helios Program. As mentioned above, the technical coordination of the two countries’ efforts is accomplished via the Helios Joint Working Group activities. The Helios Joint Working Group organizational structure is depicted in the bottom half of Fig. 1. In accordance with the international agree- ment, the Joint Working Group Meetings are co-chaired by the U. S. and West German Helios Project managers, respectively. Reporting to them are the chairmen of the various technical subgroups that support the project. These subgroup chairmen are the same individuals who have been assigned the equivalent functional responsi- bility in the formal project organization depicted in the top half of Fig. 1. wever, the subgroup panel member- ship within each these technical subgroups is fairly evenly divided between the U. S. and West German representatives in order to achieve internationally opti- mum solutions to problems facing the project. During the semiannual Helios Joint Working Group sessions, these subgroups meet both individually to resolve prob- lems within their own areas of specialization, and jointly to resolve problems associated with the interface be- tween the areas of technical responsibility. When nec- essary, the activities of the subgroups are augmented by special task or study groups assigned to investigate in detail a particular aspect of the program. A recent and successful example of the latter was the establishment of a special study group to develop a typical near-earth phase sequence of events following launch in order to determine that the spacecraft design as currently con- templated will f u W all operational constraints upon the mission. This study group was chaired by the Mission Analysis and Operations Subgroup and its membership was comprised of representatives from each of the other subgroups. Once it has accomplished this initial task, the ase Study Group will probably become NATIONAL GOVERNMENT LEVEL (PROGRAM MANAGEMENT) MINISTRY FOR PROJECT LEVEL SUBGROUP LEVEL JOINT SUBGROUP LEVEL* STUDY GROUP LEVEL* AERONAUTICS 1-T -1 SCIENCE AND AND SPACE EDUCATION ADMINISTRATION (BMBW) MANAGEMENT CENTER LEVEL --- ----- PROJECT ORGANIZATION JOINT WORKING GROUP -------- ORGANIZATION CHAIRMAN LEVEL SPECIAL STUDY GROUPS NEAR-EARTH PHASE * ONLY PRESENT ACTIVITIES PERTAINING TO THE TDS ARE DEPICTED 20 AfW BMBW DFVLR GfW GSFC HJWG KSC LeRC MA&O TDS ARBEITSGEMEINSCHAFT FUER WELTRAUMFORSCHUNG (COOPERATIVE FOR SPACE RESEARCH) DER BUNDESMINISTER FUER BILDUNG AND WISSENXHAFT (FEDERAL MINISTER FOR EDUCATION AND SCIENCES) (GERMAN RESEARCH AND EXPERIMENTAL INSTITUTION FOR AEROSPACE) GESELLSCHAFT FUER WELTRAUMFORSCHUNG, mbH (SOCIETY FOR SPACE RESEARCH) GODDARD SPACE FLIGHT CENTER HELIOS JOINT WORKING GROUP KENNEDY SPACE CENTER LEWIS RESEARCH CENTER MISSION ANALYSIS AND OPERATIONS TRACKING AND DATA SYSTEM DEUTSCHE FORSCHUNGS-UND VERSUCHSANSTALT FUER LUFT-UND RAUMFAHRT E. V. dormant until the final launch trajectories are developed. At that time, the group is expected to reconvene to prepare the final near-earth phase sequence of events. The technical decisions reached during the Helios Joint Working Group Meetings are reviewed by the co- chairmen. Upon their approval, these decisions are routed via the respective project office to the formal organization (top portion of Fig. 1) for implementation. In the case of decisions affecting the U. S. Tracking and Data Sys- tem (TDS), these are routed to the Jet Propulsion Lab- oratory as the cognizant NASA field center for Helios TDS management. The NASA TDS function for Helios has two major subdivisions: (1) support from the near- earth phase facilities, and (2) support from the Deep Space Network (DSN). The near-earth phase facilities have TDS responsibility from launch up to that portion of the trajectory wherein the DSN has continuous visi- bility of the spacecraft, at which time the DSN assumes responsibility for the TDS function. The near-earth phase facilities are composed of selected stations from the NASA MSFN and STADAN Networks and from the Air Force Eastern Test Range (AFETR). The selection of stations from these facilities is a function of the par- ticular trajectory flown and the individual project's data requirements. The selection is individually negotiated by the JPL Cape Kennedy Field Station for each flight project, but in general is composed of both tracking sta- tions (including radars) and telemetry reception stations. The near-earth phase facilities may or may not have a requirement to send commands to the spacecraft-de- pending upon the criticality of the flight sequence during the near-earth phase of the particular mission. However, the AFETR at all times retains responsibility for range safety. Data from the near-earth phase stations are fed via a combination of NASCOM and/or AFETR com- munications circuits back to Cape Kennedy, Florida, where flight operations are conducted for the near-earth phase of the mission. During this period, radiometric data are fed to the Cape Kennedy Real-Time Computer Complex, where early trajectory computations are per- formed. Near-earth phase telemetry data are displayed at the Mission Control Center at Cape Kennedy and are also routed to the DSNs Space Flight Operations Facility (SFOF) in Pasadena, California. When the spacecraft reaches the pre-determined point in the trajectory, the TDS responsibility is transferred to the DSN. By this time, mission operations responsibility has been trans- ferred from Cape Kennedy to the SFOF. The other part of the NASA Tracking and Data System is the Deep Space Network. Its function in supporting Helios is to provide near-continuous tracking, telemetry, and command support from initial acquisition through the end of the primary mission. In accomplishing these objectives, the DSN will request and receive TDS sup- port from the stations being implemented in West Ger- many. Also, as noted before, the West Germans have responsibility for mission operations. During Phase I, the period from launch through 2 to 4 weeks following launch, the West German Mission Operations Team will reside in the SFOF. At the conclusion of Phase I, Helios mission operations will be transferred from the SFOF to the German Control Center in Oberpfaffenhofen (near Munich), West Germany. This will initiate Phase 11, which will conclude at the end of the primary mission. During Phase 11, the German Control Center will act as a remote terminal to the SFOF for the conduct of mission operations; however, network operations will continue to be conducted from the SFOF since the DSN retains TDS responsibility through the end of Phase 11. Because of these additional operational interfaces, the DSN has representative membership in the TDS and MA&Q Subgroups (center portion of the lower half of Fig. 1). In addition, the DSN has been assigned the responsibility for training over a dozen West German specialists. These factors explain why the DSN's support to the Helios Project is slightly more complicated or- ganizationally than the support provided to a typical U. S. flight project. However, the organizational structure depicted in Fig. 1 has been functioning very satisfac- torily to date and there is no anticipation of the need for a significant change to occur during the lifetime of the project. A description of the current configuration of the spin- stabilized Helios spacecraft is helpful in understanding the support requirements that the spacecraft design places upon the Deep Space Network. The present Helios spacecraft concept differs only slightly in appearance from one of the two proposed configurations in the Mission Definition Report. The report had assumed that the spacecraft would be launched by an Atlas/Centaur/ TE-364-4 launch vehicle combination. By the time of the second Helios Joint Working Group Meeting (April 1970), it became possible for the Helios Project to consider employing a Titan-IIID/Centaur/TE-364-4 launch ve- hicle combination. Since this was a possibility rather than a firm commitment, the Helios project managers decided that instead of permitting the spacecraft design to in- 21 crease significantly in either size or weight, they would target the Helws mission profile to have a perihelion closer to the sun-i.e., 0.25 AU instead of 0.3 AU. This decision markedly altered the thermal design of the spacecraft (which now must withstand 16 solar constants1 at perihelion instead of the previous 11 solar constants- due to the inverse square law effect), but would not preclude flying the new configuration to the original 0.3-AU perihelion in the event it was necessary to return to the AtZas/Centaur/TE-364-4 launch vehicle. As a re- sult, the alternate cylindrical shape for the main body of the spacecraft recommended in the Mission Definition Report was discarded in favor of the spool shape shown in Fig. 2. The conical sections of the spacecraft body in the current design reflect unwanted solar energy away from the spacecraft structure while at the same time provide for a reasonable angle of incidence between the sun rays and the solar cells that are mounted on these conical surfaces. Also, because of the greater solar energy, a fewer number of solar cells were required. 1A solar constant is defined as the average solar energy received at earth distance from the sun, i.e., the solar flux at 1.0 AU. Therefore, the current design contains a mixture solar cells and 50% second surface (i.e., front mirrors to reflect excess solar energy from the conical surfaces. The cylindrical midsection, containing no solar cells but all of the spacecraft electronics and the bulk of the onboard scientific experiments, remains basically unchanged from that depicted in the Mission Definition Report. The maximum spacecraft design weight of 260 kg (570 lb) is compatible with either a Titan.or Atlas for the first stage of the launch vehicle combination. The overall clearance dimensions of the spool-shape space- craft configuration can be accommodated by using either of the more recently designed Viking-Mars or Intelsat N shroud systems (depending upon launch vehicle selec- tion) rather than an extended version of the S type spacecraft shroud contemplated during the Definition Group's studies. Therefore, the present Helios spacecraft design configuration meets the objectives set forth in the Mission Definition Report when using the Ath/Centaur, while at the same time permits trajectories to be flown closer to the sun-i.e., 0.25 AU and possibly 0.2 AU-when launched aboard a TitanKentaur launch vehicle combination. For the present, planning is pro- EXPERIMENT - ceeding on the assumption that the Titun/Centuur launch vehicle will be used; however, these plans are constrained to always maintain compatibility to use the Atlas/Cen.taur vehicle. As might be expected, these launch vehicle con- straints impact the thermal and structural design of the spacecraft much more s i d c a n t l y than they do the telecommunications system design. ysre Following injection of the spacecraft into its helio- centric orbit, and DSN acquisition, the spacecraft is positioned by command such that its spin axis (Fig. 2) is pointing to the pole of the ecliptic. This maneuver pro- vides proper illumination of the solar cells and positions the spacecraft’s antennas such that their patterns will intercept the earth throughout the trajectory of the spacecraft. There are three parts to the Helios spacecraft antenna system: (1) a two-antenna, quasi-omni-directional system; (2) a medium-gain antenna system whose “pan- cake” pattern lies in the plane of the ecliptic; and (3) a high-gain or “spot-beam’’ antenna that is mechanically oriented to point directly at the earth throughout the spacecraft’s heliocentric orbit. These antennas are indi- vidually described below. The omni-directional antenna system is composed of a linearly polarized dipole antenna atop the central mast (Fig. 2), and a circularly polarized horn antenna pointing downward and mounted inside of the bottom solar cell conical skirt. These two antennas are hard-wired to a fixed ratio power splitter, thence to one of the two onboard transponders (Fig. 3). The division ratio of the power splitter is adjusted such that the combination will provide a nearly uniform combined antenna pattern almost completely surrounding the spacecraft. The omni- antenna system finds its primary use during the period from shroud ejection following launch up through the completion of the orientation maneuver that positions the spin axis of the spacecraft toward the pole of the ecliptic. Since the latter maneuver is completed by the time the spacecraft reaches a distance from earth equivalent to the moon’s orbit, the omni-directional an- tenna system might be considered the “near-earth” antenna system. However, this is not to imply that the omni-directional antenna system will not be used during later phases of the mission when signal margins permit. The medium-gain antenna consists of a single longi- tudinal helix antenna mounted on the mast beneath the omni-antenna but above the high-gain antenna. This antenna produces a linearly polarized, pancake-like pat- tern which, following the above-mentioned maneuver, directs its maximum radiation in all directions within the plane of the ecliptic while minimizing its radiation toward the poles of the ecliptic. It is approximately 15 deg wide in the direction perpendicular to the ecliptic and has a downlink gain of 8 dB. It is the prior knowl- edge of this pattern that enables the h a l positioning of the spacecraft spin axis toward the pole of the ecliptic. The medium-gain antenna is connected thraigh a second diplexer to the second transponder aboard the space- craft. However, the transmitter portion of the second transponder is shared between the medium-gain antenna and the high-gain antenna described below. The Helios spacecraft high-gain antenna is a cylindri- cal, parabolic-shaped reflector that is mechanically de- spun to counter the spacecraft body’s spin-rate of 60 rpm. The de-spin angular velocity is adjusted to exactly counter-match the spacecraft body’s spin velocity in order to achieve a fixed beam direction in space. The direction that the spot-beam points within the plane of the ecliptic is a phase angle adjustment with respect to the pulses received from the onboard sun sensor that clocks the speed of the spool-shaped main body of the spacecraft. Since the angle between the sun and the earth changes as the spacecraft traverses its elliptical helio- centric orbit, the phase angle for the high-gain antenna must be updated by earth command during the mission. The high-gain antenna beam is again linearly polarized with a beamwidth of 5% deg within the plane of the ecliptic, and 14 deg normal to the plane of the ecliptic. The downlink gain of this antenna is 23 dB. It is used to transmit high-rate telemetry from the spacecraft to the earth, and operates in conjunction with the space- craft receiving on the medium-gain antenna, using the second transponder mentioned above. Since the second transponder contains a turnaround ranging loop, it is possible to employ either the DSN lunar or planetary ranging system in conjunction with the spacecraft medium-gain antenna system. However, at perihelion dis- tances and beyond, the spacecraft’s radio system config- uration will be such that it will receive ranging signals via the medium-gain antenna and return them to earth via the high-gain antenna. The foregoing Helios spacecraft antennas are con- nected to the onboard radio system, as depicted in Fig. 3. As mentioned previo , the spacecraft contains two complete transponders. pite the fact that these transponders operate on the same frequencies (both up- link and downlink), they are not completely redundant TOP OMNI- ;El!? ++ i MEDIUM- GAIN ANTENNA (PANCAKE) I HI GH-GAI N ANTENNA (SPOT BEAM) BOTTOM OMNI-ANTENNA (HORN) SPIN AXIS I. TURNAROUND COMMAND DETECTOR 2 Q n AUXILIARY OSCILLATOR LJ AMPLIFIER 1 (1 W) CONVOLUTIONALLY CODED TELEMETRY sysae in that the transponders are partially hard-wired and partially switchable. To explain the degree of redun- dancy provided by the present Helios spacecraft radio system design, it is best to describe the signal flow under various mission conditions. From launch through the completion of the maneuver that orients the spin axis to the pole of the ecliptic, the spacecraft is operating with the omni-directional antenna system and with transponder 1. Prior to separation of the spacecraft from the TE-364-4 third-stage, uplink and downlink communication is via the top omni-antenna (since the bottom omni-horn-antenna is masked by the third-stage rocket motor). At separation, the spacecraft is spinning with the spin axis oriented along the velocity vector of the outgoing trajectory, which lies in the plane of the ecliptic. Since, under these circumstances, the medium-gain antenna pattern intercepts but a small seg- ment of the earth, the omni-antenna system is used for initial DSN acquisition. The two-antenna omni-system provides nearly spherical antenna pattern coverage ex- cept for a null directly ahead of the spacecraft. Since the “look-angle” from the spacecraft to an earth station will traverse this forward null very rapidly as the space- craft departs the vicinity of the earth, the existence of this forward null does not impose a serious initial acqui- sition constraint. Following separation, onboard sun sensors cause the spacecraft’s attitude control system to 24 the reliability of the switching circuits necessary to utilize the redundant component. Since these studies are still in process, the Helios Project spacecraft radio system depicted in Fig. 3 is subject to change with time. Nevertheless, an understanding of the basic design philosophy of the Helios spacecraft radio system as presently conceived is helpful in assessing its operational impact upon the DSN. For instance, it is interesting to note that the redundant components associated with the uplink from the DSN to the spacecraft have separate, hard-wired paths, while the redundant components in that portion of the transponder associated with the down- link are connected to switching matrices which permit the use of a number of alternate paths or operating modes in the event of a failure along one of the down- link paths. perform the first of two orientation maneuvers-namely, to position the spacecraft’s spin axis such that the space- craft’s solar cells receive maximum energy from the sun. During this (step I) maneuver, the spacecraft’s spin axis remains basically in the plane of the ecliptic. Since, depending upon the trajectory, the initial DSN acquisi- tion of the injected spacecraft can occur prior to, during, or subsequent to the completion of the step I maneuver, a nearly uniform spacecraft omni-directional system is needed. Following completion of the step I maneuver, the spacecraft’s attitude remains fixed while the Helios Mission Operations Team in the SFQF evaluates the con- dition of the various onboard systems, using spacecraft telemetry. Following this, selected scientific experiments aboard the spacecraft are activated in the near-eai-th science mode. The near-earth science phase is complete by the time the spacecraft reaches lunar distance from the earth. At this time, the science instruments are deactivated and the spacecraft is readied for the second orientation maneuver (step 11). During the step I1 maneuver, the spin axis is commanded to precess in a direction that will orient it to the pole of the ecliptic. As this is done, the medium-gain antenna’s pancake-like pattern will slowly begin to intercept the earth. At this time, the spacecraft is commanded to operate with trans- ponder 2 in the diplex mode via the medium-gain antenna. Using the received signal strength at the DSS as an indicator, the spacecraft is commanded to continue precessing its spin axis until the pancake antenna pattern causes a maximum signal strength indication to be achieved on earth. The maximum is determined by causing the spacecraft to precess beyond its optimum orientation and then return to the optimum position- at which time the spin axis will be oriented toward the pole of the ecliptic. Following completion of the step I1 maneuver, the de-spin velocity and phase angle of the high-gain antenna are adjusted by command to direct the “spot beam” toward earth. Since the spinning space- craft will be self-stabilized (gyroscopically) in this position (unless unexpectedly perturbed), no further orientation maneuvers should be necessary. All ten on- board scientific instruments are then activated and com- manded to return data via the high-gain antenna at the maximum information rate permitted by the telecom- munications system capability through the remainder of the mission. The foregoing orientation maneuver con- siderations explain in part the rationale behind the radio system block diagram shown in Fig. 3. Other aspects will be discussed in the paragraphs that follow. the Go command detectors are different. These two features ensure that commands will be executed only via the pre-selected channel. The last major distinction The benefits achieved by having redundant compo- nents aboard the spacecraft must be traded off against Let us consider first the uplink portion of the space- craft radio system. As mentioned previously, both trans- ponder loops operate on the same center frequency. Also, both receiver 1 and receiver 2 are continuously energized in order to ensure an uplink command capa- bility to the spacecraft. Since these receivers are hard- wired to separate antennas, the selection of the uplink transponder path is determined by spacecraft logic cir- cuits that select the stronger of the two signals from the receiver automatic gain control values. Prior to the completion of the step I1 orientation maneuver, it is anticipated that the stronger signal will be via the omni- directional antenna and receiver 1 since the medium- gain antenna pattern will not have yet been oriented toward earth. Following the step I1 maneuver, the reverse should be true since the medium-gain antenna has an uplink advantage of approximately 6 dB over the omni-directional system. The transition characteristics between these two antennaheceiver paths are dependent upon both the actual trajectory being flown and the sequences employed in the orientation maneuvers. For these reasons, it is quite possible that future studies will show the need for some kind of mode control within the spacecraft receiver loops during the orientation maneuvers in order to prevent the interruption of two- way lock due to unwanted antenna switching. However, with respect to commands, the following precautions have already been taken: (1) The subcarrier frequency for command detector 2 has been made intentionally different from that for command detector 1; and (2) since the command bit rates for the two channels are sub- multiples of their subcarrier frequency, the bit rates to between the two uplink channels aboard the spacecraft is that the turnaround ranging loop components are connected only to receiver 2. This is because uplink signal-to-noise ratio considerations via the omni-antenna system very rapidly diminish the usefulness of that path once the spacecraft has been injected into heliocentric orbit. Since the Helios spacecraft reaches a distance from earth equal to the moon’s orbit approximately 9 h after launch, it was felt that the additional complexity and weight associated with a ranging loop via the omni- antenna was not justified. Next, let us consider the downlink portion of the Helios spacecraft radio system. In contrast to the uplink portion of the system, the active downlink path through the radio system is under ground-commandable mode-control. As implied by Fig. 3, a considerable number of paths or modes are possible in the downlink portion of the radio system. For the moment, only the more significant modes or combinations will be discussed. First, the downlink portion of the radio system may be operated in either the non-coherent or the coherent mode in association with one of the uplink receivers. The non-coherent mode utilizes onboard auxiliary crystal oscillators to derive the transmitter’s frequency. This mode is used principally prior to the initial DSN acquisition and again during solar occultation periods. The coherent mode employs the uplink frequency (as received aboard the spacecraft) multiplied by the ratio of 240/221 in order to translate it to the nominal downlink frequency. The coherent mode is required for two-way doppler tracking and for ranging. The coherent mode is the primary mode of operation during the cruise phase of the mission. Regardless of whether the non-coherent or coherent mode is selected, only one exciter/modulator is activated at one time. The active exciter/modulator feeds one of four possible amplifier circuits, only one of which is active at a given time. The four power amplifier circuits are composed of redundant modules of two classes of amplifiers: (1) 1-W solid-state amplifiers whose primary service is during the near-earth phase of the mission, and (2) 20-W traveling- wave-tube amplifiers used during the main phase of the mission. Following the spacecraft power amplifiers is a switching matrix circuit which will permit the downlink to be sent to earth via any one of the three antenna systems described in Section I l l , Part B . Modulated onto this downlink carrier signal is the spacecraft‘s single- channel, convolutionally coded telemetry-which con- tains a preprogrammed commutation of both science and engineering data. Also on the downlink carrier signal is the turnaround ranging modulation from receiver 2, which is sent to exciter/modulator 2 to be returned to earth via either the medium-gain or the high-gain antenna system. This completes, for the moment, the basic func- tional description of the Helios spacecraft radio system. This article has attempted to provide the reader with a basic understanding of the international character of the Helios project-how it became established and how it is organized. In addition, the reader has been given an overview of the spacecraft configuration and a brief description of its onboard radio system. The next article will discuss the launch windows and probable launch dates for the two proposed Helios spacecraft missions, the nominal post-injection trajectories and attitude maneuvers of the spacecraft as they relate to the support to be provided by the DSN, and a discussion of the DSN tracking load imposed by support to other flight projects during the Helios time-period. These two articles will then form the baseline for detailed descriptions of the various subsystems within the Helios spacecraft radio system and their interfaces with the DSN, starting with the third article. However, since the Helios spacecraft configuration and mission profile design are subject to evolutionary improvement between the present time and the launch of the first spacecraft in the summer of 1974, future articles will also include any evolutionary design changes that affect the spacecraft/DSN tele- communications link, or the interface between the DSN and the Helios Project’s Mission Operations Team. These documents are on file in The Helios Project Office, Goddard Space Flight Center, Greenbelt, Md., and in The Helios Project Office, Gesellschaft fuer Weltraumforschung, mb , Bonn-Bad Godesberg, West Germany. Helios Program Mission Definition Report. National Aeronautics and Space Administration-Der Bundesminister fuer Bildung und Wissenshaft, Washing- ton, D.C. (Bonn-Bad Godesberg, West Germany), Apr. 1969. Project Plan for Helios A and B. Goddard Space Flight Center, Greenbelt, Md., Nov. 1970. Helios Spacecraft Requirements Document, GfW TN 70/14. Gesellschaft fuer Weltraumforschung, mbH, Bonn-Bad Godesberg, West Germany, Jul. 1970. Preliminu y Support Instrumentation Requirements Document. Gesellschaft fuer Weltraumforschung, mbH, Bonn-Bad Godesberg, West Germany, Sep. 1970. Preliminu y Experiments Command and Data Handling Requirements Document. Gesellschaft fuer Weltraumforschung, mbH, Bonn-Bad Godesberg, West Germany, May 1970. Project Helios Minutes for First Joint Working Group Meeting at Bonn-Bad Godesberg, Sep. 2-3-26, 1969. Gesellschaft fuer Weltraumforschung, mbH, Bonn-Bad Godesberg, West Germany. Project Helios Minutes for Second Joint Working Group Meeting at Goddard Space Flight Center, Greenbelt, Ma yland, Apr. 27-30, 1970. Goddard Space Flight Center, Greenbelt, Md. Project Helios Minutes for Third Joint Working Group Meeting at Bonn-Bad Goahberg, West Germany, Oct. 5-8, 1970. Gesellschaft fuer Weltraumfors- chung, mbH, Bonn-Bad Godesberg, West Germany. 7 D. J . Mudgway Mission Support Office This article discusses the capabilities of the DSN as a significant factor in the radio frequency and data management design and the engineering requirements of the two orbiters and two landers for the Viking missions. Also described is the DSN involuement in the extremely complex lander acquisition sequence in which trade-offs are made between the total lander “on” period of 2 h and competing factors of round-trip light time and telemetry and command lock-up times. es The Viking 1975 mission to Mars uses two spacecraft, each consisting of an orbiter and a lander, launched up to 10 days apart and arriving at the planet up to 30 days apart about a year later. On arrival, each spacecraft is placed in orbit, and the scientific instruments on board are used to obtain data to aid in the selection of a suit- able landing site. The lander is then separated from the spacecraft and injected into a suitable landing trajectory from which the complex landing sequence depicted in Fig. 1 can be carried out. Each orbiter will then act as a relay station between either of the landers and earth, receiving data from each lander on a UHF relay circuit, recording and retransmitting the same data on an S-band link to the DSSs. A lower rate, direct S-band link from the landers to the DSS can also be used during orbital operations when desired. However, each lander can support this latter link only for 2 h in each 24-h period. The stringent requirements for planetary quarantine which apply to the landers preclude the replacement of lander elements, such as the radio system, after the lander capsule has been sealed prior to launch. This does not apply to the orbiter, however, and a spare orbiter radio system is kept available in case either orbiter should develop a fault in this area. re From the foregoing brief summary of the Viking 1975 mission, it is apparent that frequency assignments for S-band communications with the DSN must satisfy the following basic requirements : (1) Two 2-way channels, one for each of the two orbiters, both to be operated simultaneously from any one DSS. (2) One 2-way channel, to be shared between the two landers. Each lander uses it only during the daily 2-h period during which it is in direct contact with earth. (3) One 2-way channel, to be assigned to the spare orbiter radio system. This could be used on either orbiter as required. SPACECRAFT I N MARS ORBIT BlOSHlELD CAP SEPARATION LANDER SEPARATION PARACHUTE DEPLOYMENT .=- In making these four frequency assignments, many other factors were also involved. These included the capability of the DSSs to support simultaneous multiple up and down links, bandwidth requirements for ranging, command, and telemetry, doppler offsets, and existing channels already assigned to current flight projects. The 64-m-antenna DSSs will have the capability to support two simultaneous uplinks and four simultaneous downlinks. However, the high-power Klystron amplifiers used for transmitting the uplink are known to produce intermodulation products under certain conditions of nonlinear operation. Therefore, it was necessary to select frequencies for which these undesirable products would be minimized, and would not appear in the band pass of adjoining receive channels. In the case of the channels selected, there are no first-degree intermodulation prod- ucts of order less than 700 lying within the chosen receive band. Excluding the doppler effects, the R F bandwidths re- quired on each channel are: Ranging 3.6 MHz Telemetry (high rate) 4.1 MHz Command 4.0 kHz Therefore, to prevent spectral components falling outside the designated space bands, the lowest uplink transmit frequency must be at least 1.8 MHz from the lower transmit band edge (2110 MHz) and the highest down- link receive frequency must be 2.05 MHz from the higher band edge (2300 MHz). Therefore, the frequencies chosen must lie between channels 9 and 21. The frequency offset due to two-way doppler effects can become appreciable in missions of this type, and . BI must also be taken into consideration in assigning fre- quencies to avoid interference between adjacent channels. A two-way doppler plot for a typical Viking trajectory is shown in Fig. 2 for the period from launch to en- counter. A cyclic variation of 2 8 . 0 kHz due to earth rotation is superimposed on the steady-state offset throughout the mission. -350 -300 -250 It can be seen from these plots that, during the cruise periods of the mission, the two-way doppler offset may exceed 300 kHz. Furthermore, the specified tuning range of a DSIF receiver is 9 parts in lo5 (which is equivalent to about 207 kHz), and the bandwidth of adjacent deep space frequency channels is approximately A185 kHz. I I I I I I I I I i I I I I I I - - To all of these effects must be added an uncertainty in the actual orbiter or lander carrier frequency of +3 x 10-5. Finally, although the space band contains some 29 channels spaced approximately 270 kHz apart, these are not all available to a new flight project. Channels 14, 17, and 23 are assigned to Mariner Mars 1971, channels 15 and 21 to the Helios Project, and channels 6 and 7 to Pioneer. , The DSN has taken all these factors into consideration in making the tentative frequency assignments in Table 1. for Viking spQeec~Qf~ Channel 1 OA 1 OB 13A 13B 16A 16B 19A 198 Frequency, MHz 2293.51 8519 21 11.948303 2294.629630 2112.971451 2295.740741 2 1 13.994599 2296.051852 2115.017747 ler Recommended use Orbiter 1 Orbiter spare landers 1 and 2 Orbiter 2 Typical one-way doppler frequency shift due to orbital motion for the orbiter is shown in Fig. 3, as a function of time from periapsis. The corresponding doppler rates are shown in Fig. 4. LAUNCH 8/1/75 ARRIVAL 6/6/76 P TWO-WAY DOPPLER 2( 16 li E N I Y 4 CL w A p. E ( g -4 G ? -E -12 -16 - 2c +TO EARTH 24 TIME FROM PERIAPSIS, h TIME FROM PERIAPSIS, h line B In order to continuously maintain a coherent two-way link with the DSS, the orbiter transponder must be able to accommodate doppler excursions of about -40 kHz peak-to-peak and doppler rates of about 20 the ground receiver at the DSS must accommodate ap- proximately twice these amounts. In both the uplink and downlink doppler, an additional allowance must be made for an earth-Mars separation rate equivalent to about -70 kHz at the beginning of orbital operations, changing to about -1-20 kHz at the end of the mission. Although the ground receiver can accommodate a tun- ing range of 207 kHz at S-band, with RF tracking rates in excess of 50 Hz/s, it is imperative that the ability of the Viking orbiter to track under these conditions be carefully examined, and a determination made as to the need for a programmed local oscillator on the DSS ground transmitter. The doppler offset seen by the Viking lander due to Mars rotation will exceed &1.7 kHz, together with +3.28 kHz due to earth rotation and the effect due to earth-Mars separation. In this case, the rate of change of doppler is much lower and unlikely to exceed the lander RF loop capability. The capabilities of orbiter, lander, and DSS to track the orbital doppler are dependent on the signal-to-noise ratio and allowable phase error. Careful analysis of the trade-offs must be made to ensure the required capa- bility of the telecommunication link during orbital operations. As mentioned previously, each lander is constrained by power and thermal considerations to a daily “on” period of about 2 h. With an earth-Mars round-trip light time of 40 min;it becomes apparent that a carefully planned lander acquisition sequence is necessary if the maximum time for ground commanding and telemetry data transmission is to be realized. The lander transmitter is turned on at a specified time each day by a beacon signal from the orbiter passing overhead, and is turned off at the expiration of the 2-h transmitting period. The lander receiver is turned on about 1 h prior to the transmitter. The lander acquisition procedure takes advantage of this information to turn on the ground transmitter about 1 h prior to the expected lander on period, sweeping the ground carrier right across the lander receiver pass- band and returning to the expected best lock frequency of the lander receiver. Thus, when the orbiter beacon activates the lander transmitter, it should come on in the two-way coherent mode. On reaching earth, the DSS receiver, subcarrier de- modulator, and bit synchronizer must be locked up, after which a final adjustment to the telemetered static phase error (SPE) can be made prior to starting any command sequences. Estimates for the time required to accomplish these functions vary from 1 to 5 min, although it is expected that the time will diminish as proficiency improves and uncertainties in the lander carrier and subcarrier fre- quencies are reduced. The sequence of events described above is shown in Fig. 5. RECEIVER LOCKED ONE-WAY LANDER TIME TIME, h DEEP SPACE STATION TIME RECEIVER AT ZERO SPE, COMMAND DETECTOR IN LOCK, BEACON TURNS TRANSMITTER ON, IDLE SEQUENCE STARTS IDLE SEQUENCE ENDS, SEND SCIENCE DATA RECEIVE FIRST COMMANDS, CONTINUE TO SEND DATA I END DATA, TIMER TURNS TRANSMITTER OFF I I LOSE RECEIVER LOCK, END DATA ADJUST SPE IF NECESSARY, START COMMANDING, CONTINUE TO RECEIVE DATA I BDA LOCKED UP, RECEIVE SCIENCE DATA RECEIVER LOCKED TWO-WAY, SDA IN LOCK, RECEIVER IDLE SEQUENCE I SET TRANSMITTER FREQUENCY TO ZERO SPE, COMMAND MODULATOR O N TRANSMITTER ON, BEGIN FREQUENCY SWEEP SPE = STATIC PHASE ERROR SDA = SUBCARRIER DEMODULATOR ASSEMBLY BDA = BLOCK DECODER ASSEMBLY In order to avoid loss of valuable science data during these lock-up activities, it has been proposed that the lander might transmit a few minutes of idle-sequence bit prior to sending the first science data. A procedure such as this would permit early lock-up of the DSS telemetry equipment in preparation for the arrival of the science data at the conclusion of the idle sequence. R. 6. Hartley DSN Engineering a n d Operations Office The support provided by the Deep Space Network (DSN) to the Manned Space Flight Network ( M S F N ) during the Apollo 14 mission is described. Support was provided from the three 26-m (85-ft) DSN/MSFN Wing stations, the Goldstone 64-m (210-ft) antenna, the Ground Communications Facility, and the Space Flight Operations Facility. Premission and mission activities are discussed and a brief mission description is included. 1. The DSN support provided to the MSFN for the past Apollo lunar missions has been described in Ref. 1 and earlier issues of the Space Programs Summary Volume I1 series. This article describes the support provided for the Apollo 14 (AS-509) mission, the third successful manned lunar landing. Apollo 14 was the seventh manned Apollo mission flown above the three-stage Saturn V launch vehicle and carried astronauts Alan B. Shepard, Jr. (Commander), Stuart Allen Roosa (Command Module Pilot), and Edgar Dean Mitchell (Lunar Module Pilot). The mission goal was to land in the Fra Mauro uplands, the same goal that the abortive Apollo 13 mission was unable to achieve. The Apollo 14 mission successfully accomplished its objectives in spite of a large number of minor prob- lems, which are mentioned in the following discussion. Launch from Cape Kennedy Pad 39A occurred at 21:03:02.9 GMT on January 31,1971, at a launch azimuth of 75.56 deg. The launch had originally been scheduled for a 20:23 GMT launch, but a launch hold of some 40 min was required due to the new weather restrictions developed after the lightning strikes of Apollo 12. This was the first time a manned Apollo mission was not launched on time. Injection into translunar trajectory occurred midway through the second revolution in the Earth parking orbit with a 5-min 49-s burn of the S-IVB- stage engine. Following translunar injection (TLI), the Command/ Service Module (CSM) separated from the booster and attempted to dock with the unattended Lunar Module (LM). For the first time on Apollo, this docking proce- 33 dure was unsuccessful, with the capture latches of the docking probe failing to seize the Lunar Module drogue. Without a successful docking there would be no lunar landing, although the crew was in no danger. The dock- ing had to be accomplished within several hours before the S-IVB attitude-control system became inoperative and the S-IVB and Lunar Module started to tumble. Under direction from Mission Control Center in Houston, the astronauts made five additional docking attempts in various configurations. On the sixth attempt, docking was successful. After docking, the probe was removed from the Command Module/Lunar Module tunnel for examina- tion. No abnormal behavior or contamination could be detected, and the mission was given the “go-ahead” for lunar landing. After separation from the spacecraft, the S-IVB (the launch vehicle third stage) was directed toward a crash on the Moon as a calibration of the seismometer left there during the Apollo 12 mission. The impact occurred at 07:40:55 GMT on February 4, 1971, at lunar coordinates 8 deg 03 min S, 26 deg 03 min W, approximately 174 km southwest of the seismometer which showed vibrations for some 2 h. Midcourse correction 1 was deleted due to the accu- racy of the translunar injection maneuver. Midcourse correction 2 was executed at 03:39 GMT on February 2, 1971, removing the spacecraft from its “free-return” tra- jectory to save fuel for the lunar landing sequence. The remainder of the translunar cruise was nominal except for an observed dip from 37.0 to 36.7 V in one of the Lunar Module ascent stage batteries. This dip, first noticed during the Lunar Module checkout at 10:56 GMT on February 3, caused some concern and a test was executed at 02:38 GMT, February 4, which confirmed that the battery would properly support a load. Other minor problems during translunar coast included difficulty in maintaining correct pointing of the space- craft high-gain antenna and a high rate of pressure decrease during a cabin leak check. The latter problem was caused by a urine vent valve being accidently left open. Midcourse correction 3 was deleted but a small midcourse correction 4 was executed at 02:Ol GMT, February 4, with the burn lasting less than 1 s. A successful lunar orbit insertion (LQI) burn put the spacecraft in a 315- by 110-km (169.6- by 58.9-nmi) elliptical orbit. Two orbits later a descent orbit insertion (DQI) burn changed these dimensions to 110 by 17.2 km (59.0 by 9.3 nmi). During lunar orbit, additional pointing problems with the high-gain Command/Service Module antenna were experienced. During lunar orbit 12, the Command/Service Module and Lunar Module separated with astronauts Shepard and Mitchell in the Lunar Module preparing for descent to the lunar surface on orbit 14. Difficulty was experi- enced maintaining lock on the Lunar Module high-gain antenna, but behavior was normal on the next orbit. An abort pushbutton sent erroneous inputs to the Lunar Module guidance computer on four occasions. The button may have been defective or contaminated since tapping on the switch panel cleared up the problem. The com- puter was reprogrammed to ignore these abort inputs. During the descent phase, the landing radar came on in the “near” mode [lo668 m (3500 ft) maximum] instead of the “far” mode. This problem was first noticed when the radar failed to acquire lock when passing through 9.15 km (30,000 ft). The radar was turned off and on, and lock was acquired. The landing was normal and was within 20 m of the intended landing site. After more than 5 h of prepara- tion, the two astronauts began their first excursion during which surface samples were collected, the ALSEPl and TV camera were deployed, and the “‘thumper” was acti- vated. The “thumper” is a hand-held rod which taps the lunar surface as a small explosive charge within it is detonated. This device proved to be especially unreli- able, and the astronaut was able to detonate only 13 of the 21 charges. Due to this problem and several others including difficulty unfolding the S-band erectable an- tenna, a twisted urine tube in the Commander’s suit, a broken wrist cable in the Lunar Module Pilot’s suit, and an EVA (extra-vehicular activity) radio problem, the astronauts fell behind their planned timeline and were unable to complete all assigned tasks. During the ensuing rest period, an S-band bistatic ra- dar experiment was conducted between the orbiting Command/Service Module and DSS 14. This experiment is discussed below. Also during the rest period, the Lunar Module astronauts conducted a pressure check of the space suits to confirm that an abnormally high leak rate on the Lunar Module Pilot’s suit had not increased to the danger level. All was in order and the astronauts be- gan the second excursion 2 h and 28 min ahead of sched- ule. This excursion was to take the astronauts 1000 m to the edge of Rim crater. Although this goal was not 1ApoZZo lunar surface experiments package. quite reached, many rock samples of new and different types were collected. , Lunar liftoff, rendezvous, and docking were normal with none of the problems of the earlier docking, Fol- lowing transfer of the lunar samples to the Command/ Service Module, the EM was jettisoned and directed to- ward a crash on the Moon near the ALSEP seismometer. One orbit later, on orbit number 35, the crew executed the trans-Earth injection burn to start the return trip to Earth. Apollo 14 landed at approximately 172 deg W, 27 deg S, midway between Samoa and New Zealand. Table 1 shows the mission event times and Table 2 gives the television coverage. Care must be exercised when using ground elapsed time (GET), which on pre- vious missions recorded the time from launch. With Apollo 14s launch slip (the first on Apollo), the clock car- ries a different meaning. In order to make the sun angles upon lunar landing identical to the premission plans, the translunar trajectory was made 40 min faster than planned. The GET clock was then set forward 40 min and 3 s at 03:38:03 GMT on February 3. A new clock, actual elapsed time (BET), records the time from launch, and the GET clock is merely a convenient reference time which corresponds very closely to the planned mission timeline. First midcourse Second midcourse Begin bistatic fre- quency measure- End bistatic fre- quency measure- Begin LM inspection Third midcourse Fourth midcourse Second LM inspec- Descent orbit Ground elapsed time, h:min:s oo:oo:oo 02:28:30 02:34:19 Deleted 30:36:07 52:ZO:OO 52:25:00 55 :15 :03 60:59:00 Deleted 77:38 :13 78:14:58 82:23:03 82:36:43 83:17:55 86:50:55 Actual elapsed time, h:min:s oo:oo:oo 02:28:30 02:34:19 30:36:07 52:ZO:OO 52:25:00 54:35:00 60:18:57 76:58:10 77:34:55 8 1 :43:00 81 :56:40 8 2,37:52 86:10:52 Greenwich Mean Time, h:min:s Jan 31 121 :03:03 Jan 31/23:31:33 Jan 31 /23:37:22 Feb 02/03:39:10 Feb 03/01 :23:03 Feb 03/01 :28:03 Feb 03/03:38:03 Feb 03/09:22:00 Feb 04/02:01:13 Feb 04/02:37:58 Feb 04/06:46:03 Feb 04/06:59:43 Feb 04/07:40:55 Feb04/11:13:55 Event Undock CSM circularization Powered descent initiation Touchdown Begin EVA-1 ALSEP activated ALSEP high-bit rate on ALSEP high-bit rate off End EVA-1 Begin bistatic radar End bistatic radar Begin EVA-2 End EVA-2 LM ascent CSMlLM docking LM separation LM deorbit burn LM crash Trans-Earth injection Fifth midcourse Splashdown Ground elapsed time, h:min:s 1 04 :28 :03 105:46:48 1 08 :42 :01 108:55:14 114:20:00 1 17:05:00 1 17:36:00 118:14:00 1 19:04:00 130:05:00 131 :04:00 1 3 1 :48 : 14 136:19:00 142:25:43 144:lZ:OO 146:25:00 147:54:00 148:22:25 149:16:04 166:14:00 21 6:42:01 Actual elapsed time, h:min:s 103:48:00 105:06:45 108:01:58 108:15:11 1 1 3:39:57 116:24:57 1 16:55:57 117:33:57 11 8:23:57 129:24:57 130:23:57 131:08:11 135:38:57 141 :45:40 143:31:57 145:44:57 147:13:57 147:42:23 148:36:01 165:33:57 21 6:01:58 Greenwich Mean Time, h:min:s Feb 05/04:51:03 Feb 05/06:09:48 Feb 05/09:05:01 Feb 05/09:18:14 Feb 05/14:43:00 Feb 05/17:28:00 Feb 05/17:59:00 Feb 05/18:37:00 Feb 05/19:27:00 Feb 06/06:28:00 Feb 06/07:27:00 Feb 06/08:11:14 Feb 06/12:42:00 Feb 06/18:48:43 Feb 06/20:35:00 Feb 06/22:48:00 Feb 07/00:17:00 Feb 07/00:45:25 Feb 07/01 :39:04 Feb 0711 8:37:00 Feb 09/21 :05:01 Feb 1 /00:07 Feb 1/00 :09 Feb 3/09:04 Feb 5 1 1 4 6 6 Feb 6/07:34 Feb 6/20:12 Feb 6/20:27 Feb 8/01 :21 Feb 8/23:31 aGDS = Goldstone MSFN station HSU = Honeysuckle MSFN station, Australia MAD = Madrid MSFN station, Spain Inspection of docking probe Interior of spacecraft EVA-1 EVA-2 Rendezvous Docking lnflight demonstrations Stationa GDS GDS HSU HSKIMAD GDSIHSU MAD MAD GDS GDS As was done during previous Apollo lunar missions, DSSs 11,42, and 61 were committed to support Apollo 14 under direct MSFN/MSC control starting at launch minus 2 wk through the end of the mission. The Mars station, DSS 14, was required to receive voice, telemetry, biomedical, and TV and relay the data to the Goldstone Prime MSFN station (6DS). Specific ents existed for lunar lan television, crash. Coverage was also television during translunar coast. As part of a continuing study of lunar potential anoma- lies (mascons), IDS§ 14 was required to provide precision doppler recordings of the Command/Service ing several low lunar orbits and of the during the descent phase and later durin additional internal DSN requirement was levied for more doppler data during all low Command/S orbits and several adjoining high orbits in s L study of mascons (with the same prin tor who originated the formal requirem can see from the timeline (Pig. I), one pass from DSS 62 was necessary to provide almost complete coverage of this requirement. A related requirement was for high- speed strip-chart recordings of DSS 14 received signal level during orbit 3 of the Command/Service Module, descent and touchdown of the Lunar Module, and the crash of the Lunar Module. f On April 14, 1969, Stanford University submitted a formal proposal to NASA to conduct an experiment, Downlink Bistatic Radur Study of the Moon, using DSS 14. The specific goals of this experiment are: (1) To determine the Brewster angle of the lunar crust at S-band. (2) To measure the spectral properties of bistatic radar echoes from low-altitude orbit. (3) To gain operational experience with Apollo systems and operations as an aid in the design of future bistatic radar experiments. The experiment should reveal fundamental new scien- tific information on the upper few centimeters of the lunar crust through a short wavelength determination of rewster angle of the lunar surface, and should pro- ngineering data necessary for optimizing the de- sign of future bistatic radar experim orbits (approximately %Wkm altitud results should provide a lunar S-band, bistatic radar cali- bration which would have considerable utility in the GREENWICH MEAN TIME FEB 4/00 00 06 00 1200 18 00 FEB 5/00 00 06 00 12 00 I I I I I I I b//////////////--/d 14 DSS VIEW PERIODS (m =SCHEDULE) t I I 42 42 I I 62 DSN REQUIREMENT HIGH ORBITS LOW ORBITS HIGH ORBITS -I ------- + ------- EVENTS I OCCULTM LO1 A DO1 S E P A ACIRC ATOUCHDOWN 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 l 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 75 80 85 90 95 100 105 110 11 GROUND ELAPSED TIME, h interpretation of similar experiments conducted in the future at the planets. It was proposed that the downlink CSM S-band signal would be directed toward the lunar surface. At DSS 14 the reflected signal (and a portion of the direct signal) would be received in two polarizations simultaneously using the polarization diversity S-band cone. The outputs of the two receiver chains would be recorded on FR 1400 recorders. An obsolete time sync agreement in effect since ApoZZo 11, which provided for routine measurements with a DSIF portable cesium clock at all Goldstone/ApoZZo sup- port sites, was amended on June 9, 1970. The new agree- ment, which will be valid for all future ApoZZo missions, is as follows: A one pulse-per-second tick is distributed via DSN microwave to DSS 14. The pulse originates in the DSIF Standards Laboratory, arrives at the sites with known offsets, and is accurate to better than 5 ps. This signal is made available to the MSFN at the MSFN microwave interface at DSS 14. The portable cesium clock will be used only to calibrate the micro- wave delays when changes have occurred to the microwave system. Routine support of the portable clock will be discontinued. In the case of a bona fide timing emergency, the portable cesium clock will be available with a delay of 2 h during the normal work- ing day and 48 h otherwise. Following any clock trip a TWX giving the results with respect to the Na- tional Bureau of Standards and the expected National Bureau of Standards-U.S. Naval Observatory offset will be sent out to interested parties. wenty-Kilowatt Uplink Original plans for the operation of the Wing stations were for two simultaneous 10-kW radiated uplink signals. This was accomplished using two 20-kW transmitters fed into a combiner which introduced a 3-dB loss. Thus, 10 kW of power from each transmitter was radiated from the antenna. The other 10 kW from each transmitter, which was lost in the combiner, was dissipated in a 20-kW water load. When transmitting a single uplink, there was no longer a need for the combiner and its ac- companying 3-dB loss, and a nonstandard configuration was entered at some Wing sites which bypassed the com- biner, allowing a single 20-kW radiated power uplink. Unfortunately, since this was not a standard configura- tion, the combiner water load interlock was not disabled; that is, any failure in the water load would shut down the transmitter, even though the water load was not in use. No changes could be authorized because the radia- tion of a single 20-kW uplink had never been requested or committed. 7 On June 12,1970, GSFC levied an official requirement upon the DSN for the 20-kW capability. An Engineering Change Order was issued to disable the water load inter- lock when in the 20-kW configuration. The modifications were completed in November 1970. Certain operational limitations remain: There exists no backup 20-kW capability since the installed microwave switching gear allows only one transmitter (the DSN transmitter) to be switched around the combiner. All uplinks must be turned off before any micro- wave switching can occur. Site ert 0 ALSEP communications are usually accomplished using the 30-Et antennas of the MSFN. In September 1969, the DSN received a requirement for the Wing sites to track ALSEP during the Active Seismic Experiment on ALSEP No. 4 during the Apollo 14 mission period. During this new experiment, the ALSEP transmits at 10.6 kbits/s, a higher rate than past ALSEPs and beyond the capa- bility of the 30-ft stations. The experiment is dubbed the “thumper” because of a series of small charges which are actuated by the astronauts to generate artificial seismic waves. &@ Between Apollos 13 and 14, the DSIF was reorganized to eliminate the separate positions of Operations Engi- neering, Operations Planning, and System Data Analysis (SDA) for Apollo. Instead the DSIF planning efforts are represented by a single DSIF Operations Project Engi- neer for Apollo, and some of the operations functions, such as predict generation, have been assumed by the DSN mission independent operations organization. esti DSSs 11, 42, and 61 were placed under configuration control for the Apollo 14 mission as of 0O:Ol GMT on January 17, 1971, and subsequently placed on mission status by the MSFN as of 0O:Ol GMT on January 21, 1971. Prior to January 21, each Wing site conducted extensive maintenance and testing of the equipment common to the DSN and MSFN. The new requirement for Wing support of the ALSEP indicated a need for ALSEP premission testing. The downlink tests pose no problem, but the ALSEP uplink frequency (2119 MHz) is at the extreme end of the normally used band (MSFN: 2100-2110 MHz; DSN: 2110-2118 MHz). Some tuning of the klystron power amplifier is therefore needed. Unfortunately, the klystrons may be retuned only a finite number of times before the cavity adjust- ment screws will bind up, requiring costly and time- consuming repair. In order to maintain these klystrons in good condition for the mission, the DSN limited the MSFN to one ALSEP uplink test at each Wing site. These tests, conducted on January 11 and 12, failed at all three Wing sites due to identical intersite microwave prob- lems unrelated to the klystrons. It is unfortunate that the results of the first test were not applied to the second and third site. Nevertheless, the MSFN requested addi- tional testing and, after management discussions, one additional test at each site (January 23 and 24) was allowed. Later investigation may show that this testing is related to the transmitter problems experienced during the mission. ss 1 In addition to the bistatic experiment tests and the predict tests mentioned below, DSS 14 conducted two tests before the Apollo 14 mission (see Table 3). The first was an officially required participation in Day 5 of the MSFN network readiness test. DSS 14‘s participation turned out to be nothing more than a data transmission test between the DSS 14 communications center and the Goldstone MSFN station. The second test was an internal DSIF configuration vedication test on January 19. The third test was an extensive data flow test with a space- craft simulator used at DSS 14 to generate data which were fed through the entire system ending at the Gold- stone MSFN station. Due to delays the test required more than the alloted 24 h on January 21, and the tele- vision and Lunar Module tests were rescheduled on January 27. Configuration control was imposed on DSS 14 on January 17. S A meeting was held at JPL on May 8, 1970 to discuss the feasibility of the proposed experiment. In attendance were representatives of MSC, GSFC, NASA Headquar- ters, North American Rockwell, Stanford University, and the DSN, DSIF, and GCF. The potential problem areas were identified and a number of action items were assigned. Shortly afterward a support requirement was issued at MSC (May 19, 1970). The final details on the configuration at DSS 14 were discussed at another meet- Data 12/21/70 12/22/70 12/23/70 01 /08/71 01 /08/71 01/19/71 01/21/71 01 /27/71 01/28/71 STATE VECTOR 7094 I 14 tests MAGNETIC TTY PAPER TAPE PDP-7 (MEDIA TAPE e CONVERSION 8 CENTER) Time, h:min ELECTRICAL INTERFA': 16:OO-24:OO 22:00-02:01 02:Ol-06:OO 22:30-05:30 1420-14:20 14:18-22:17 2O:OO-07:lO COMMUNICATIONS Dss 14 ~ PROCESSOR Test Bistatic cable installation Bistatic equipment checkout Bistatic trock of Pioneer VI Day 5 of network readiness test Bistatic test Configuration verificotion test Data flow test (24 h) Doto flow test Bistatic final Cali bration Comments Receivers 3 and 4 not yet installed no 152.4 cm/s (60 ips) narrow- band frequency modulation FR 1400 modules Data transmission test During Pioneer V I track; used Pioneer VI signal; FR 1400 not available TV and LM tests ing at JPZ on November 9, 1970 with approximately the same representation as the May 8 meeting. A number of installation periods and tests were con- ducted from late December 1970 until launch on January 31. These tests are included in Table 3. Predict data for DSS 14 is generated in two ways. The prime method uses state vectors supplied by Houston MSC or Goddard Space Flight Center, and generates DSN predicts in the SFOF computers. The backup system involves a 29-point acquisition message generated at MSC, transmitted directly via TTY to DSS 14, and converted at DSS 14 from X-Y to HA-dec coordinates with an atmosphere refraction correction added if necessary. In November 1970 the DSIF served notice that it would not allow further use of the 29-point conversion program because the software had never been properly certified or documented, even though it had been suc- cessfully used on several Apollo flights. To make matters worse, the SFOF IBM 7094/7044 computer system was about to be removed to make way for a second IBM 360/75, and it was uncertain if the predict software for the 360 would be ready for Apollo. Several compromises were necessary. The 7044 was removed, but the 7094 was merely moved to a new location in the SFOF. The 7094, however, lacked a high- speed printer (used for checking the predicts), adequate tape drives (to lessen the required tape changes), and a 7044 to transmit the predicts. The missing 7044 was overcome by the procedure shown in Fig. 2. A state vector received in the SFOF was input to the 7094. The output tape was then carried to the PDP-7 computer in the Media Conversion Center of the SFOF where the predicts were transferred to TTY punched paper tape. This tape was taken to a tape reader in the Communications Center, where a T T Y header was added, and the data was transmitted. A test of this sys- tem on January 18 was successful except for a bad TTY punch in the Media Conversion Center. The test was repeated on January 19 with complete success. In an effort to obtain clearance to use the 29-point conversion program, a number of tests were conducted starting in December 1970. The last tests on January 12 and 22 re- sulted in acceptance on an emergency backup, best-effort - basis only. ITY TAPE READER (COMMUNICATIONS CENTER) SFOF N~TWORK ANALYSIS AREA 9 S DSSs 11, 42, and 61 successfully supported all phases of the ApoZZo 14 mission. The problems experienced are noted in Table 4. The many transmitter problems are being investigated at this time. 5ta &in 5s 14 Seven passes were tracked as shown in Table 5. The only schedule change occurred on February 4 when DSS 14 was requested to be on track earlier than the sched- I uled 03:OO GMT in order to observe the second Lunar Module checkout. This second checkout was to investi- gate the Lunar Module battery problem, and it was thought that low power levels might have resulted in a degraded telemetry signal. Accordingly, DSS 14 was tracking at 00:43 GMT on February 4. DSS 14 experi- enced no problems that had any effect on Apollo 14 sup- port. DSS 62 tracked the Command/Service Module as planned on February 4 from 13:32:18 to 22:12:47 GMT. The required doppler data was successfully taken and sent to JPL. redid tions It is apparent that the extraordinary care given to pre- dict generation problem before the mission paid off, because no problems were experienced during the mis- sion (unusual for ApoZZo). The personnel were eagerly cooperative, and it is felt that this spirit is responsible for maintaining and operating the system with so many known weak points. The 29-point acquisition messages were also used on occasion at DSS 14. Several comparisons were made be- tween the two types of predicts with no significant dis- crepancies except for the Lunar Module powered descent. This difference was anticipated since the 7094 predict program cannot model a long duration burn with a changing force vector. Thus, the 29-point message was much more accurate and was used for powered descent. The SFOF areas and equipment used for ApoZZo 14 included the Operations Area, the Network Analysis Area, the displays, the relocated 7094, and the Media Gonver- DSS 1 1 42 - 61 Jan 31/22:31-22:34 Jan 31 /23:53-07:35 Feb 01 /20:43-08:42 Feb 02/21 :10-09:02 Feb 03/21 :17-08:54 Feb 04/21:13-10:05 Feb 05/21:53-10x54 Feb 07/00:01-11:21 Feb 07/23:37-11:32 Feb 08/23:30-12:06 Feb 01 /04:45-1225 Feb 02/05:31-13:18 Feb 03/05:41-13:41 Feb 04/05:42-13:57 Feb 05/06:37-14:45 Feb 06/07:33-1547 Feb 07/08:02-16:ll Feb 08/08:06-16:29 Feb 09/08:24-16:41 Feb 09/17:08-18:48 Feb 09/20:35-20:49 Feb01/11:58-02:00 Feb 02/ 12:33-M:17 Feb 03/12:45-02:26 Feb 04/13:32-03:07 Feb 05/13:50-04:05 Feb 06/ 14:4344:5 1 Feb 07/15:10-04:54 Feb 08/15:28-04:58 Feb 09/17:5619:25 Problem None PA 4 (power amplifier) high-voltage rectifier interlock; no apparent reason. PA 3 arc detector trip; no apparent reason None Maser 1 exhaust valve frozen; will replace maser PA 3 arc detector trip at 2224; switch to PA 4 after 19 seconds PA 4 high-voltage ac avercurrent trip; no apparent reason None None None PA 4 high-voltage ac overcurrent trip; no apparent reason; PA was in standby Both klystron power supplies tripped on beam overvoltage; transmitters were not radiating None None None None None None None None None (ALSEP track) None None None None None PA 2 excessive reflected power trip during pretrack; retuned klystron None None None None 00:30:52-06:30:00 2 2:59:00-04:30:00 00:43:00-09:55:00 21 :07:15-10:57:40 21 :42:34-11:51:40 22:40:00-06:OO:OO sion Center. The SFQF support is limited to predict gen- eration and some off-line monitoring. As mentioned above, the support was excellent. atio The DSN GCF provided voice and teletype circuits as required to support the operations mentioned above. In addition, JPL acts as West Coast Switching Center for the NASA Communications Network and handles many non-DSN circuits in support of Apollo. The only known GCF problem was a bad set of communications pro- cessor log tapes covering parts of the precision doppler data collection. These data were later recovered from DSS 14 and 62 tracking data tapes were sent in by mail. 1. Hartley, R. B., “Apollo Mission Support,” in The Deep Space Network, Space Programs Summary 37-64, Vol. 11, pp. 7-11. Jet Propulsion Laboratory, Pasa- dena, Calif., Aug. 31, 1970. 0. H. von Roos Tracking and Orbit Determination Section Possible influences on the measurements of the total electron content due to mag- netic fields, spatial inhomogeneities, and pulse shape distortions are investigated and found to be negligibly small at Deep Space Network tracking frequencies. 1. There are essentially two methods now in use to mea- sure the effect of the interplanetary plasma on the propa- gation of radio signals used for tracking a spacecraft. One method is the differenced range versus integrated doppler (DRVID) calibration utilizing the difference in phase and group velocity of the electromagnetic waves (Ref. 1). The other method uses two different frequencies (dual fre- quency method) taking advantage of the frequency de- pendence of the phase and group velocity (Ref. 2). The effects of the interplanetary plasma on the electromag- netic wave propagation can be theoretically expressed by the fact that the (relative) dielectric constant E is given by where 4se2 m 04 = - N = plasma frequency o = angular frequency of the radio wave and N = number of electrons per cms The refractive index is then given by I since the plasma frequency is u p z 0.18 N = l O ~ m - ~ and the carrier frequency o is much larger than Op, the former being 400 MRz for the dual frequency experiments and 2000 MHz for the DRVID method. There ensues, then, a first order theory of the effects of the solar plasma on the electromagnetic wave propagation as put forward by Eshleman in comprehensive form (Ref. 3). This first order theory neglects magnetic field influences, influences due to spacial and temporal inhomo- geneities in the electron density 37, and, finally, puke shape distortion due to dispersion. We shall address our- selves to these higher order effects and show that they are indeed negligible at the high carrier frequencies involved. 42 rotation of the polarization is proportional to the &fer- ence of the two wave vectors: e equation for the elec- tric field of a monochromatic electromagnetic wave in the presence of a constant magnetic field in convenient vector notation is given by (6) - c whereas the electron content measurements, as in the dual frequency and DRVID methods, essentially depend on the wave vectors k+ and k- separately, in which case the magnetic field influence is negligibly small. (3) where L ( v -io) (v - i .)2 + 0: A = c=:- v ( v - i o ) + U : v - io (v - L ) 2 + a: and oP = plasma frequency 0 = angular frequency of the wave v = effective collision frequency for the electrons wc = cyclotron frequency h = unit vector in direction of the constant magnetic A field The magnitude of the interplanetary magnetic field is of the order of tesla (W gauss) resulting in a cyclotron frequency of wC z 176 see-I. The collision frequency is v < 1 sec-I. Therefore, for w z 2 X log sec-I, the influence of the magnetic field and the influence of collisions are totally negligible as far as the wave propagation is con- cerned. This is even true for the ionosphere and solar corona where magnetic field strengths are of the order of lo-* tesla (1 gauss) so that wC z lo6 sec-I. For these cases Eq. (3) reduces readily to where the ratio W,/W = 5 X 1Q* indicates the smallness of the magnetic field iduence. The first term on the right- hand side of Eq. (5) leads to an observable Faraday rota- tion effect, however (Ref. 5). The reason for this is that the The field equation (3) reduces, therefore, to Turning now to the spacial inhomogeneities of the elec- tron density, it must be realized that the smallest scale of such fluctuations is at worst of the order of kilometers. Measurable changes rather take place over thousands of kilometers. Therefore, considering the small carrier wave- lengths involved, a first order Wentzel-Kramers-Brillouin method is quite adequate for the solution of Eq. (7). Sup- pose that the electron density varies in the direction of propagation of the signal (the x direction). With the ansatz E = A (x) exp [i (kx - W t ) + i+ (x)] (8) where with N o a reference electron density chosen such as to make the difference IV (x) - N o small, we obtain an expres- sion for the correction in amplitude A (x) and phase 4 (x) from Eq. (7): 2i (k + 4’) A‘ + (@” - 2k 4’) A We neglected the second derivative of the amplitude A” because of its smallness (see, for instance, Ref. 6). Setting we obtain from Eq. (10) and obtain in complete analogy to Eq. (11) Noting that +’ is of the order of + / L where L is the scale length of changes in the electron density which measures in kilometers and that on the other hand k-l is of the order of centimeters, we see that Eq. (12) can be replaced by the error being of the order of (+/kL z But Eq. (13) yields as solution where of course the prime means differentiation with respect to z. The amplitude turns out to be We assume for convenience that N ( z ) > No. It is clear then that a surface of constant phase, the wave front, is given by Eqs. (17) and (18): so that disregarding an immaterial integration constant. From Eq. (8) we see immediately that there is no effect in first order. In second order we obtain from Eq. (12) i A‘ A 2k - -i+’ + - [2 (+ ’ )Z + i+”] -- which when integrated yields The first term under the integral sign in Eq. (16) is always imaginary (see Eq. 11) and will, therefore, always lead to a phase shift. If the electron density N ( x ) is always larger than the reference electron density N o throughout the ray path, the signal will always be attenuated. In the opposite case the second term under the integral becomes imaginary and leads to an additional phase shift. However, the effect is extremely small, being of the order X/L z at best. Turning now to a situation in which the electron density inhomogeneities are transverse ( z direction) to the radio beam direction, we put E = A ( z ) exp [ i (kx - at) + i+ ( z ) ] (1-7) [ N (2) - No] [ N ( z ) - N o ] = tan z z a (21) ax 1 dz k -= - - where a is the angle between the direction of propagation and the x coordinate (the undisturbed direction of propa- gation). To roughly estimate the angle a, let us put N ( z ) = N o (1 + z2 /D2) where D, the scale length, is assumed to be lo3 km. For a beamwidth B of 26 m, the the DSN antennas, we obtain from Eq. (21) 1.8 X radians UP B W D a=--= assuming N o = 10 ~ m - ~ and u = 2 X lo9 sec-l. diameter of (22) The apparent change in the range due to such small angles is very small indeed. To estimate its value, it suffices to consider that for a distance of 1 AU, the lateral deviation of the ray path is only about 0.2 km, assum- ing an angle as given by Eq. (22). The change in the range or length of the ray path is then only fractions of a centimeter. Solar plasma clouds originating from solar flares are also of concern because of their vastly larger electron density. Typically, their dimensions are of the order of lo6 km and their electron densities are 10 to 100 times that of the ambient plasma wind. Here we use a ray tracing technique to estimate their possible influence on the deflection of radio beams. A simplsed geometry of a plasma cloud is given in Fig. 1. x = o x = d A radio signal enters a plasma slab, confined between the planes x = 0, and x = d, normally to its surface. Due to a lateral electron density inhomogeneity N ( z ) , the beam will be deflected toward a region of lower electron concentration and will eventually emerge in a different direction characterized by the angle (Y in Fig. 1. The refractive index of the plasma is again given by since the radar frequency .W is so much larger than the plasma frequency 4re2N/m. In order to obtain an expres- sion for the angle (Y and to ultimately evaluate the effect of the bending on the ray path, it is quite sufficient to use Fermat’s principle 8 nds=O (24) s If we express the ray in Fig. 1 by z = z (x), the Euler equa- tion for it becomes very simple: 1 dn z” n dz 1 + ( z ’ ) ~ --= where a prime means differentiation Integrating Eq. (25) once yields (-w with respect to x. no = n at the position x = 0 (see Pig. l), and z’ = dz/dx = 0 at x = 0. It follows from Eq. (26) that is the desired relationship between x and x. Since x is by definition real, n must be greater than no. The beam is indeed deflected into regions of a higher refractive index or a lower electron concentration. According to P. F. MacDoranl the time rate of change of the electron content I of a typical solar plasma cloud can be as much as (28) dI - = 3 X 1015 m-2 sec-I dt Assuming a cloud velocity of 300 km-sec-l and an exten- sion of some lo6 km (values which are generally accepted; see, for instance, Ref. 7), we obtain a characteristic value for the electron density gradient of about dN dz - z 1om-4 (29) To estimate the angle (Y in Fig. 1, we now use Eq. (23) together with a linear relationship N = N o (1 - yz). The integral (27) becomes Defining and putting zo = 0 for convenience, we obtain from Eq. (30) We also have (see Fig. 1): where z is the solution of Eq. (32) with x = d. The solution of Eq. (32) in turn is given by lPrivate communication. With the substitution o0 - o = w', we obtain from Eqs. (38) and, finally, tan a: = sinh (pd) Just to see how small the angle a! is, we take typicaP values for the plasma cloud as given by MacDoran. Let N o = 300 ~ r n - ~ and w = 4n lo9 Hz. From Eq. (29) the value of y turns out to be (36) 1 = 3 lo-? m-l - and (40) (35) Let d = lo9 m, the scale length of a typical plasma cloud, then tan a! = sinh pd = sinh (lo-?) (37) so that a! is rad signifying a lateral deflection of the beam at a distance of 1 AU of about 10 lun, somewhat larger than before (Eq. 22) but still totally negligible. Finally, we address ourselves to the question of pulse degradation by dispersion in the interplanetary plasma. It is clear that a pulse or pulse train may be represented where it is recognized that the first exponential deter- mines the phase velocity and the first term in the second exponential determines the group velocity and the last term represents a distortion of phase and amplitude. For mathematical convenience we take a gaussian pulse by a Fourier integral. Accordingly the electric field of the radio signal is The integral (41) becomes with the pulse spectrum (42) where Wk 20 +(W) =a(. - - - ut (39) i[S (*, - *) 200 - Wet]} which is a solution of Eq. (7) using the approximation The pulse spectrum F (0) is sharply peaked at the carrier frequency o0 with an inverse bandwidth o1 small com- pared to the carrier. For the DSN transmitter ol/oP = 2.5 X surely a small number. Because of these prop- erties of F (0), the phase + (w) can be expanded in a Taylor series about o0 and we have X W$X + ( O ) = (; - t). -E + (; - t + -) 2coc (00 - w) (43) The term E = ~ $ 4 x/2c0: responsible for the pulse degra- dation can quickly be seen to be ridiculously small even for distances x of the order of 1 AU. For o1 z % lo6 sec-l, oo z 2 X lo9 sec-l, oP z 1.8 X 105 sec-I (for No = 10 ~ m - ~ ) , we have E = 1.6 X x where x is measured in kilometers. The influence of dis- persion on the pulse shape is negligible. The situation would change drastically if the carrier frequency would be lowered and the modulation frequency (inverse band- width) would be enlarged. For instance, a twice as high modulation frequency together with half the carrier fre- quency would result in a 4 X 8 = 32-fold enhancement of the pulse shape effect. It must be realized that Eq. (43) is only valid if E < < 1. In that case Eq. (43) may be written as: --:['-t+*]z) 4 c 2cog t + ""1'- 2cw: +)} (44) where of course oto: x %; c e=-- (45) In case of a slowly varying electron density along the ray path, we have I (46) &e' o;x = - m where is the electron content in cm-' since we use cgs units throughout. At the time the pulse arrives at the receiver. In that case, x is the distance between the spacecraft and the earthbound receiver. If the pulse (44) is beat against a reference pulse of the same shape (Eq. 42), there results a signal which may be expressed as 0 = exp [ - f (t')'] cos .(f(f)' - +) - exp [ - f (t' + s ) ~ ] (49) In Eq. (49) the high frequency component has been aver- aged out and 6 is the delay time between the received pulse and the reference pulse. If the pulse maxima coin- cide (6 = 0), there results a net signal and, since E is small, Expression (50) can be written as = - 2 8 ($- (t')' - z) 1 ' exp [ - 3 (t')'] (51) The pulse Eq. (51) is symmetric about its maximum (t' = 0). If the delay time S is not zero, the pulse shape is asymmetric. Therefore, by changing S , it is possible to measure the group velocity accurately if the quantity E is not too small. For the DSN, however, E is far too small as we have seen previously and therefore a pulse shape analysis is out of the question. In the foregoing, we have shown that higher order effects of the interplanetary plasma on radio signals as utilized in the DSN are negligible as far as a determina- tion of the electron content is concerned. We have also seen that a lateral change in electron concentration engenders very small angular deviations of the ray path. They also lead to a negligible change in the apparent range. The first order theory is therefore completely ade- quate to deal with the charged particle content. 1. MacDoran, P. F., “A First-Principles Derivation of the Differenced Range Versus Integrated Doppler (DRVID) Charged-Particle Calibration Method,” in The Deep Space Network, Space Programs Summary 37-62, Vol. 11, pp. 28-34. Jet Propulsion Laboratory, Pasadena, Calif., Mar. 31, 1970. 2. Koehler, R. L., Interplanetary Electron Content Measured Between Earth and the Pioneer VI and VII Spacecraft Using Radio Propagation Effects, Report SEL 67-051. Stanford Electronics Laboratories, Stanford University, Calif., May 1967. 3. Eshleman, R., “Radar Astronomy of Solar System Plasmas,” in Solar System Radio Astronomy. Edited by J. Aarons. Plenum Press, New York, 1965. 4. Ginzburg, V. L., Propagation of Electromagnetic Waves in Plasma. Gordon and Breach, Science Publishers, Inc., New York, 1965. 5. Stelzried, C. T., Faraday Rotation Measurement of a 13-em Signal in the S o h Corona, Technical Report 32-1401. Jet Propulsion Laboratory, Pasadena, Calif., July 15, 1970. 6. Wait, J. R., Electromagnetic Waves in Stratified Media, p. 86. The Macmillan Go., New York, 1962. 7. Land, J. A,, and Croft, T. A., J. Geophys. Res., Vol. 75, p. 4623, 1970. W. lushbaugh Communications Systems Research Section A hardware version of an optimal convolutional decoder is described. This decoder implements the Viterbi Algorithm for maximum-likelihood decoding of short constraint length convolutional codes. I t 8 capable of decoding at data rates up to a megabit for codes of constraint-length 3,4, or 5 at code rates ?h or %. (Mod 2) of the K - 1 bits in the shift register, and the new data bit, are computed and transmitted. The new data bit therein shifted out. Clearly, the encoder can be viewed as a finite-state machine with ZK-l states. For each of the 2K-1 states that the encoder can assume, there are only two predecessor states possible e.g., if K = 4 and the pres- ent state is 001, the two possible predecessors are 010 and 011. Thus each state has a possible zero or one predeces- sor, the zero or one being associated with the Ieast sig- nificant bit of the state number. The Obtained by (Ref* ') for is then shifted into coder the oldest bit the decoding of short constraint length convolutional codes using the Viterbi decoding algorithm created con- siderable interest in the practical application of the algo- rithm. Layland (Ref. 2) described an all software decoder Of &is type rate % code. This article describes a hardware version of a Viterbi decoder capable of a megabit data rate for con- straint lengths 3,4, or 5, at code rates M or 4s. Of a kbits/s data rate for a = I The optimum convolutional decoder contains infonna- tion pertaining to all possible states of the encoder i.e., there are ZK-l sets of likelihoods or metrics, and 2K-1most- A convolutional encoder for a code of constraint length K and rate 1/V consists, typically, of a shift register of length K - 1, coupled with V parity-check adders. For each new bit to be encoded, the V linear combinations likely bit streams (called survivors) leading to each of the states. At each step in the algorithm, a computation ele- ment associated with each state examines the two possible predecessors, adds the correlation between the branch symbols from each predecessor and the new V received symbols, and chooses the higher likelihood as the new metric for that state. The survivor associated with the winner is also copied as next state survivor and the zero or one associated with the winner is added as the new- est bit in this survivor stream. It has been shown experi- mentally (see Ref. 2), that oldest bits of the 2"-l survivors converge to each other and to the maximum likelihood decoded answer. esi Two main systems designs were originally considered for the Viterbi decoder. These were the serial computa- tion of the Z K - l states which would use parallel operations on each of the state metrics, and the finally adopted design of parallel computations of the states with a com- puting unit associated with each of states using serial arithmetic. This design was adopted because the design aim of completing one branch (bit) calculation in 1 ps could easily be met, and because the hardware for one state calculation could be put on a printed circuit board and enable the design to be expanded to larger constraint lengths. A picture of the state board is shown in Fig. 1. Sixteen of these boards were used in the present design, and in general 2K-1 are needed for a constraint length K machine. er st The logic diagram of the state board is shown in Fig. 2. The state board consists mainly of shift register memory, 10 bits for the metric of that state, 32 bits for the survivor stream associated with that state and some serial adders. To increase the speed of the system, two lo-bit regis- ters are actually used to hold metrics, one for each of the two candidates for the new state metric. The metrics are shifted at a 10-MHz rate to be consistent with the 1 p per node. Since the two candidates are serially introduced to a state board, and the correlation with the received symbols is opposite for each, the received metric from a zero predecessor state has the branch metric (for that state) added to it while the metric from the one predeces- sor state has it subtracted. The larger of these two values then becomes the new metric for the state in question. Both of these values have to be saved, since the larger of the two is not known until the sign bits of each arrive at the end of the cycle. A comparison of the two possibilities is made serially as the branch metric is added and the result is stored into the x flipflop at the end of the cycle. This x decision then steers the correct metric out of the board on the next cycle and is used to copy the correct survivor stream into the state board in question. This x value also becomes the most recent member of the sur- vivor stream. The length of the survivor stream is 32 bits broken into two 16-bit portions, so that a 16-MHz shift rate is neces- sary to copy the survivors in 1 ps. The decoded bit is taken from the oldest survivor of the all one's state board. As is shown in Ref. 2, little degradation is suffered by choosing this answer rather than the survivor associated with the maximum likelihood state at each step. eei In addition to the Z K - l state boards in the system, four special-purpose boards were needed to complete the sys- tem. These were the timing board, which develops the various timing signals needed, the metric calculator, a metric select board, and a node sync board. et& calculator. For a rate % code there are only ossible received sets of 3 symbols for each trans- mitted bit. Moreover, each state unit of the decoder always expects to see the same set of received symbols. Thus the eight possible values of sums and differences of the three received symbols can be computed in one central location and fed to the appropriate state boards. The states that receive each branch metric are determined by the code, and to change codes, only these eight signah I I I I z a z I,; need to be rerouted. This selection and distribution of the branch metrics is performed by the code select board which consists of 16 four-way select gates, one for each state board in the system, allowing up to four codes to be hard-wired into the system and selectable with a four- position switch. The metric calculator has been designed to accept sym- bols quantized up to four bits each. When these symbols are properly scaled and three are added together the range of branch metrics for a rate 45 code is -45 to +45. Changing the code rate to rate M is accomplished by setting one of the symbols to zero on the metric calculator and using the four significant results that remain as the branch metrics. To change the constraint length of the code, state boards are associated in groups of 16/2='-l where K' is one of the other possible values of constraint length i.e., either 4 or 3. Each of the groups of states now corresponds to a superstate with each state unit within it receiving the same branch metrics. It can be shown that all states within a superstate perform identically after a transient of 16/2x'-1 steps in the algorithm. 2. Node sync. Layland has shown in Ref. 3 that node synchronization of a Viterbi decoder can be obtained using the decoder state likelihoods integrated over sev- eral hundred code branches assuming only that the data source is sufficiently random. The design finally adopted, however, does not use integration over a fixed number of nodes but instead compares the uphill climb of the metrics to the sum of the absolute values of the received symbols Figures 3 and 4 show the experimental densities of the state likelihoods minus for a K = 4 and K = 5 code. It was observed from these curves that if +4 is added to both curves, then the in-sync version would be shifted positive while the out-of-sync curve would remain negative. Thus an in-sync indication is the positive drift of the quantity OUT OF SYNC 0 K = 4CODE 2 dB -15 -30 I I $ l r i l INCREMENTAL STATE LIKELIHOODS - 1 I I I IW -30 0 -15 V INCREMENTAL STATE LIKELIHOODS - 1 where Mk is the kth metric and M, is the metric at some starting point, In the actual design, 10 bit numbers are used for this quantity which is initially loaded with some M, + 256 and the sign bit is monitored for positive or negative overflow. These overflows are in turn divided by three to increase the integration time to the order of mag- nitude suggested by Eayland. For each V - 3 code there are 6 Merent permutations of sending the three symbols as well as a twofold sym- metric way of interpreting the order of the taps on the encoder. For this reason a standard way of interpreting each code is desirable. The present decoder has been wired according to the conventions defined by the follow- ing algorithm: (1) Write the code in binary on V lines and write the weight of each term below, e.g., for a K = 6 and V = 3 code 1 0 1 1 1 1 1 0 1 1 0 1 1 1 0 0 1 1 Weight of eachtap: 3 1 2 2 2 3 (2) Take the K digit number that results and its reverse e.g., in .this case, 3 1 2 2 2 3 and 3 2 2 2 1 3 and choose the larger of the two as the standard order for writing the code. The newest bit of the encoder will now be associated with the most sig- nificant bit of this number. ewrite the code as in step 1 but in the standard order e.g., 1 1 1 1 0 1 P O 1 1 0 1 1 1 0 0 1 1 (4) Now interpret each row as a binary number, most significant bit on the left, and reorder them by mag- nitude. The smallest of these will be s,, the next Sz, etc., e.g., ! & = l o 1 1 0 1 s , = 1 1 0 0 1 1 s 3 = 1 1 1 1 0 1 This is the uniquely defined standard form for the code and the encoder is to be wired to transmit these 3 symbols in the order SI, Sz, S3. Table 1 shows the standard form of the three codes presently wired into the machine. I Code I I sz I s a K = 3 , V = 2 101 111 - K = 4 , V = 3 I 1011 I 1101 I 1111 K = 5, v = 3 10101 1101 1 11111 S Figure 5 shows a picture of the completed system. The 16-state board and four special boards, which occupy two slots each, just fills a standard chassis with 25 slots. A single 5-V power supply is used and the system draws 12 A. System debugging has been completed and simula- tion has shown that the system operates in agreement with error rate predictions from previous software simulation. er, J. A., “Sequential Decoding: Short Constraint Length Convolutional es,” in Supporting Research and Development, Space Programs Summary 37-54, Vol. 111, pp. 171-177. Jet Propulsion Laboratory, Pasadena, Calif., Dec. 31, 1968. ayland, J. W., “Information Systems: Capabilities of an All-Software Optimum Convolutional Decoder,” in The Deep Space Network, Space Programs Sum- maries 37-62, Vol. 11, pp. 64-66. Jet Propulsion Laboratory, Pasadena, Calif., March 31, 1970. 3. Layland, J. W., “Information Systems : Synchronizability of Convolutional Codes,” in The Deep Space Network, Space Programs Summary 37-64, Vol. 11, pp. 4440. Jet Propulsion Laboratory, Pasadena, Calif., August 31, 1970. G. lutes Communications Systems Reseorch Section Frequency dividers with improved phase stability were recently developed for use in the hydrogen m e r frequency standard. The commonly used methods of frequency division were found to have excessive phase noise and long-term drift and would seriously degrade the inherent stability of the frequency standard. 1. Frequency dividers with improved phase stability were recently developed for use in the hydrogen maser fre- quency standard. The commonly used methods of fre- quency division were found to have excessive phase noise and long-term drift and would seriously degrade the in- herent stability of the frequency standard. quency to pass through the gate at the exact time and interval to achieve the desired division. A block diagram of the improved frequency divider is shown in Fig. 1. The improved frequency dividers consist of a digital divider in parallel with an analog gate. The analog gate is opened by the digital divider allowing the input fre- The first amplifier, AI, is an isolation amplser which provides a constant 50-Q input impedance and a voltage gain of approximately seven. The output of A1 is im- IN D SHAPING DIGITAL - CIRCUIT DIVIDER TIMING CIRCUIT 56 1 2 3 4 5 6 GATE INPUT I I I I I I I GATE I I I I CONTROL I I I I I I I I I I I I I I I I I I I GATE OUTPUT ~ GATE I- A G A T E I OPEN I I OPEN1 iwide- pressed on an analog gate which is opened by a digital signal to allow one half cycle of the signal from A1 to pass through the gate every nth cycle, where n is the division ratio. The clock for the digital divider is taken from the out- put of A1 and goes through a timing circuit which is used to set the time relationship between the analog signal and the digital signal at the analog gate (Fig. 2). The signal is then amplified by A2 and shaped properly to drive the digital divider. The output of the digital divider is one pulse every nth cycle of the input frequency and has a pulse width of one cycle, as shown in Fig. 2. The output pulse from the digital divider goes through an inverting amplifier, A3, which drives the analog gate. The gate is opened and allows the positive part of the input signal to pass through. This results in a signal at the input of the tuned amplifier, A4, which is the positive half sine wave every nth cycle of the input signal (Fig. 2). The output of the tuned amplifier drives a distribution amplifier which is not shown. The phase noise of the improved frequency dividers was compared to several other modules presently in use. An improvement in phase noise of nearly two orders of magnitude over a typical frequency divider now in use is indicated by the graph of Fig. 3. This graph shows the single sideband phase noise density relative to the carrier as a function of frequency for the modules which were compared. The measurement method is described in Ref. 1. SINGLE SIDEBAND NOISE DENSITY RELATIVE TO CARRIER I '\ X- DISTRIBUTION AMPLIFIER FOR DSIF LX g FREQUENCY DIVIDER IN DSIF X- DISTRIBUTION AMPLIFIER IN HYDROGEN MASER -1 60 I I I I 0 20 40 60 80 10 FREQUENCY, Hz The bandwidth of the prototype is approximately 10%. Much wider bandwidths could be achieved since it is lim- ited only by the timing circuit and the tank circuit of the tuned amplifier, A4. Other test results are shown in Table 1. This type of divider does not have the problems associ- ated with more commonly used designs, such as spurious modes of operation, unreliability, sensitivity to power sup- ply variations and temperature effects. This design utilizes the reliability of a digital divider but eliminates the phase noise inherent in digital cir- cuitry. A considerable improvement in frequency dividers is indicated by the test results, which show substantially better phase stability, reliability, temperature stability, reproducibility, bandwidth capability, and insensitivity to power supply variations. easuremenl Input 2 output z Bo n dwidth Phase shift with 4emperafure Amplitude variation with temperature Distortion Power supply Nominal 50 Q 50 Q See text - - <5% THD rt 15 ond f 5 V Measured <1.1:1 VSWR ' 0.075+, that Rcomp ( p ) exceeds Rm(p) , the largest rate at which reliable com- munication can be achieved by an MD algorithm. For binary antipodal signalling over a white gaussian chan- nel, with binary detector quantization, the situation is more complicated for MD algorithm: there exists a dimen- sionless rate R = 0.5377' at which the minimum &,/No required for reliable communication is minimum, 2.547'. In order to surpass this with sequential decoding-again with binary quantization-one must use convolutional codes of rates < 0.3196. I Now before giving the details of this calculation, let us admit that these results depend upon an unproved hypothesis. That hypothesis is that €or a fixed rate R, the largest possible value for the minimum distance of a block code of length n is asymptotically equal to n .Hi1 (1 - R), where H, is the binary entropy function. We really only know that the best minimum distance is at least this good. owever, this hypothesis is widely believed to be true; and if it should turn out to be false, the results of this note can easily be modified and will not be qualitatively changed. The details of the calculation are quite easy. For the BSCl with error probability p , capacity is 1 - H , (p), and Rcornp is By our above assumption, for large n an MD algorithm will correct Vi R a Hi1 (1 - R) errors, and so by the law of large numbers if p < M Hi1 (1 - R ) reliable communication can be achieved. Thus we define Rnn, (p) = 1 - H , (2p); this is the supremum of the set of rates at which MD algorithms will succeed in driving the error probabili- ties to zero. Figure 1 is a plot of Reornp (p)/Cap (p) and RMD (p)/Cap ( p ) for 0 6 p 0.25. For p A 0.25 R m = 0, while As stated above, the two curves cross at p = 0.075355+. For the white gaussian channel, with binary antipodal signalling and binary detector quantization, the calcula- tions are only slightly more difficult. If the energy of one binary symbol is Es, then the probability of detector error is where 1BSC = binary symmetric channel. and % N o is the spectral density of the noise process. Instead of dealing with capacities, we wish to know the smallest Eb/N0 for which the probability of error can be made arbitrarily small. With block (or convolutional) codes of rate R the energy available per channel symbol is E8 = Eb - R. Thus with maximum likelihood decoding, reliable communication can be achieved if R 6 1 - H , (p); thus the minimum E#, is 1 - (Q-l [Hi1 (1 - €I)]), (capacity) For Rcornp the equation is i.e., 1 - 2'4 (2R - 1)"])2 (Rromp) 2R 2 Finally, for MD decoding, we need These values are plotted in Fig. 2. Note that both the Rcornp and the Cap curves are monotone decreasing with R, re- flecting the gains which accrue with increasing band- width occupancy. However, the MD curve has its mini- mum at R = 0.537724' for which Eb/No = 2.547'. In order to surpass this with Rcomp, we see that R L 0.3196' is required. I t is interesting to compare Fig. 2 with Fig. 6.49 in Wozencraft and Jacobs (Ref. 1, p. 442) where similar behavior was observed in the performance of BCR codes versus convolutional codes. 1 .o 0.8 0.6 0.4 0.2 i i I i i 0.05 0.1 0.15 0.2 0.25 TRANSITION PROBABI LlTY RATE, bits/synbol 1. Wozencraft, J., and Jacobs, I., Principles of Communication Engineering, John Wiley & Sons, New York, 1965. R. 1. McEliece Communications Systems Research Section 1. E. Savage Engineering Division, Brown Universify With the intention of finding binary block codes which are easily decoded, we examine decoding functions consisting of one level of symmetric functions. W e find that all codes so decodable with fixed error correction capability t have rate less than 1/(2t + 1) and that this rate is achieved b y the repetition code which has two code words and length 2t f 1. Decoding functions conskting of two or more levels of symmetric functions include all binary functions and can therefore decode arbitrarily good binary codes. . I The design of error control systems often begins with the selection or invention of a code with good error cor- rection capability, and this is followed by the invention of a decoder. Decoders invented in this fashion are often quite complex. This causes one to wonder whether this complexity is the result of an unfortunate choice for the code. In this report we invert the conventional approach to coding by fixing the decoder type and then examining the type, and characteristics of codes decodable with them. We choose to consider decoders for binary, block codes whose decoding function f = ( f l , f 2 , * . , f i ~ ) is a collection of functions, each of which is symmetric on some subset of the components of a received word. We choose to let the code block length be n and denote by M the number of code words in a code. When the decoding function consists of a single sym- metric function ( k = l), we show that all codes decodable with it which have minimum distance d = 2t + 1 must satisfy M I 1-1 2t + 1 = 2t + 1 + 1 It can also be shown that equality can be achieved. As a consequence of Eq. (l), if d is fixed, then the achievable code rate A decreases with increasing n. This contrasts sharply with the BC codes, for example, where pi ap- proaches 1 with increasing n when d i s fixed. When the decoding function consists of k symmetric functions, we show with a “sphere packing”-like bound that the maximum code rate is achieved when the domains of definition of f l , f 2 , . . . , f k are nonoverlapping. This maximum rate is bounded by 1 R L - j and is achievable with a code which consists of the con- catenation with itself n/d times of the code containing t h e 3 word and 1 word of length d. Thus the best sym- metrically decodable codes are both repetitious and not very good. Decoding functions consisting of two or more levels of symmetric functions include the majority-logic decod- able codes since the first level can be used to form the syndrome of a linear code using modulo-two sums, which are symmetric functions, followed by one or more levels of majority functions which are also symmetric. With this report we document a case study of decoders and hope to encourage interest in this unconventional approach to coding. 1. S A binary, symmetric function f ( x l , * , 2,) of n binary variables has the same value on all n-tuples ( x l , x2, . . . , x,J with the same Hamming weight. Thus, if Wt(X1, . * * ,x,) = wt(q1, * . . ,!Ifl) Consequently, a binary, symmetric function can assume at most n + 1 values, corresponding to the n + 1 different weights of binary n-tuples. Also, symmetric functions are rather easy to realize with logic elements. With, at most, n counters, counting to at most n and a few additional logic elements, any symmetric function can be realized. The symmetric functions form the only known class of functions of low complexity yet rich enough to offer some hope that they can be used to decode codes with good rate and error correction capability. Consider a binary code C with IC1 = M codewords of length n. We suppose that received words R are decoded by a symmetric function of n variables, f (xl, . * * , xn). If the code is to correct t errors, then if E is an error pattern containing L t ones, f (c + E ) = f (c ) for each MC. But since, as we have seen, the value of a symmetric function depends only on the weight of its argument and since an error pattern of weight t or less can change the weight of c by up to t, we see that the weights of the codewords Ci must satisfy ~ w ( c i ) - ~ ( c i ) I 1 2 t + 1 , i#f (2) and the maximum number of weights in the range [0, n ] which can be chosen to satisfy Eq. (2) is (3) Equality can be achieved by choosing one word of weight 0, one of weight 2t + 1, . . , one of weight q (2t + l), where = l&] Notice that the rate of such a code is bounded by and we can actually achieve rate l / (2 t + 1) with the repe- tition codes (00 . . . O and 11 * . . l } with decoding func- tion f = majority vote. We now consider a somewhat broader class of decod- ing functions. The decoder is to consist of k functions fi, and each f i is to be a symmetric function of ni of the variables xl, x2, * , xn. We denote the domain of the function f i by Di. If the code C is to correct t errors, then the projection Ci of the code onto each domain Di will itself correct t errors and must be decodable by f i . Hence by the results in Section ZIZ, Since M L M , M, 1 Mk (Footnote l), which by Eq. (7) yields k (4) If the domains Di are disjoint, this bound can be achieved, since we have seen in Section IIZ that it is achievable for k = 1, and so we may take C = C, X C2 X X c k . If however, the Di are not disjoint the various projections Ci cannot be chosen independently. Nevertheless we argue that nothing can be gained by having the Di’s overlap, as follows: Let E be the subset of the variables { xl, . . . , x,} which are involved in more than one of the functions f i , and let C, be the projection of C onto E . For each word eeE let Ci, be the set of codewords in Ci which are “compatible” with e in the sense that each codeword in Ci, agrees with e on Di n E . Then the total number of codewords in C compatible with e is A M , , M, , * * * M r e and so Since all codewords in Ci, agree in mi = I Di n E I posi- tions, but their weights must be separated by 2t + 1, we must have 1 ni - mi + 1 2t + 1 M i , .L Also, since Ci = UCi, and then ni + 1 [ G I Combining Eqs. (5) and (6) we obtain (7) ,Of course, this assumes all variables are used. If some are not, we just connect the unused ones to a symmetric function whose value is constant. k 1 This is the same form as Eq. (4), with a total block length of i = 2 This is , A n since for i k 2 the sets Di - Di n E are dis- joint from each other and from Di. Hence if the Di are allowed to overlap, we can get no more codewords than we could with disjoint Di at the same (or smaller) block length. There remains the problem of maximizing Eq. (4) over all values of k and all choices of ni subject to n,+ * a - +nk=n For each i write ni = qi (2t + 1) + ri with OAT^ <2t + 1. Then we may replace each f i with qi functions of 2t + 1 variables each and obtain possible codewords instead of [SI = qi + 1 Since 2 a h a + 1 for all integers a LO, if the product in Eq. (4) is to be maximized no ni needs to exceed 2t + 1. So if we w r i t e n = q ( 2 t + 1 ) + r , 0 . 1 r < 2 t + 1 7 and take n, = . . = n, = 2t + 1, nqCl = r, we get M = 29 as the maximum possible number of codewords, and so the rate of the code satisfies But since rate l/(2t + 1) can be achieved by a repetition code of length 2t + 1 and “majority vote” decoding, we see that no code that corrects t errors and is decodable by a sub-symmetric decoder is better than a repetition code. Hence, good symmetrically decodable codes are repetitious. 6 1. S. Reed Electrical Engineering Department, University of Southern California In this article a method is devised for testing for a possible fault in a gate in a larger switching circuit, that does not require isolating the suspicious gate from the rest of the circuit. The techniques involve a Boolean difference calculus remi- niscent of, but not identical to, ordinary difference calculus. , x n ) is a realization of the mapping f I 2(n) + 2 where 2 is the set of two elements, i.e., 2 = (0, l}, and is Cartesian products of 2, taken n times, n t i m e s Mapping f is called either a switching or Boolean func- tion. Switching circuits are composed of a number of sub- switching circuits. These elementary switching circuits often occur in modules and are called gates. A typical switching circuit is illustrated in Fig. 1. It happens that this circuit consists only of nand gates. In terms of its gates the circuit of Fig. 1 is the following set of five equations: f = g , g 2 g1 = x,g g 2 = x4g - g = X,X& u = xzx3 - Variables xl , x2, x3, x, are called the primary or input vari- ables to the circuit. The outputs of the gates u, g , gl, gz are known as internal or secondary variables, and finally variable f is the output of the circuit. The bars in Eq. (1) denote complementation, i.e., if ~ € 2 , E = 1 @ u where @ is sum, modulo 2. Suppose f is a switching function of the n binary vari- ables and u is an internal variable of the circuit. The depen- dence of f on both x and u is represented functionally as where x denotes the binary variables x l , xz, . . . , xn. Nec- essary and sufficient conditions for f to be a dependent function of u are well known ( To illustrate the notion of functional dependence, con- sider again the circuit of Fig. 1. The output (3) b 65 . 1. is a dependent function of both function gl (x) and func- tion g 2 (x) where x = (xl, x2, xs, x4). Similarly (see Eq. l), are both dependent functions of a primary variable and of the same gate g ( x ) . Equations (3) and (4) illustrate functionally another important concept. This is circuit fan-in and fan-out. Since g l ( x ) and g 2 ( x ) in (3) “feed the single gate f , the out- puts gl and g2 are said to fan-in to f . In Eq. (4 ) gates gl and g2 are both fed by the same gate g. Schematically Fig. 1 thus requires the output lines of g to fan out from gate g in order to simultaneously feed gates gl and g2 . The generalization of the concept of fan-in and fan-out to n gates is evident. %s Let y = f (2, u (x)) be a function, dependent on an in- ternal gate variable u(x) where x denotes the set of pri- mary variables x1,x2, . . . ,xn. Our purpose is to devise a test for a possible error or fault in the physical realiza- tion of gate u. If a gate such as u can be isolated from the rest of the circuit f , the testing of u would be a straightforward mat- ter. However, today a subcircuit u of a digital integrated circuit usually cannot be disconnected without serious damage to the overall circuit f . Assar~~moiv A. The only terminals of the circuit f avail- and the output y . able for testing are the primary input variables xl, . . . X?l In order to impress both values 0 and 1, respectively, on u in a reliable manner with test equipment, it is con- venient to assume there is only one or no fault in the cir- cuit making up f . If u is the gate variable to be tested, one assumes the only possible error in the gates of f resides in the output of gate u. The above single-fault assumption is formalized as follows: e Let y = ~ ( x , u ( x ) ) be a switching func- tion dependent on gate function u (x) where x = (XI, x,, . . ’ ,XJ. If gate u is being tested, there is only one possible error or fault in the gates which physically compose f, and that is at the output of gate u. Now consider the possible errors or faults which can occur at the output of gate u. The types of error possible in u are perhaps best illustrated by the symbol transition diagram of Shannon. Such a diagram is shown in Fig. 2. In Fig. 2 impressed values and received values correspond to transmitted symbols and received symbols, respectively, in the usual Shannon diagram for a binary (or two symbol) channel. u (x) is the desired value or the value which the tester endeavored to impress upon the gate u, whereas uR(x) is the actual value at the output of gate u as a function of input configuration, ~€2”. If one could dis- connect the output of gate u from the rest of the circuit f and connect the output of gate u to a tester, then uR (x) would be the actual value received by the tester from gate u. If the symbol or value k is impressed on gate u, the probability that the same value is received is given by qk where k = 0, l . The probability is p k = 1 - qk that the wrong or opposite symbol is received. In general, the probabilities in Fig. 2 allow for the pos- sibility of intermittent errors. We are interested for the present only in the static or permanent type of error. These are the probability one type errors. Digital computer designers denote such errors as faults. R U 0 rn fer u 66 JPk TEC T 32-952 e. All possible probability one type of errors or faults are illustrated in Fig. 3. The type 1 fault is u stuck-at-zero. Type I1 fault is the u stuck-at-one. Finally, the type 111 error is u complemented. Type 111 type errors occur quite frequently in final LSI (large scale integrated) circuit layouts. The bar for com- plementation is often omitted or inserted erroneously dur- ing the final conversion of switching circuit equations to planar circuits. Type I and I1 faults occur more usually as electronic faults. We can now prove the following theorem: . Let y = f (x, u (x)) be a switching function dependent on a gate function u (x) where x = (xl, xo, . . . , xn). Under Assumptions A and B a single fault in u of any of the three types I, 11, or I11 of Fig. 3 can be detected if and only if there exist two input test configurations xo and x1 in 2” such that R U 0 0 TYPE I STUCK-AT-ZERO (s-a-0) 1 R U Proof: Suppose xo and xl exist such that (1) and (2) are true and u has a fault, e.g., a type I fault. From Fig. 3 we have U R (x0) = u (x0) U R (XI) = u (XI) for a type I fault. By Assumptions A and B only the out- put y = f of the circuit is observable and the observed value of y is for input configuration ~€2” . If xo and x1 are test config- urations, it is apparent that a fault in u can be detected if and only if for either k = 0 or k = 1. Substituting the above value of uR (XI) into Eq. (5 ) yields y R (XI) = f (XI, U(X1)) which is not equal to y (xl) = f (xl, u (XI)). Thus by Eg. (6) any type I type error can be detected if xb exist such that (1) and (2) are true for k = 0 and 1. In a similar fashion it can be shown that type I1 and I11 errors will be detected and suaciency has been shown. It is evident that condition (1) is necessary. By Fig. 3 for a type I error in u U R (x0) = u (x0) U R (21) = z i (XI) and for a type I1 type error in u U R (x0) = a (x0) U R (XI) = u (XI) Hence for a type I type error in Ti Y R (x0) = f (x0,Z (XO)) YR (XI) = f (xl, ?J (XI)) and for a type I1 error Thus to detect both a type I and I1 error it is necessary 5 To develop a calculus for the Boolean difference, recall first the fundamental expansion formula of Boolean alge- bra. If xc2 and f ( x ) is the mapping f I 2 + 2, then f ( 4 = f ( O ) x @ f ( l ) x (8) = f (0) @ [ f (1) @ f (0)lx respectively. Thus the necessity for conditions (1) and (2) = f ( 0 ) @ A f I E = o * x is established and the theorem is proved. where A f l , = , denotes the difference with respect to x, evaluated at x = 0. The second and third forms of Eq. (8) can be recognized as Newton’s interpolation formula over the field of two elements, e.g., Ref. 6, pp. 66-70. Condition (2) of the above theorem is true if and only if bf (x , = f (9, (x”) @ f (9, (9)) = 1 (7) The partial Boolean difference was defined by the left side of Eq. (7) in Section 11. We now determine some ele- mentary rules and properties of the Boolean difference. Suppose f (u) and g (u) are mappings of 2 into 2. where @ denotes sum, modulo 2, or what is often called the exclusive or operation of Boolean algebra. The left side of Eq. (7), the expression f ( x , U ( x ) ) @ f (x , u ( x ) ) , is called the partial Boolean difference with respect to u and is denoted by the suggestive notation, A f 6 , u ) . The partial Boolean difference was first defined and used by For this rule use definition (7) and formula (8). tf = f (u) @ f (E), f (u) = f (0)u @ f (1)E. The rule follows from u @ Ti = 1. the author (Ref. 2) in analyzing certain error-correcting codes. Aker’s in Ref. 1 develops some of the calculus and Boolean functional analysis associated with the partial Boolean difference. ULE HI. A is a “linear” operator. In terms of the partial Boolean difference, Theorem 1 can be restated as follows: t t ( f @ g ) = y @ p . Let y = f (x,u(x)) be a switching function a gate function u ( x ) where x = ( x I , x2, . . . , xn). Under assumptions A and B a single fault u of any of types I , II, or III of Fig. 3 can be detected if and only if there exists two input test configurations xo and x’ in 2“ such that Aa = 0. This assertion is immediate by definition and the asso- ciativity of addition, modulo 2. . If a d is a constant with respect to variable u, ZL (1) u ( 9 ) = k (2) ef (9, (x”) = 1 for both k = 0 and k = 1. Section IZI will be devoted to the development of a dif- ference calculus for Boolean algebra which is quite anal- ogous to the classical difference or differential calculus of many real variables. The chain rules of this calculus simplify the computation of the partial difference required in Theorem 2, making possible the automatic generation of test configurations for all gates of a circuit. As stated, Theorem 2 formalizes rather sufficiently a number of related developments by other authors (Refs. 35). Rule I11 is immediate and Rule IV can be verified directly by calculation. Rule IV can be generalized induc- tively to a product of n terms. The following rules of differencing are direct conse- quences of Rules I through IV. ULL ’Bd. ef = Af This follows from Rules I1 and 111. u u u T 32-P5 b. I! I. Let ul,uz, . . . ,un be n binary variables, i.e., variables with domain Rules VI and VI1 are immediate by Rule I. An important tool for computing the partial differences, needed in Theorem 2, is provided by the partial difference "chain rule" of the next theorem. E ~ ~ E ~ 3. Let f, g,, g,, . . . , gk and u be Boolean func- tions of n binay variables x,, x,, . . . , xn. Suppose func- tion f is dependent on functions g,,g,, . . , g , and in turn functions g,, g,, . . . , gk are dependent on function u. Then the partial Boolean difference with respect to u satis- fies the chain rule, where A(l") 8,].8,?' ' . g i , denotes the mth partial difference with respect to func- tions gil,giz . . . gi,. rdp(~j: A proof for this theorem is provided here only for can be developed for arbitrary k. k = 2. A similar pro For more details see For this case f is functionally of the form f = f (gl (u), gz (u)). Use Eq. (8) to expand f (gl, gz) with respect to each variable, separately, as follows: Formula (9) can evidently be generalized to k variables (Ref. 2). Take the difference of both sides of Eq. (9) with respect to u, then since f (0,O) and the coefficients of gi, gz and glgz are in- dependent of u. But by Rule IV 6 (g1g2) = ($1) e gz @ gl e @ ($2) Substituting this in Eq. (10) and collecting coefficients, we obtain The theorem for k = 2 is proved once expansion (8) is used to identify the coefficients of Agl and Agz with Af and Af, respectively. 81 82 The above theorem and previous rules provide an ade- quate machinery for calculating partial differences. In the classical difference calculus there are existence theorems associated with certain classes of difference equations and boundary conditions. Similar theorems can be proved for the present difference calculus of Boolean algebra. An example of such a theorem is the next one. This theorem provides a "global" criterion for the existence of test con- figurations xo and x1 which satisfy both the "boundary" condition (1) u ( Z ) = k and the difference equation (2) ~ f ( X & ) ) = 1 of Theorem 2. . There exist two test configurations xo and d, satisfying (I) and (2) of Theorem 2 if and only if functions u (x ) and f (x, u (x ) ) are such that (3j-G) e f (x , 4 z 0 (4) U ( 3 C ) " i f ( X , U ) #O. f : If (1) and (2) of Theorem 2 are true, then u ( x o ) 4f (xo, u) = 1 and u (XI) if(XI, u) = 1 and (3) and (4) - J hold. Conversely if (3) is true, there exists xo such that u (xo) a $f (xo, z() = 1. But this implies (1) of Theorem 2. Similarly (4) implies (2) of Theorem 2 and theorem is proved. - s Let f be the switching circuit of Fig. 1. f is a circuit of four primary input variables x,, x2, x3, and x4 with circuit equations given by Eq. (1). As an application of Theorem 2 we will find test con- figurations xk = ($, x;, e, x:) for k = 0 , l which will test for a fault in gate u. To do this we must compute the par- tial difference if and set it equal to one. This is done using the chain rule of Theorem 3 and the difference calculus These two configurations are displayed in two rows of the incompleted Table 1. Note that relation = 1 and u ( Z ) = k for k = 0 , l are already satisfied in remains to complete the table, using Eqs. (l), and to show that +f = 1 can be satisfied. liable of test confi s for gate u le 2. Test configurations for gate u (~2x3 = 0) A completed table of test configurations for gate u is shown in Table 2. The two blanks in the table may be fiIled with elements of the set The solutions of conditions (1) and (2) are not always unique. For this example the test configuration, xo = (x;, xo,, x;, x;) = (1,1,1,1) is unique and the configuration x1 is not unique and can be chosen to be x1 = (x;, xi, x;, xi) = (1,0,0,1) To test for the presence for a fault in any primary input, internal gate, or output one applies Theorem 2 repeatedly, finding test configurations for all such variables. A minimal set of test codgurations is obtained by taking minimal union over the set of all test configurations. If n is the number of variables to be tested, the minimal set never exceeds 2n test configurations. 7 1. Akers, S. B., Jr., “On the Theory of Boolean Functions,” J . SIAM, Vol. 2, No. 4, 2. Reed, I. S., “A Class of Multiple-Error-Correcting Codes and the Decoding Scheme,” IRE Trans. PGIT-4, 1954. 3. Armstrong, D. B., “On Finding a Minimal Set of Fault Detection Tests for Combinational Logic Nets,” IEEE Trans. Elec. Comp., Vol. IC-15, No. 1, Feb. 1966. 4. Sellers, F. F., Hsiao, M. Y., and Bearnson, L. W., “Analyzing Errors with the Boolean Difference,” Annual IEEE Computer Conference, Chicago, Ill., Sept. 1967. 5. Chiang, A. C. L., Reed, I. S., and Banes, A. V., “Data Sensitation, Partial Boolean Difference and Automated Fault Diagnosis,” to be published IEEE Trans. on Elec. Comp. 6. Van der Waerden, B. L., “Modern Algebra,” Vol. I, Frederick Unger Publishing Co., New York, 1949. Dee. 1959. 2-iF526, Yo&. II A. Sward Communications Systems Research Section Measurements have been made on the JPL hydrogen masers to determine the average fractional frequency departure versus averaging time. In addition, the receiver section has been modified with a newly developed frequency synthesizer. Recently measurements have been made to determine the frequency stability of the new JPL hydrogen masers. In addition, the old receiver-synthesizer section has been modified with a newly developed which will not only improve the ability of the receiver, but also complexity. This article is a continuation of the hydrogen maser development frequency synthesizer performance and reli- decrease its size and the articles relating to found in Refs. 1 to 3. I Figure 1 is a plot of the standard deviation of the aver- age fractional frequency departure versus averaging time for two masers. With the power output of the two masers at -85 and -89 dBmW, the measured short-term sta- bility approaches the theoretical limit predicted by Cutler and Searle (Ref. 4). S equency synthesizer is being built in conjunction with Dana Laboratories which will provide the maser with a frequency settability of 7 parts in 10-l8. Its frequency range will be 400 to 510 k Hz. The synthesizer, when subjected to temperature steps of 2SoC, was found to have a maximum drift rate of 0.3 X degrees of phase/second at 50 corresponds to a frequency stability of 1.6 p This is two orders of magnitude improvement over pres- ent units and enables the synthesizer to achieve resolution. A diagram of the new receiver is shown in Fig. 2. The synthesizer is driven dire MHz signal at the output of the 1 amplifier. Output frequencies from z, 10 MHz, 5 MHz I I I I I I I I I 405.751 KHz 10-11 - t 10-12 - \ c! b 1 0 - l ~ - 1 0 - l ~ - 1 0 - l ~ I I 1 I I I I I I 0.01 0.1 1 10 1 02 1 o3 I o4 1 o5 1 o6 TIME, s FREQUENCY DIVIDER I I 1 I I I I 100 MHz i 5 FREQUENCY MULTIPLIER 20 MHz FREQUENCY DIVIDER AND DI STRl BUT1 ON AMPLIFIER 1. Erpenbach, W., Finnie, C., and Petty, S., “Frequency Generation and Con- trol: Atomic Hydrogen Frequency Standard,” in The Deep Space Network, Space Programs Summary 37-58, Vol. 11, pp. 5255. Jet Propulsion Laboratory, Pasadena, Calif., July 31, 1969. 2. Sward, A., “Frequency Generation and Control: The Hydrogen Maser Fre- quency Standard,” in The Deep Space Network, Space Programs Summary 37-59, Vol. 11, pp. 40-43. Jet Propulsion Laboratory, Pasadena, Calif., Sept. 30, 1969. 3. Lutes, G., “Frequency Generation and Control: Distribution Amplifiers for the Hydrogen Maser Frequency Standard,” in The Deep Space Network, Space Programs Summary 37-61, Vol. 11, pp. 68-72. Jet Propulsion Laboratory, Pasadena, Calif., Jan. 31, 1970. 4. Cutler, L. S., and Searle, C. L., “Some Aspects of the Theory ,and Measurement of Frequency Fluctuations in Frequency Standards,” Proc. ZEEE, Vol. 54, Feb- ruary 1966. I U. Timor Communications Systems Research Section The performance of the sequential ranging system can be improved by using a maximum likelihood receiver; however, the complexity grows exponentially with the number of components N needed to determine the range unambiguously. A new truncated maximum-likelihood receiver, based on the Viterbi decoder for convolu- tional codes, is presented and is shown to achieve a maximum-likelihood perform- ance uhile having a fixed complexity independent of N . The improvement in signal-to-noise ratio, compared to the present receiver, is 1.5 dB for PE < lo-=. Ranging systems for deep space applications achieve the required resolution by transmitting, either simulta- neously or sequentially, a multi-component signal. Gold- stein (Ref. 1) described a sequential ranging system, which transmits N + 1 squarewaves of increasing periods T , 2T, * * , 2NT. The highest frequency component yields the most accurate measurement, but with an added dis- tance of M 0 R, where R is proportional to T and M is an unknown integer. The other N components are used to estimate M and thus remove the ambiguity. In the above system, the estimation is done sequentially, that is, at each step a binary number ak is estimated from the signal component which is present at that step. The sequence a,, . . , aN, considered as a binary representa- tion of M, yields an estimate of M. The performance of the system-that is, the probability of estimating M correctly-can be improved by using a maximum-likelihood estimator, which estimates the whole sequence a,, , ax simultaneously. However, the com- plexity of such a system is proportional to 2N and is not practical for large N . We will present a suboptimum estimation procedure, which outperforms the sequential receiver and approaches asymptotically, as the signal-to-noise ratio increases, the performance of the maximum-likelihood receiver. This method is based on the Viterbi algorithm for decoding convolutional codes of short constraint length (v), and has a complexity of the order of 2", no matter how large N is. The improvement in the signal-to-noise ratio required to achieve a given error probability PE is 1.5 dB throughout the range of interest ( P E < step whenever $-I = 1, where e k - 1 is the estimate of the previous step. Thus The time-of-flight (TQF) of the signal at to can be represented by k - 1 j=1 ffk = %k + 2 (aj - ej) 2(i+1-k) (5) TQF = (M + E ) T (1) where T is the period of the first (highest frequency) squarewave, M is a positive integer and O d E < 1. Let { a N M, that is - . . , a,} be the binary representation of where ak is 0 or 1. To measure the TQF, and therefore the range, it is 3 * , aN} . The present re- enough to measure Q and {al, ceiver does this sequentially. We start by transmitting the T-period squarewave to obtain an estimate 6’of E. The receiver correlates the in- coming signal with the receiver coder squarewave and its 90 deg-shift to obtain a pair of outputs xo, yo, from which E is estimated. The correlator is then shifted by *to have a phase of 9- E . We will assume that the integration time is long enough to obtain E - e= 0. To estimate a,, * . , aN, squarewaves of periods ZkT, k = 1, * - , iV are transmitted sequentially. The outputs of the in-phase and quadrature correlators at the kth step are L...i ’ y k = r k $ . f % (3) x k = sk f nk respectively, where nk and mk are independent white gaussian noise samples of zero mean and variance u2, and ffk depends on a,, * * * 9 ak* It was shown that the probability of error is minimized by shifting the correlator waveform by 90 deg at the kth and we estimate The procedure is terminated when QN is obtained. The resulting (7) and ?yield the measured TOF. sti The “estimate-and-shift” sequential method has a strong error propagation property. Suppose the estimate 2, of a, is in error, e.g., a, = 0 but & = 1. The correlator wave- form is shifted by 90 deg at the second step, therefore the no-noise outputs will be A‘ if a2 = 0 and B’ if a, = 1 instead of A and B, respectively (Fig. 1). Since the receiver estimates a2 = 0 whenever x2 10, the probability of wrongly estimating a, is %. It can be shown that the error in estimating a, will adversely affect the estimation ‘k I . 1 . if of a3,u4, * * . . However, the effect will diminish as k increases. This error propagation does not affect the performance of the system, since one wrong estimation is enough to cause an error in the measurement of the TQF. How- ever, since & affects all (x,, yn) for n h k, an estimate of a k based on {x,, y,; n k} is superior to one which is performed at the kth step, and therefore depends on ( x k , yk) only. Assume, as before, that E and let be the correlators output vectors anc. t - e TOF binary vector respectively. By Bayes rule we have and the maximum-likelihood estimate is the binary vec- tor a* which maximizes (9) or equivalently minimizes where s k y r k , k = 1, . . * , N can be expressed in terms of a (Eqs. 4 and 5). Note that the &, i = 1, . . . , k - 1 that appear in Cyk (and hence in s k and r k ) are on-the-spot estimates which determine whether the correlator waveforms are shifted in the next step. They depend on the xi's alone and not on a, and are known to the receiver when P (a) is evaluated. Since we have to choose the most likely of 2N possibili- ties, the complexity grows as 2N, and the method is not practical for large N . I The contribution of a k to (sn, r,) is halved at every successive step, since its coefficient in an is proportional to 2(k-n) (Eq. 5) . In other words, the value of a k affects the correlator outputs for all n k; however, this effect diminishes exponentially as n increases. Thus, there exists some integer V, which depends on the signal-to-noise ratio, such that the contribution of a k to (sn, r,) for n > k 4- v is negligible. We therefore can approximate Eq. (5) by for all k > v + 1. Thus each (sk, r k ) depends on a k and the previous v com- ponents and we have a finite state machine with 2" states, corresponding to all possible binary vectors (ak-1, . . * , a k - v ) , and two outputs per state depending on a k . The progress of this machine, during few successive steps can be de- picted by its trellis diagram (Fig. 2). STATE STEP k STEP k + 1 STATE a k-1 '.. 'k-3 a k = O 'k+l = k- l ak+l ... a - 0 0 0 0 0 1 0 1 0 0 1 1 1 0 0 1 0 1 1 I O I l l a k = 1 'k+l = e 77 During step k, each state (ak -1 , - . , ak-v) can advance to one of two states (0, a k - 1 , . * , a k - v+J or (1, ak-1 , * . e , ak-v+ lJ depending whether a k is 0 or 1. The corresponding corre- lation outputs (Sk, Tk) will depend on the starting state as well as on a k . The similarity to convolutional coding with a constraint length of v is immediate. Many decoding procedures have been developed for convolutional codes, however, for short constraint length ( v < 10) the Viterbi algorithm (Ref. 2) is the most efficient, and is actually a maximum- likelihood estimate of the truncated estimation problem (Ref. 3). The Viterbi algorithm can be briefly described as fol- lows: With every state we associate a metric (accumu- lated likelihood function up to this step) and a survivor (the most likely sequence leading to this state). At step k, each state ?k = ( a k , . . . , a k - v + l ) can be reached from two states S6,-, = (ak -1 , * - , ak-v+ l , a), where 8 is 0 or 1. After the received signal is correlated to yield (xk, q k ) , we com- pare the two possible ways to reach e k and keep the most likely of them, by properly updating the metric and the survivor of g k . This is done for each one of the 2” states, and is repeated every step. The final decision is made after the Nth (last) step, by comparing the 2” metrics and selecting the survivor of the largest, to yield the most likely estimate of a. Thus a ranging receiver based on the Viterbi algorithm can yield a (truncated) maximum-likelihood performance with a complexity of 2”, which is independent of the number of components N to be estimated. ~ ~ ~ ~ ~ @ S The signal is received in the presence of additive white gaussian noise of zero mean and spectral density No. If the signal power is S and the integration time of each component is Tk = 7, k = 1,2, . * , N the probability that the sequential receiver will correctly estimate the whole sequence {al, * ,aN}, is given by If e- E #! 0 the performance is degraded; however, we can assume that the integration time of the hi hest fre- quency component is long enough to obtain 8~ E . An analytic expression for the performance of the maximum- likelihood receiver or for the Viterbi algorithm cannot be obtained in closed form. Therefore, the error probabilities for various signal-to-noise ratios were obtained by com- puter simulations. The noise was generated by a multi- plicative congruential generator (Ref. 4), and quantized in steps of u2/32, where 02 is the noise-to-signal ratio. The results for the maximum-likelihood estimate of 10 components ( N = lo), and a truncation to v = 5 are shown in Fig. 3 together with the performance of the se- quential receiver. The improvement gained by maximum- likelihood estimation compared to sequential estimation is 1.5 dB for the measured range of error probabilities. This improvement is also achieved by the Viterbi algo- rithm (with v = 5 truncated memory and therefore a Y = 5 2A where 7 smaller complexity), for signal-to-noise ratios which yield P E < 10-2. maximum-likelihood techniques; however, the complexity grows exponentially with the number of components N needed to determine the range unambiguously. The sug- gested method, which is based on the Viterbi decoder for convolutional codes, performs like the maximum- likelihood receiver while having a finite complexity inde- pendent of N . The improvement in signal-to-noise ratio, compared to the present receiver, is 1.5 dB. s A truncated maximum likelihood receiver for sequen- tial ranging has been presented. The performance of the sequential ranging system can be improved by using 1. Goldstein, R. M., “Ranging With Sequential Components,” in The Deep Spuce Network, Space Programs Summary 37-52, Vol. 11, pp. 4 U 9 . Jet Propulsion Laboratory, Pasadena, Calif., July 31, 1968. 2. Viterbi, A. J., “Error Bounds for Convolutional Codes and an Asymptotically Optimum Decoding Algorithm,” IEEE Trans. Infor. Theory, Vol. IT-13, No. 2, April 1967. 3. Forney, G. D., “Review of Random Tree Codes,” Final Report for NASA Ames Research Center under contract NAS2-3637. 4. Chambers, R. P., “Random Number Generation,” IEEE Spectrum, February 1967. J. Y. Otoshi Communications Elements Research Section This article presents a simple formula and graph useful for predicting the transmission loss of a circular hole array in a metallic flat plate having either a 60- (staggered) or 90-deg (square) hole pattern. The formula is restricted to the case of an obliquely incident plane wave with the E-field polarized n o m 1 to the plane of incidence. The theoretical formula was experimentally verified by testing samples having hole diameters vaying from 1.6 to 12.7 mm, porosities varying from 10 to 51 %, and plate thicknesses vaying from 0.08 to 2.3 mm. The agreement between theory and experiment was typically better than 1 dB at S-band and 2 dB at X-band. To those involved with the development of low-noise antennas for deep-space communications and radio as- tronomy, the subject of leakage through antenna mesh materials is of great interest. This subject is also of interest to those concerned with microwave radiation hazards due to leakage through various types of mesh materials. Meshes have many applications; some examples are reflective surfaces on antennas, Fabry-Perot inter- ferometers, microwave oven doors, RF screen rooms, and RF protective garments. meshes (Refs. 1-5). Using existing experimental data, Mumford (Ref. 4) has derived an empirical formula and nomograph for predicting transmission through wire grid meshes at normal incidence, To this author’s knowledge, a similar type summary of experimental results has not been made for transmission through flat plate meshes having round hole perforations. It is the purpose of this article to present a simple theoretical formula applicable to perforated flat plates. This formula is verified by experimental data obtained by both free-space and wave- guide measurement techniques. Meshes are usually of two types: (1) meshes formed by wire grids, and (2) meshes formed by round holes in a flat metallic plate. A significant amount of experi- mental and theoretical work has been done on the study of microwave transmission properties of wire grid-type For the case of a normally incident plane wave, an array of small holes in a metallic sheet behaves as an inductive susceptance in shunt with a TEM (transverse . II electromagnetic) transmission line. The normalized sus- ceptance of this array is given as (Ref. 6) b l a B - 3abh, --- Yo rd3 where a,b = spacings between holes (see Fig. 1) d = hole diameter ho = free-space wavelength Equation (1) is valid when the hole diameter is small compared to the hole spacings a and b. For the case where a and b approaches d, the corrections described in Ref. 6 can be applied. Generalizing Eq. (1) to the case of an obliquely inci- dent plane wave with the E-field polarized normal to the x x Q -0 0 0 plane of incidence and also accounting for the effect of plate thickness, the approximate expression for trans- mission loss is where 'ei = angle of incidence t = plate thickness and d,a,b << An. The last term of Eq. (2) is the plate thickness correction term given by Marcuvitz (Ref. 7 ) and was derived by treating the small hole as a circular wave- guide beyond cutoff. It should be pointed out that Marcuvitz's transmission loss equation (Ref. 7 ) for a thick circular aperture in rectangular waveguide and Eq. (2), above, for the array of small holes give results that differ by about 5 dB. The percent porosity of a mesh is usually known or can be easily calculated for a 60- or 90-deg hole pattern from the expression where a and h are the dimensions shown in Fig. 1. Substitution of Eq. (3) into Eq. (2) results in an equation from which a general plot of the type given in Fig. 2 can be made. The curves in Fig. 2 are useful for pre- dicting transmission loss of meshes having either the 60- or 90-deg hole pattern, but are valid only for the case where the E-field is normal to the plane of incidence. Using the mesh on the DSS 14 64-m (210-ft) antenna as an example, at 2295 MHz and 30-deg angle of inci- dence, the following values are applicable: P , = 51.0 d - = 0.0365 i o COS f3i = 0.866 t -J = 0.480 Then the intersection of 2 = 51 (0.0365) (0.866) = 1.61 and t / d = 0.48 on the graph gives a transmission loss of 43 dB. An estimate of the contribution to zenith antenna 10-1 2 4 2 4 6 101 Fig. 2. ~ran$mi$$ion loss curves for perfora pattern (E-field normal 60- or 90-deg ho noise temperature due to leakage through this mesh would be 0.06 K. eri Table 1 is a summary of experimental results obtained by waveguide and free-space measurement techniques. The mesh test samples were made from flat aluminum plates having 60- and 90-deg hole patterns, hole diam- eters varying from 1.6 to 12.7 mm, porosities varying from 10 to 51%, and plate thicknesses varying from 0.08 to 2.3 mm. Waveguide measurements were performed by JPLl using a network analyzer and techniques described in IMeasurements were performed by R. B. Lyon of the JPL Com- munications Elements Research Section. ‘d = .o .9 .8 .7 .6 .5 .4 .3 .2 .1 1 .o 0.9 0.8 0.7 0.6 0.5 0.4 0.3 0.2 0.1 0 Ref. 8. Test samples preyiously described in Ref. 9 were remeasured. Some of these samples are shown in Fig. 3. The accuracy of the waveguide measurements is esti- mated to be - C O S dB. For additional verification, free-space measurement data from a Dalmo Victor Co. report (Ref. 10) are in- cluded. The estimated accuracy of the free-space mea- surements was stated to be about k0.5 dB. Theoretical values for Table 1 were calculated through the use of Eq. (2). It can be seen that the agreement between theoretical and experimental results is typically better than 1 dB at S-band and 2 dB at X-band. Some of the deviations at X-band are attributed to increasing inaccuracy of the waveguide method when the number of basic cells in the waveguide sample becomes small and the sample is not imaged to the waveguide walls. issie Measured minus theoretical transmission, dB Hole pattern, deg Approximate porosity, % Theoretical transmission,a dB t d Comments 0.04 0.04 0.07 0.07 0.1 0 0.1 0 10.0 22.6 10.0 22.6 10.0 22.6 0.345 0.345 0.1 9 7 0.1 97 0.138 0.138 50.3 43.3 40.8 33.8 35.8 28.8 -2.3 -2.5 0 -0.5 -0.9 -1.0 Dalmo Victor data (Ref. 10) 0 60 30 33 40 5 0 51 51 25 50 25 50 51 25 50 48.3 54.9 48.9 51.5 30.2 42.9 41.3 35.4 34.4 28.4 28.8 30.6 24.3 -0.4 0.1 -0.8 -0.8 -0.9 -0.7 -0.4 -0.5 -0.5 -0.1 0 -1.3 -1.0 0.025 0.025 0.025 0.025 0.038 0.038 0.051 0.051 0.076 0.076 0.076 0.101 0.101 0.025 0.025 0.051 0.051 0.076 0.076 0.101 0.101 0.400 0.632 0.496 0.632 0.085 0.480 0.316 0.316 0.21 1 0.21 1 0.227 0.158 0.158 0.632 0.632 0.3 16 0.3 16 0.21 1 0.21 1 0.158 0.158 DSS 14 antenna 35.1 60 25 50 25 50 25 47.3 25 50 57.5 51.4 41.3 35.3 34.5 28.9 30.3 24.3 0.7 -1.2 1 .o 0.1 -0.5 0.4 -1.3 -1.1 90 35.1 0.045 0.045 0.089 0.089 0.134 0.134 50 50 30 40 51 51 0.048 0.256 0.400 0.496 0.085 0.480 28.1 34.8 37.7 38.3 19.7 32.3 2.8 2.8 -1.7 -1.0 - 2.0 -1.3 38.5 60 DSS 1 64-m antenna diameter and spacings. ig. 3. 30 test san The waveguide test samples for the data in Table 1 were cut so that the hole pattern would be oriented with respect to the E-field as indicated in Fig. 3. Tests on some samples at other orientation angles showed that for practical purposes, the transmission loss is independent of the orientation angle (see Ref. 8, p. 56). It should be re-emphasized that the data presented in this article are applicable to the free-space configuration where the E-field is polarized perpendicular to the plane of incidence. At present, there does not appear to be published experimental or theoretical data on the trans- mission loss of perforated plates for the parallel polari- zation case. Based strictly on an equivalent plane sheet theory, however, one would not expect the transmission losses of the perforated plates for the two polarization cases to differ by more than 3 dB when the angles of incidences are less than 30 deg. Good agreement has been obtained between experi- mental and theoretical values for transmission losses of perforated flat plates having various hole diameters, porosities, and plate thicknesses. The theoretical formula is restricted to the case of an obliquely incident plane wave where the E-field is polarized normal to the plane of incidence. Even with this stated restriction, the formula and curves presented in this article are useful for predicting leakage through antenna surfaces and RF shields. 1. Broek, G. J., and Vooren, J., “On the Reflection Properties of a Periodically Supported Metallic Wire Grating with Rectangular Mesh Showing Small Sag Due To Its Own Weight,” IEEE Trans. Antennas a d Propagation, Jan. 1971 (in press). 2. Kaplun, V. A,, et al., “Shielding Properties of Wire Screens at SWF,” Radio Engineering and Electron Physics, Vol. 9, pp. 1428-1430, 1964. 3. Kontorovich, M. I., et al., “The Coefficient of Reflection of a Plane Electro- magnetic Wave from a Plane Wire Mesh,” Radio Engineering and Electron Physics, pp. 222-231, Feb. 1962. 4 4. Mumford, W., “Some Technical Aspects of Microwave Radiation Hazards,” Proc. IRE, p. 445, Feb. 1961. 5. Decker, M. T., “Transmission and Reflection by a Parallel Wire Grid,” J . Res. NBS, Sec. D, Vol. 63D, No. 1, pp. 87-90, Jul.-Aug. 1959. 6. Culshaw, W., “Reflectors for a Microwave Fabry-Perot Interferometer,” ZEEE Trans. Microwave Theory and Techniques, Vol. MTT-7, p. 226, Apr. 1959. 7. Marcuvitz, N., Waveguide Handbook, M. I. T. Radiation Laboratory, Vol. 10, p. 409. McGraw-Hill Book Co., Inc., New York, 1951. 8. Otoshi, T. Y., and Lyon, R. B., “Improved RF Calibration Techniques: A Study of the RF Properties of the 210-ft-diam Antenna Mesh Material,” in The Deep Space Network, Space Programs Summary 37-66, Vol. 11, pp. 52-57. Jet Propulsion Laboratory, Pasadena, Calif., Nov. 30, 1970. 9. Otoshi, T., “RF Porosity Studies,” in Supporting Research and Advanced Development, Space Programs Summary 37-20, Vol. IV, pp. 135-137. Jet Propulsion Laboratory, Pasadena, Calif., Apr. 30, 1963. 10. Pon, C. Y., Final Report on Transmission through Perforated Reflectors and Loss Due To Paint for AASDSZF Program. Dalmo Victor Co., Belmont, Calif., Dec. 28, 1961. uning Serv C. Finnie Communications Elements Research Section Servo design details are described for a prototype hydrogen maser cavity tuner for use with the hydrogen m e r frequency standards developed at the Jet Propulsion Laborato y. This article is a continuation of a tuner design dis- Where it is desired to remove a linear temporal change or drift in the input parameter, the count direction may be double switched to give cussion initiated in Ref. 1. A period counter forms the basis of the tuning system. It has a sampled transfer characteristic given by W 7 0 + (T&+l - T4n+z - T411+3 + 747~+4) n=o (1) (3) W 1 Tout = - ( T o + ETm n=O S where Tout = output count T~ = input variable at sampling time NT T~ = initial value The counter will perform a summation of difference measurements if the count direction is reversed after each measurement, and the count is sampled after alternate measurements : Alternate adjacent samples are subtracted and the differences are accumulated in a buffer which samples the counter at a 27 sampling rate. A diagram of the com- plete servo is shown in Fig. 1. The maser section of Fig. 1 represents the cavity pulling relationship described previously (Ref. 1). The output frequency f o of the maser is modulated by switching between two cavity pulling factors K , and K,. The switch sequence is that of Eq. (3). Therefore, changes in the maser output frequency will be coherently summed in the updown counter of the tuner, while static and linear changes in the maser fre- quency will produce zero and zero-time-average counts in the tuner, respectively. 6 (n + 1 ) ~ - HYDROGEN MASER avify tuner servo The receiver divides the maser frequency, fo, by the ratio ( K 3 ) between the maser (1420.405 MHz) and the voltage-controlled oscillator (VCO) (100 MHz) of the receiver. The 100-MHz VCQ is then mixed with the reference oscillator f T to produce a nominal 100-s beat period Tb. This period is measured in the tuner by count- ing a 1-kHz internal clock for the interval between zero crossings of Tb. The proportionality constant K , between the frequency and period variable is given by where Tb average is much greater than modulation or noise produced on T b . In the count sequence, the counter counts up for the first period, allows the dissociator pressure to reach equilibrium for one period, then counts down the next. After completion of the down count, the residual in the counter is transferred to a digital buffer and an analog- to-digital converter. The voltage from the analog-to- digital converter in turn is applied to a varactor which tunes the microwave cavity. The sequence is then in- verted with the down count coming first. K 5 is derived from the counter clock rate, the analog amplifier of the digital-to-analog converter, and the position of the counter digits selected for analog conversion. The last gain constant K , is the sensitivity of the tuning varactor to its control voltage from the digital-to-analog converter. The open-loop transfer function for the system becomes where 'a is the loop gain. The closed-loop transfer func- tion (sampled) becomes ae-2sr f c - - _ f n 1 - (1 - a)e-2sr and the response to the input step (sampled) is The convergence of the loop is, therefore, a linear decreasing staircase function. Figure 2 is a graph of the loop error as a function of time and loop gain. A simplified design criteria is given by a l x m z ) N E where v1 = allowable variation of maser long-term stability u2 = measurement variation of T b E = change in maser output frequency for the least significant bit of the digital-to-analog converter The loop gain is approximately equal to the input measurement of Tb transferred to the output frequency. Hence, if the reference oscillator is one hundred times 100 10-1 10-2 1 o - ~ 10-2 10-1 1 00 101 TUNING TIME, days 102 ig. In more noisy (of/ f) than the hydrogen maser, a degradation in tuning accuracy must be accepted or tuning times on the order of 40 days are required to realize the inherent accuracy of the maser. The above assumes the noise introduced into the maser output by the tuner to be 10% of the maser 100-s stability. 1. Finnie, @., “Frequency Generation and Control: Atomic Hydrogen Maser Fre- quency Standard,” in The Deep Space Network Progress Report, Technical Report 32-1526, Vol. I, pp. 73-75. Jet Propulsion Laboratory, Pasadena, Calif., Feb. 15, 1971. 88 M. S. Reid Communications Elements Research Section The system operating noise temperatures of the S-band polarization ultra cone at DSS 11 (Pioneer Deep Space Station), the S-band research operational cone at DSS 13 (Venus DSS), and the polarization diversity S-band and multi-frequency X-band/K-band cones at DSS 14 (Mars DSS) are reported for the period October 1, 1970 through January 31, 1971. The system operating noise temperature performance of the low noise research cones in the Goldstone Deep Space Communications Complex (GDSCC) is reported. The operating noise temperature calibrations were per- formed with the ambient termination technique (Ref. 1). The cones on which this technique was used during this reporting period are: (1) S-band research operational (SRQ) cone at DSS 13. (2) S-band polarization ultra (SPU) cone at DSS 11. (3) Polarization diversity S-band (PDS) cone at DSS 14. (4) Multi-frequency X-band/K-band (MXK) cone at DSS 14. The averaged operating noise temperature calibrations for the various cones, and other calibration data, are presented in Table 1. The reporting period for the SRO and SPU cones is October 1, 1970 (day 274) through January 31, 1971 (day 31). The reporting period for the PDS cone is from its initial operation on the 64-m an- tenna, September 15,1970 through January 31,1971. The reporting period for the MXK cone is from its initial installation on the 64-m antenna, March 15, 1970 through February 28, 1971. The calibration data were reduced with JPL computer program number 5841000, CTS20B. Measurement errors of each data point average are recorded under the appro- priate number in Table 1. The indicated errors are the standard deviation of the individual measurements and Station Cane I Configuration E v e m o d e ~ I DSS 11 DSS 13 DSS 14 SPU sxo PDS MXK I Diplexed I Frequency, MHz Maser serial number 2295 2295 2388 2295 7840 8448 96S5 9682 96S2 96S3 150x1 150x1 Maser temperature, K Maser gain, dB 3.7 5.2 5.2 4 10 7 55.8*0.23/0.10 51.0?i).53/0.17 37.1 * 1.1810.10 5 3 . 4 ~ 0 . 5 0 / 0 . 1 0 37.0&1.7/0.55 41.3 6 measurements 10 measurements 137 measurements 5 measurements 10 measurements of the means, respectively. They do not include instru- mentation systematic errors. The averages were com- puted using only data with the system operating noise temperatures of the MXK cone at 8427 and 8448 MHz will average approximately 3 K lower than the average temperature at 7840 MHz. ~~ Fallaw-up noise 0.06 ?Z 0.003/0.002 temperature con- 6 measurements tribution, K System operating 18.3?0.10/0.06 noise temper- 4 measurements 0.1 1 * 0.0610.02 0.55 2 0.09/0.05 0.01 * 0.07/0.05 1.7 * 0.60/0.20 0.58 9 measurements 137 measurements 4 measurements 10 measurements 1 measurement 16.5&0.37/0.12 16.4k0.7610.08 24.1 *0.55/0.05 27.7*1.3/0.51 2 3 . 6 2 0 . 2 5 9 measurements 137 measurements 4 measurements 13 measurements 1 measurement 0.1 K. (1) Antenna at zenith. z 25 0 - Z 9 (2) Clear weather. Z $ 20 g z 5: 15 (3) No RF spur in the receiver passband. +tu 2: Z 10 (4) Probable error of computed operating noise tem- The system operating noise temperatures of the SPU cone at DSS 11 are plotted in Fig. 1 as a function of time in day numbers. Similarly, Figs. 2 and 3 are system operating noise temperatures of the SRO cone at 2295 z and 2388 MHz, respectively. Figure 4 shows the PDS cone data and Fig. 5 the MXK cone data. 0 0 X % % * In all the figures, data that satisfy the four conditions stated above are plotted as asterisks, while data that fail one or more conditions are plotted as circles. The one exception is the single data point at 8448 MHz in Fig. 5 which is plotted as a square; the remainder of the data in the figure is for 7840 M z. It is expected that DAY IN 1970 35 Y ui $ 30 L w 25 0 Y 0 - Z (3 Z3 z 20 cz 30 cd 3 + VI 35 - 4 $2 2 01 Z ? + w 25 3 + VI 2: 2 Z 0 15 10 20 270 300 330 360 025 0 050 100 150 200 250 300 DAY NUMBER tern operating noise DAY IN 1970 30 25 20 15 10 080 140 200 260 320 015 O i DAY NUMBER ~ p e r a ~ i n ~ wise ~ernperature of 1. Stelzried, C. T., “Improved RF Calibration Techniques: Daily System Noise Temperature Measurements,” in The Deep Space Network, Space Programs Summary 37-42, Vol. 111, pp. 25-32. Jet Propulsion Laboratory, Pasadena, Calif., Nov. 30,1966. . I I J. J. Volkoff SFOF/GCF Development Section Light emitted from a cathode-ray tube (CRT) is comprised of reflected environ- mental light incident upon the CRT system and light generated at the phosphor screen and transmitted through the CRT system. The total photon energy which leaves the system and those energies which are dissipated in the various elements of the CRT system are deriued. An expression for the contrast ratio of the CRT system is also presented and discussed. - . Ink Photons generated by the impingement of electrons upon the phosphorescent material of the phosphor screen pass through the cathode-ray tube (CRT) system as out- put light. A fraction of the photons from the environment (ambient light) incident upon the CRT system is reflected from the CRT system and contributes light to the gen- erated output light. The combination of these two outputs makes up the total light emitted from the CRT system. A common CRT system is comprised of a phosphor screen, a glass faceplate, and a filter screen which is utilized to enhance the contrast ratio of the CRT system. When light, either generated at the phosphor screen or entering from the ambient, encounters an element of the CRT system, its energy is partially transformed into heat due to the energy-absorbing nature of the CRT element. The remaining energy is partially transmitted if the ele- ment is not opaque, and the residual portion is reflected. Conservation of the total energy for this process can be expressed by where &(A) , ~ ( h ) , and p ( X ) are, respectively, the absorp- tivity, transmissivity, and reflectivity coefficients at wave- length h of the element (Ref. 1). Since light undergoes an infinite number of reflections in a CRT system, it follows that there occurs an infinite number of sequential events differentiated by absorp- tion, transmission, and reflection mechanics. These series describe the total energy transformations within the CRT system. The analysis in this article is divided into two sections, the energies resulting from the environmental light inci- 9 dent upon the system, and the energies resulting from the light generated by the system. In these analyses, all the energies transmitted and reflected from the system, and all the energies dissipated by absorption processes by the phosphor screen, the glass faceplate, and the filter screen elements are considered. 1. ies vir Consider a coherent beam of light at wavelength incident upon the CRT system. The energy E (A) of this light is (2) E (A) = n (A) h v where n(A) represents the number of photons at A, h is Planck's constant, and v is the frequency of light. The fre- quency may be expressed as the velocity of light in the medium divided by the wavelength. Using c and A as representing the velocity and wavelength of light, respec- tively, under vacuum conditions, the frequency of light may then be written as C v = - A (3) light. While [ 1 - p, (A)] T f (A) E (A) enters into the face- plate, [l - T, (A)] [l - p, (A)] T f (A) E (A) is absorbed by the faceplate and 7, (A) [l - p, (A)] T f (A) E (A) = E, is inci- dent upon the surface of the inner faceplate (phosphor screen side). Of E,, the portion p, (A) E , is reflected and [ 1 - pg (A)] E , is the energy finally incident upon the phosphorescent material. The phosphorescent material absorbs and reflects inci- dent energy since its transmissivity coefficient is essen- tially zero. Thus, for every beam incident upon its surface, a fraction [l - p, (A)] is absorbed, and ps (A) is reflected. Subsequently, a portion of the reflected beam is re- reflected from the inner surface of the faceplate back for another reflection from the phosphorescent material. Thus, an infinite series of subsequent reflection and absorption energy sequential events occurs. It can be shown that the series of energy reflections and absorp- tions can be expressed in the form such that the sum total energy of E , which is reflected from the faceplate and phosphor screen into the faceplate at the inner side is F , (A) E,, where Since the velocity of light in standard air is so nearly equal to the velocity of light in vacuum, for practical pur- poses, the wavelength of light in the two mediums shall be considered equal and shall not be differentiated in The total amount of energy E , which is absorbed by the phosphorescent material is F , (A) E,, where [1 - p,(A)l [1 - Ps (41 1 - P, (A) Pa (A) this study. Fs (A) = (5) The fraction of the total light energy transmitted through the filter screen and incident upon the faceplate is T f (A) E (A), where T f (A) is the transmissivity coefficient at A of the screen and the direction of incidence is taken to be normal. The remaining energy ./(A) E ( A ) is ab- sorbed by the screen. Since the reflectivity coefficient of a filter screen is negligibly low, the energy absorbed by the screen can be rewritten as [1 - T f (A)] E (A). Continuing with the light incident upon the glass face- plate, the portion p, (A) r f (A) E (A) is reflected from the outer surface of which p, (A) T% (A) E (A) is transmitted back through the screen as the first term of the total light output from the system contributed by the environment and F , (A) + F , (A) = 1. As the reflected beam F, (A) E , passes through the face- plate, [ 1 - T~ (A)] F , (A) E , is absorbed by the faceplate, and [ 1 - p, (A)] T, (A) F , (A) E , emerges from the outer sur- face of the faceplate toward the filter screen. However, pg (A) T, (A) F , (A) E , is the portion of energy reflected from the outer surface of the faceplate toward the phosphor screen for another series of reflections and dissipations. By continuing similar arguments for the reflected beam, series expressions for these energies which are dissipated and transmitted in the various elements of the CRT sys- tem can be developed. These series are given below for a monochromatic beam of light. The total energy Lo (.A) emerging from the CRT system is the series of energies formulated from the analysis above, as follows: which can be written as issipated into the Filter Screen The total energy Lf (A) dissipated into the filter screen in the form of thermal energy is resolving to The total energy L, (A) absorbed by the glass faceplate is written as which becomes ner by the or Screen The total energy L, (A) absorbed by the phosphor screen is similarly a series of energies which, when com- bined, results in the following equation: Energy is conserved since i@S Let a coherent beam of light having wavelength A be generated at the surface of the CRT phosphor screen by the impingement of electrons upon the phosphorescent material. The energy E, (A) of this beam of light may be expressed by E , (A) = n, (A) h Y (14) where n, (A) is the number of photons generated. Of the beam of light which impinges upon the inner surface of the glass faceplate, the portion p, (A) E , (A) is reflected and the remaining portion [l - p, (A)] E, (A) passes into the faceplate. Direction to the faceplate sur- face is assumed to be normal. The reflected portion inci- dent upon the phosphor screen is partially absorbed by an amount equal to [ 1 - p, (A)] p, (A) E , (A) and the other portion ps (A) p, (A) E , (A) is reflected back upon the inner facepIate surface. This reflected beam undergoes a similar reflection and absorption sequence between the faceplate and phos- phor screen, resulting in an infinite number of sequential events. The combined series of energies passing into the faceplate may be expressed by the form K , (A) E (A), where Similarly, the total energy absorbed by the phosphores- cent material by all these primary sequential reflections and absorptions is K s (A) E , (A), where Of K , (A) E, (A), the portion absorbed by the faceplate is [ 1 - T , (A)] K, (A) E , (A) and the remaining portion E,. (A) = T, (A) K , (A) E , (A) becomes incident upon the outer surface of the faceplate. Of &(A), the portion [l - pg(A) ] E,(A) passes onto the fiIter screen and p, (A) E , (A) is reflected back toward the inner surfaces of the faceplate. The amount 7, (A) p, (A) E,. (A) incident upon the inner surface is then divided into two light beams, one beam passing through the inner surface to the phosphor screen for sec- ondary reflections back through the system, and one beam reflected to the outer surface. It can be reasoned that the primary light beam K , (A) E , (A) entering into the faceplate undergoes an infinite number of reflections from the inner and outer surfaces of the faceplate, and for each sequence of reflec- tions, some fraction of light is passed toward the phos- phor screen or toward the filter screen respective of the inner or outer faceplate surface. In the interim of these reflections within the faceplate, a series of energies is absorbed by the faceplate. Thus, three total energies from the primary light beam entering the faceplate from the inner side occur: (1) a total energy reflected from the face- plate toward the phosphor screen Eph (A), (2) a total energy passed from the faceplate toward the filter screen Et (A), and (3) a total energy absorbed by the glass faceplate E, (A). These series of energies are combined and formu- lated as follows: Of the beam E f (A), the portion [l - T f (A)] E f (A) is ab- sorbed by the filter screen, and ~f (h) E (A) finally emerges from the CRT system as light. Continuing back at the inner surface of the faceplate, after Eph(A) impinges upon the phosphor screen, a por- tion [ 1 - p,. (A)] Eph (A) is absorbed by the phosphores- cent material, and the remainder ps (A) Eph (A) is reflected back toward the faceplate. This reflected portion under- goes an infinite number of reflections and absorptions between the faceplate and phosphor screen similar to that undergone by E , (A), which was described in the above analysis. Thus, K , (A) p, (A) Eph (A) is absorbed by the phosphorescent material, and K , (A) p, (A) E (A) enters into the faceplate, which in turn undergoes interreflections between the inner and outer surface similar to the above analysis (Eqs. 17-19). Again, a portion of this energy entering the faceplate will be divided into three portions: (1) a portion reflected back to the phosphor screen; (2) a portion absorbed by the faceplate; and (3) a portion pass- ing from the outer surface of the faceplate, being partially absorbed by the filter screen and the remaining energy finally emerging from the CRT system as a term of the series of light output. Another set of terms can be similarly developed to fur- ther the expression of the energy terms evolved from the interreflections of the phosphor and faceplate surfaces. The energy series for each element are given below. The total energy ru (A) of the light emerging from the CRT system is expressed as which becomes where into the Filter Screen The total energy rf (A) dissipated into the filter screen is which reduces to 6 J L. II late The total energy r, (A) absorbed by the glass faceplate may be formulated in a series of energies as follows: which resolves to y the ~ h Q s ~ h ~ ~ Screen The total energy rs (A) absorbed by the phosphor screen is similarly reduced from a series to the following form: Again, energy is conserved since ies The total energies produced by both the environmental light incident upon the CRT system and the light gen- erated by the CRT system in the forms of light output and eneigy absorptions by the CRT elements can-be obtained by summating the energies produced for each form. Let d (Et )n at A represent the total light energy out- of A and A + dA. Thus, the incremental energy d (Et),,/dh put for the considered sample in the wavelength region (31) at A is the sum of the monochromatic energies Lo (A) and ro (A) (Sections 11-A and 111-A). By integrating this result over a given spectrum (A, to h2), the total light energy out- put from the CRT system over that spectrum is obtained. Thus, the total light energy output and total energies (Et ) , = SA’ [Ls (A) + rs (A)] c lA (32) hl absorbed in each element can be written as follows. . ’BOB The contrast ratio of the CRT system is defined as the quotient of the maximum luminance divided by the mini- mum luminance. Luminance output from a GRT system can be varied by modulating the flux of electrons incident ( E ~ ) ~ = W~ (A) + To (A)] dx (29) hl upon the phosphor screen. By assigning G to represent the maximum degree of photometric modulation or gain of the CRT system, it follows that the maximum and mini- mum number of photons generated by the electron beam are related by G according to the equivalent The maximum light energy output can be expressed, from Eq. (29), as The minimum light energy output from the CRT system is then Converting these energies to luminance by introducing the luminous efficacy K , the contrast ratio C, of the CRT system, including environmental light, can now be written as follows: It is apparent that Eq. (36) is too involved to be used to practically estimate the contrast ratio performance of a CRT system. For reasons of simpliiication, it shall be assumed that the distribution of n (A) and n, (A) with respect to A is uniform. This permits the assignment of an average value over the considered spectrum for each optical property of the various elements of the CRT sys- tem. For example, the average transmissivity coefficient yf of the filter screen under this condition can be written as ( (A) dx Equation (36), under these conditions, can then be modi- fied to the form - - Lo +To c, = - ( s o ) m a x L, + -G where the bar over each total energy s i d e s that aver- age values for the optical properties are used. By including only the primary reflections (first-order terms) associated with the primary beams of the environ- mental light and generated light of the CRT system, an expression for the contrast ratio (C,)’ which includes only these first-order terms can be derived (Ref. 2). This expres- sion is given as (39) T~ k L + rmax (C,)’ = r m a x T f k L + G where and L and r are the luminance levels of the environmental light and generated light, respectively. Also, the spectral distributions of L and r are assumed to be equal. It is of interest to compare the contrast ratio given by Eq. (38), which includes the series of interreflection effects upon the elements, with the contrast ratio given by Eq. (39), which considers only the first-order effects of the primary beam. Representative values for the optical properties of a typical CRT configuration which utilizes a P4 phosphor is given in Table 1 (Ref. 2). Assume that the selected CRT has a filter screen with a mean transmissitivity Sf == 0.50. By substituting the values for the properties (Table 1) into Eqs. (38) and (39), esti- mated contrast ratios, including fractional differences, are obtained. These ratios are shown in Table 2 for three dif- ferent environmental lighting conditions. It can be seen from Table 2 that the value of contrast ratio using Eq. (39) results in a slightly higher contrast ratio value than that which results from using I Property I Value I Lighting condition Dark I Dim I Brighf Ratio Environment-to-generated- E, (Eq. 38) (E,)’ (Eq. 39) light ratio (L/rmax) IT, - (E)’] /E, In the case where a filter screen is not used, the results obtained from the two equations are about double those shown in Table 2. It may be concluded that Eq. (39) may be used to obtain a reasonably accurate estimate of the contrast ratio capability of a CRT system. The results obtained will be slightly optimistic. For the CRT configuration described in Table 1, the light output and energies dissipated in the various CRT elements may be estimated from Eqs. 28-31. Consider a practical intensity of environmental light to be 10% as great as the light generated by the CRT. Under these conditions, the total photon energy (incident upon and generated by the CRT system) would be porportioned as follows: 32% is transmitted and reflected from the CRT as light output, 36% is dissipated by the filter screen, 40% is absorbed by the glass faceplate, and 2% is absorbed by the phosphorescent material. For a CRT system which has no filter screen, under the same light conditions given above, the total photon energy would be porportioned as follows: 65% is light output, 42% is absorbed by the glass faceplate, and 3% is absorbed by the phosphorescent material. 1. Born, M., and Wolf, E., Principles of Optics. Pergamon Press, London, England, 2. Volkoff, J. J., “Contrast Ratio Determination for the SFOF Video Image Dis- play,” in The Deep Space Network, Space Programs Summary 37-65, Vol. 11, 1959. pp. 91-93. . I C. F. leahey SFOFIGCF Development Section The Mark I I I A Simulation Center is capable of simultaneously simulating two spacecraft and three deep space stations using the Univac 1108 and the E M R 6050 computers. The control consoles of the Mark II system were inadequate for con- trolling a simulation of the sine required for the Mark IIIA system. A new control and display system was designed using interactive cathode-ray tube data terminals and high-speed printers. This design upgrades the control and display system for future use in more complex missions. Development, capabilities, and operation of this system are described. The purpose of this article is to describe the devel- opment, capabilities, and operation of the interactive alphanumeric television (IATV) system. This equipment configuration is an assembly of the DSN Simulation Center (Simcen), designated the Mark IIIA S' imcen, which is located in the SFOF. The installation of the IATV system is part of the current development activity that is taking place in the DSN Simcen in preparation for Mariner Mars 1971 and Pioneer F support. This activity was described in Ref. 8. The Mark I1 Simcen configuration was capable of simu- lating one spacecraft and one DSS. The Mark HIIA con- figuration is capable of simultaneously simulating two separate spacecraft and three DSSs. Since the control consoles which existed in the Mark I system were inadequate for controlling a simulation of the size required for the Mark IIIA system, a new control and display system had to be designed. Originally, the plan was to fabricate the necessary control consoles using discrete components. After doing a preliminary design on these consoles, using the requirements specified by the DSN, it was determined that hardware implementation costs would approach the $100,000 mark and that the configuration of the consoles could not be easily upgraded for follow-on projects with greater complexity. The use of interactive cathode-ray tube (CRT) display terminals was decided upon since they would provide the desired 3exi- bility to upgrade the control consoles for more complex missions with few, if any, hardware modifications, and a lesser cost for software changes than if the job were done with discrete components. The IATV display system is functionally diagrammed in Fig. 1. The system contains eight CRT-keyboard display terminals and two printer terminals, and operates from a buffered input/output word channel of the EMR 6050 computer. The equipment for the IATV system was obtained from an outside source and, with the exception of the channel adapter, is standard off-the-shelf equipment. The channel adapter was designed and built for JPL by the manufacturer . EMR 6050 I Each of the eight CRT display stations of the IATV display system is comprised of three physically separate units (Fig. 1): (1) CRT/TV display unit (14 in.). (2) Station controller. (3) Keyboard. Each station controller contains a 1024 byte core memory for storing characters to be displayed on the associated CRT/TV display unit. The station controller is the cen- tral control unit of the station and responds directly to the commands and data received either from the key- board or the computer (through the remote multiplexer). The same type controller is also used as the primary con- troller for the two hard copy printout stations. TRANSFER RATE: 100 kbps BIT-SERIAL CONNECTION TYPICAL DISPLAY STATION MULTIPLEXER STATION CONTROLLER MECHANISM The two nonimpact, high-speed, hard copy printout sta- tions of the IATV display system are implemented using four physically separate interconnected functional units (Fig. 1): (1) Station controller. (2) Printer controller. (3) Nonimpact printer mechanism (two units). The station controller and the printer controller are time- shared by the two printer mechanisms. The operations of each of the five major functional units used in the IATV display system are briefly described in the following subsections. The five major functional units are: (1) EMR 6050 computer. (2) Computer interface channel adapter. (3) Remote multiplexer. (4) Display communications station. (5) Printout station. 6050 computer. The EMR 6050 computer inter- face with the display station is by means of a buffered input/output word channel. A number of data, address, and control signals are involved in the interface between the computer and the channel adapter. The data is trans- ferred in parallel between the computer and the channel adapter, 24 bits at a time. Three 8-bit characters are packed into each 24-bit computer word. The characters can be for display or printout or for control of the sub- systems of the IATV system. a. ZATV interrupts. The IATV display system is as- signed two external device interrupt levels in the com- puter and is capable of generating four distinct interrupt conditions. Two of these conditions are indicated by bits in the channel adapter status word, which will be dis- cussed later. These conditions are as follows : (1) B M = BL interrupt. This condition occurs when the channel adapter receives a BM = BL signal on the BM = BL line from the computer, which indicates that the information block in memory has been filled (input) or that the last word is available (output). (2) End of operation interrupt. This condition occurs when a block read or write operation is terminated by the transfer of all the characters or words, and the channel adapter returns to the not busy condi- tion and has been instructed to return an end-of- operation interrupt. (3) Time out interrupt. This interrupt provides d e channel adapter with the ability to terminate the read operation if a data character is not received from the remote multiplexer within a given time period. This time period is determined by the chan- nel adapter. When a time out occurs, a bit is set in the channel adapter status word. ( 4 ) Interrupt character received interrupt. This condi- tion occurs when the channel adapter receives an interrupt character from a station controller. A bit is set in the channel adapter status word when this occurs. 2. Computer interface channel adapter. The computer interface channel adapter is compatible with the buffered input/output word channel of the EMR 6050 computer and acts as a device controller on this channel. The chan- nel adapter performs four major interrelated functions: (1) Transforms the bit-parallel interface to a bit-serial communications channel and conversely. (2) Manages the flow of data between the buffered input/output word channel and the devices (sta- tions) serviced by the bit-serial communications line (through the remote multiplexer). (3) Responds to the command and control signals gen- erated by the buffered input/output word channel in a functionally and electrically compatible manner. (4) Generates the correct transmission format which is compatible with the “line discipline” used on the bit-serial communications line. a. Commands. The computer can issue the following control commands to the channel adapter: (1) Initiate block write. This command initiates the transfer of a block of data from the computer to the channel adapter. (2) Znitiate block read. This command initiates the transfer of a block of data from the channel adapter to the computer. Enable interrupt character received interrupt. This command enables the channel adapter to transmit an interrupt to the computer when an interrupt character is received by the channel adapter from a station controller. Disable interrupt character received interrupt. This is opposite from command (3). Clear channel adapter. This command resets the logic and data registers in the channel adapter to a known state. Enable end-of-operation interrupt. This command enables the channel adapter to return an end-of- operation interrupt, which was previously described. b. Status conditions. The channel adapter is capable of indicating its status to the computer by means of a status word. Seven status conditions can be indicated and are as follows: (1) Data overrun on input. Set when a write operation is attempted while the channel adapter has a char- acter waiting to be transferred to the EMR 6050 computer. (2) Input transmission parity error. Set by the detection of a character parity error during receive operation. (3) Character in read buffer. Set whenever any charac- ter is placed in the channel adapter input data register. (4) Time out interrupt. Set whenever the time out interrupt (discussed previously) is activated. (5) Interrupt cha?.acter received interrupt. Set when- ever the interrupt character received interrupt (dis- cussed previously) is activated. (6) Remote multiplexer busy. Set whenever the remote multiplexer is performing a block read or a block write operation. (7) Data lost. Set if the channel adapter fails to receive a read command before the channel adapter must begin to assemble the next bit-parallel word. emote multiplexer. The remote multiplexer func- tionally performs as an electronic switch controlled by the computer through the computer interface channel adapter. The “address” character transmitted to the re- mote multiplexer from the computer causes the remote multiplexer to select one of several long line adapter interfaces, which in turn are connected to the “remote” stations (CRT display or printout stations). The address character always precedes the “text” transmitted to the selection station. The previously selected long line adapter interface and, therefore, the corresponding station is auto- matically deselected by the SYN character which always precedes the address character. The remote multiplexer also accepts and responds to control function codes which are effectively addressed to it. The remote multiplexer functionally consists of a data “shift through” register, control logic, function decode logic, a special character generator, several status regis- ters, and a set of long line adapters. Each long line adapter provides two bit-serial interfaces. One remote type station can be connected to each bit-serial interface. Each long line adapter functionally is an integral part of the remote multiplexer and contains two bit positions of the station acknowledge status (SAS) register, four bit positions of the double rank station interrupt status (SIS) register, and the address/select logic for the two bit-serial interfaces provided by each long line adapter. The shift through register in the basic remote multiplexer is the only storage register provided for data transfer in the remote multiplexer. This register introduces a one charac- ter delay in data transfer through the remote multiplexer. The SAS and SIS registers contain data relative to the status of the remote stations, which the computer can sample. a. Operation. The remote multiplexer operates either in the transmit or receive mode. It can either transmit infor- mation to the computer or receive from the computer, but not simultaneously. It is a half duplex device and can “turn around” in one bit time period. The data trans€er rate through the remote multiplexer is 100 kbps. The remote multiplexer is designed primarily as a hard-wired logic multiplexer for data transfer between the computer and the several terminals (stations) connected to the re- mote multiplexer. In addition, it can also: (1) Execute certain control functions received from the computer. (2) Propagate to the computer interrupts generated at any one of the several remote stations connected to the remote multiplexer. (3) Present to the computer status word(s) in response to requests from the computer. isplay communications station. The display com- munications station is comprised of the CRT/TV display unit, the station controller, and the keyboard (Fig. 1). The station recognizes and responds to: remote multiplexer through one printer controller and one station controller (Fig. 1). The printer controller is con- nected to the remote multiplexer through a long line cable consisting of three twisted wire pairs. A separate pair is used for data transfer in each direction. The third pair provides bit-serial timing down from the remote multi- plexer to the station controller of the printout station. (1) 17 control characters. (2) 13 function command characters for the station controller. USASCII coding is used for all characters. The station can display up to 800 characters (40 characters per line by 20 lines) per display and can display 67 different charac- ters as follows: 26 alpha 10 numeric 28 pictorial 3 special pictorial a. Significant features. Some of 5 e significant features of the display station are as follows: (1) Capability to address any displayable location on the CRT screen. This enables the computer to transmit either a full or a partial line and/or screen (split screen capability) or to selectively update any symbol displayed anywhere on the CRT screen without need to send any “space” codes. (2) Capability to determine the location of the cursor. (3) Ability to send an interrupt code from the display station at any time. (4) Computer can enable or disable transmitting from the display station. (5) Ability to obtain a STATUS byte from the station informing the computer of the station status. (6) Ability to enable or disable refreshing of the infor- mation displayed on the CRT screen, intout station. Two nonimpact-type printout sta- tions are provided in the IATV display system. Each station is connected to the remote multiplexer and receives data for printout from the computer through the remote multiplexer. The printer mechanisms are connected to the The station controller provides storage for up to 1,024 characters of data. (To store up to 1,024 characters in the station controller, the program must first set the “cursor” to the first nondisplayable memory location in the station controller. If the cursor is set to memory location 0/0 in the station controller, then up to 960 characters can be stored in the station controller core-memory.) The printer controller contains the printer control and interface electronics for the printer mechanism. The printer controller also converts each USASCII coded character received from the core memory in the station controller into a 5 X 7 dot matrix and transmits the dot pattern generated to the printer mechanism. This printer can operate at a rate of 300 characters/s (225 lines/min at up to 80 characters/line, or at 450 lines/min with up to 40 characters/line). In the system, the computer can select any one of the two printout stations for data printout, and through the remote multiplexer, transfer data for printout by the selected station. Data for printout is first transferred at high speed (100 kbps) to the core memory of the station controller, stored, then transferred, one character at a time, to the printer controller for printout by the printer mechanism at a rate of 300 characters/s. onfroi lay 1. Control,. All operational control is accomplished through the CRT keyboard. The first 20 positions of line 20 are used exclusively for control input during operations. The last 20 positions of line 20 will be used by the system for response messages, so as not to conflict with other displays or the request portion itself. Each user of the keyboard must first idenbfy himself before submitting his message. Identscation pertains to the various elements of the operating system, such as telemetry, command, etc. Upon receipt of each message, the system will make the necessary identification and route to the applicable processor or program. isplay. All display will be accomplished on either a CRT or a line printer. a. CRT displays. CRT display capability consists of 8 CRT units (with attached keyboard). All units will have identical functionality, that is, any unit has total control and display capability. Each CRT screen has a display grid of 20 lines of 40 characters each. The system will initially supply 3 formats which are designed to cover all display needs in an efficient and standard manner. There are two types of format: GRID and ROLL. GRID formats are displays repiesenting cur- rent status; ROLL formats show parameter history as well as current status. A synopsis and example of each format used in the system follows. GRID formats (Figs. 2 and 3) Format A (24 displays, 4 X 6 MATRIX) (1) 8-character DESCRIPTOR (2) 8-character VARIABLE 15.48.19 T/M REQUEST S/C 2 FORMAT 123 AUX BAT CMD LOCK RATE IRR MlRR ON OFF 33 NOTSTOW MANACT CCO CCONE GRS2 INHIBIT INCREASE 88.50 324-01 PYGPEN DC64S DC63S AUPPEN OFF POST INS NT IN EF FALSE GES3 SVAC0l T W T l O VBSTCV 764+02 146.5 12345tO2 332241 ST A AGC ST A SPE SA GTPWR SA GFREQ 1234-02 77+00 1234-02 4321-05 ST A H A STA DEC STARUG STA DCC 138.5 35.0 1 2345678 8570 T, F123$ THIS HALF FOR REPLY (3) Fixed point-8 significant digits (4) Floating point-5 significant digits, %digit ex- ponent (5) ALPHANUMERIC-8 characters (6) ALL CONVERSIONS (BCD, BINARY, OCTAL, DECIMAL) Format B (18 displays, 3 X 6 MATRIX) (1) 12-character DESCRIPTOR (2) 12-character VARIABLE (3) Fixed point-12 significant digits (4) Floating point-9 significant digits, 2-digit ex- ponent (5) ALPHANUMERIC-12 characters (6) ALL CONVERSIONS 07.32.16 TRK REQUEST S/C 1 FORMAT 062 DSS 41 HA DSS 41 DEC DSS 41 RNG 4 1 27347 1647236 2 1 2476733 541 1-WY DOP S41 2-WY DOP 541 3-WY DOP 1234567890 1234567890 1234567890 541 RESOLVER S41 EX VCO C DSS 41 DCC 862 2898765432 1 8570 DSS 11 HA DSS 11 DEC DSS 11 RNG 1234567 1234567 123456789 S11 1-WY DOP S 1 1 2-WY DOP S11 3-WY DOP 1234567890 1234567890 1234567890 S l l RESOLVER S11 EXVCO C DSS 11 DCC 123 12345556789 1234 K, F062$ THIS HALF FOR REPLY Fig. form 1 ROLL formats (Fig. 4) Format C (16 line displays) (1) 3-column DESCRIPTORS (11 characters each) (2) First column always time (3) Variable has 11 characters (4) Fixed point-11 significant digits (5) Floating point-8 significant digits, 2-digit ex- (6) ALPHANUMERIC-11 characters (7) ALL CONVERSIONS ponent Display formats are determined at initialization time. This is in the form of format cards entered into the sys- tem via the card reader. The user may choose any of the three formats. b. Line printer displays. Line printer displays are han- dled in a manner identical to CRT displays. Formats are defined for several lines of print, each line having a separate format description card. Activation of a line printer display is also accomplished via the keyboard. 10.35.59 STA DATA REQUEST FORMAT 001 GMT AGC SPE 14 1-WY DOP 35.42 66.5 77.6 1234567890 35.43 66.5 77.6 1234567890 35.44 66.5 77.6 1234567890 35.45 66.5 77.6 1234567890 35.46 66.5 77.6 1234567890 35.47 66.5 77.6 1234567890 35.48 66.5 77.6 1234567890 35.50 66.5 77.6 1234567890 35.51 66.5 77.6 1234567890 35.52 66.5 77.6 1234567890 35.53 66.5 77.6 1234567890 35.54 66.5 77.6 1234567890 35.55 66.5 77.6 1234567890 35.56 66.5 77.6 1234567890 35.57 66.5 77.6 1234567890 35.58 66.5 77.6 1234567890 s, F1$ THIS HALF FOR REPLY or c. Operation. The execution of all displays is the respon- sibility of the several display processors, and not the individual programs, which, in general, do not initiate a display. All displays are initiated at the operator’s request. Upon receipt of a display request, the keyboard message processor (part of the system) will identify the user, unit, and format number. The format descriptors and map will be retrieved from the drum, the data will be retrieved from the program interface file, and the display will be executed. The IATV system integrated fairly easily with the EMR 6050 computer. The main problems encountered were in the checkout of the logic of the channel adapter. Since the channel adapter was a new design, these problems were not unexpected. After the system was made operable, checkout with user software uncovered a few problems in the printer sub- system of the IATV system. The most significant of the problems uncovered was the lack of an interrupt from the printer subsystem. Since the initial design of the printer subsystem was not capable of generating an interrupt to indicate when a given block of data had been printed, a wait loop was incorporated in the software sufficiently long to insure that the previous block of data had been printed before an attempt was made to print another data block. This proved to be an unsatisfactory solution be- cause of the limited memory space available in the com- puter for use as a printer data buffer. To solve this problem, the manufacturer changed the station controller used in the printer subsystem to provide an interrupt to the computer whenever a block of data has been com- pletely printed. This proved to be very satisfactory. Another minor problem with the printers was the lack of paper take-up reels. In this system, the printers are used as data logging devices, and therefore output a large quantity of paper as well as data. Since there were no take-up reels for the paper, the large quantity of paper output was somewhat of a problem. To solve this prob- lem, some surplus teletype reelers were salvaged from some unused printers and were mounted behind the printers. I The need for a viable user-computer interface is an ever-present requirement. In this case, an interface was needed in which the computer could output data in set formats and in which the user could enter changes to this data to the computer. This interface could have been implemented with discrete components as the Mark I1 system had been; but, because of the development of interactive CRT data terminals, a better alternative was available. missions. The choice of a CRT data terminal system for this inter- face proved to be satisfactory. In addition to being a satis- factory interface, it is also a flexible interface. It can pro- vide a good base for the development of other information display and control systems for future more compIex 1. Polansky, R. G., “DSN Mark IIIA Simulation Center Development,” in The Deep Space Network, Space Programs Summary 37-65, Vol. 11, pp. 9496. Jet Propulsion Laboratory, Pasadena, Calif ., Sep. 30, 1970. 7 R. levy DSlF Engineering Secfion An efficient procedure is described for reanalysis of space-truss structural frame- works. The procedure has been programmed to operate as a post-processor to determine response changes from sets of displacements developed for the initial structure by an independent structural analysis system. Examples given show sub- stantial savings in computation time when operating in conjunction with the NASTRAN structural analysis system. Efficient methods of reanalysis are useful to assess the changes in structural response as a function of modifica- tions to the properties of the individual member compo- nents of the structure. The applications encompass all design phases through inception and future alteration, particularly in conjunction with the implementation of the techniques for structural optimization. Currently, a number of diverse procedures for reanaly- sis have evolved to operate within the context of prevalent approaches to the overall formulation chosen for analyti- cal solution of the response problem. Some have been designed to be used in conjunction with force method, displacement method, or mixed formulations. Additional alternatives depend for their attractiveness upon whether or not the number of members for which modifications are considered form a relatively large or a relatively small set, or possibly whether or not the modified members can be readily localized to occur within restricted regions of the structure. Some of the procedures are mathematically exact in the sense that the solutions will be exactly equivalent to soh- tions that would be obtained from analysis of the modified structure ab initio. Other procedures develop approxi- mate solutions for which exact eventual convergence can be demonstrated iteratively. All methods usually appear to fall within either of two classes: (1) Given the inverse of the stiffness matrix for the initial structure, the inverse for the modified struc- ture is developed by perturbation. (2) The solution for the modified structure is developed as a linear combination of solutions for the initial structure. I In the following, an exact method of solution will be described that develops displacements and member forces for the modified structure from linear combinations of dis- placement function solutions developed for the initial structure. Conceptually, the method is based upon a paral- lel element approach. For each component member to be modified, a hypothetical parallel member is postulated. Cross-sectional properties of the parallel member are taken as exactly equal to the changes in properties of the parent (original) member and the connectivity is dupli- cated. The final member forces for the modified structure are the sums of the member forces on parent and parallel members. A condition of the solution is to maintain com- patibility of the parent and parallel member distortions. Entirely new members can be added by parallel members and original members can be completely removed by assigning duplicate properties of negative magnitude to the parallel member. Reanalysis is performed by a post-processor computer program that uses input displacement functions devel- oped previously by an independent structural analysis program. Because the only computed data that are input are the displacement functions, the analytical formula- tion procedure used to process the initial structure and develop these displacements is immaterial. The present implementation of the program is designed to accept input specifically in the format of the NASTRAN analysis system. A present limitation is the restriction to process only changes in the cross-sectional area property for one- dimensional bars. This limitation, however, is compatible with the design requirements of space truss structures, such as the frameworks of antenna reflectors. The advan- tages of this method are the simplicity of program opera- tion and input requirements, and the efficiency with which parameter studies can be developed for the response as a function of a spectrum of property changes for particu- lar bars. The displacements of the modified structure UM are obtained by superposition of displacements of the initial structure U , and the displacements AU caused by the internal forces of the parallel members acting as loads on the initial structure. That is, The order of the matrices in Eq. (1) is m X k, where rn is the number of unconstrained degrees of freedom, and k is the number of external loading vectors. To evaluate AU, it is convenient to express these dis- placements as the product of the displacements for unit values Us of the parallel member forces post-multiplied by the forces R of the parallel members, or [nul = [Us1 ERI (2) ( n f x k ) ( i i i x b ) ( b x k i In Eq. (2), the index b is equal to the number of property changes summed over all the members. To enforce compatibility, let eF = final distortions of parent members = distortions el = initial distortion of parent member for the exter- es = distortions of parent member for unit values of e, = distortions of parallel members for unit forces of parallel members nal loads forces of the parallel members Therefore, from superposition [GI = [eI] + [GI [RI (3) ( b x k ) ( b x k ) ( b x b ) ( b x k ) But e, can be determined directly as the distortions of the parallel members, (4) After combining Eqs. (3) and (4) and rearranging, R can be obtained by solving (5 ) The solution of Eq. (5) for R can be obtained readily in many instances because the order of the coefficient matrix is equal only to the total number of property changes. Examination of Eq. (5 ) shows that relatively little addi- tional computational effort is necessary to process more than one property change for a given member. This can be done by saving the es and e, matrices in Eq. (5) and repeating only the relatively minor effort in regenerating new e, matrices. In this manner, the responses for se- quences of property changes for given members can be obtained e5ciently. After solving, and combining Eqs. (1) and (2), the final displacements of the modified structure are obtained from [UHI = [UII + ius1 [RI (6) ( m x k ) ( m x k ) ( m x b ) ( b x k ) The final member forces can be found either by summing forces on parent and parallel bars, or else by expanding Ah the indices of Eq. (3) to cover all of the members of inter- est and then applying the internal force-distortion rela- tionship for these members. Parameter Matrix order rn Total number of bars Number of loading vectors k Number of bars in change group b NASTRAN CPU timesa Original structure: for Ur Original structure: for Ur and US In the limited case of one-dimensional bar members, the member distortion is the extension of a bar along its axis and is given by Structure II Structure I 48 1350 60 1492 2 2 6 37 32 s l l m i n 4 4 s 4 5 s 11 min41 s (cold start) Irestart) distortions for unit tensile loads, with each diagonal ele- ment of the form Reanalysis program CPU timesa Generate UDI for first area Generate UDI far each additional change area change where Ai is the change in area for the ith member and this can be positive or negative, depending upon whether the area of the ith member is to be increased or decreased. 2.7 5 37.0 s 0.3 s 2.3 s where Si, Li, Ai, and Ei are, respectively, the member force, length, area, and modulus of elasticity for the ith member. As an alternative to Eq. (7), the present im- plementation computes the extension from the grid co- ordinates and displacements associated with the bar as follows : 3 q f H ; -e; = i = 1,2, * ' . , b (8) Li ' where qf is the difference in displacements of the termi- nal nodes of the bar in the direction of the ath axis of Cartesian coordinates, H ; is the projection of the bar axis along the ath axis, and (9) The computations for Eqs. (8) and (9) are performed by using the member connection cards to define the terminal nodes and the grid cards that give the nodal coordinates. These data can be duplicates of the input for the original structural analysis program. The displacements used to generate the eI set are the responses to the applied load- ing. The displacements that generate the es set are ob- tained by applying a pair of unit loads directed towards each other at the terminal nodes associated with each parallel member; each pair of loads forms one loading vector and contributes one column of displacements to the U , matrix, which is used to compute one column of distortions in the es matrix. The unit loads that produce the es set are equilibrated by corresponding unit loads that extend the parallel mem- ber. Consequently, the e, matrix is a diagonal matrix of A flow chart of the reanalysis program described is shown in Fig. 1. The program is developed either to process property changes for individual bars or common property changes to groups of bars. Table 1 contains a summary of the central processing unit times for program operation (Univac 1108 Exec 8 computer) for two different structures that have been reanalyzed by this program. For both structures, the input displacements were read from a tape which was created by suppressing NASTRAN output punch card images by means of the executive breakpoint feature of the 1108 computer. The central processing unit (CPU) times shown include, in addition to computation time, the time used for data input (card and tape reading) and a substantial amount of printed output. Two NASTRAN runs were made in each case of Table 1. The first run was made independently of re- analysis requirements. The purpose was to check stability of the analytical model and possibly to provide some le XO tee SET A SET B SET OF BARS TO BE MODIFIED UNION OF SETS OF TERMINAL NODES FOR BARS OF SET A START (7 - MATCH AND ASSIGN COORDINATES TO UI BAR GROUP) TO BE MODIFIED SET C NODES FOR WHICH MODIFIED DISPLACEMENTS ARE REQUIRED UNION OF SET BAND SET C SET D \ INPUT: Us / insight into which members were the most desirable candidates for revision. When the checking can be elimi- nated and the members to be changed are known in advance, reanalysis could start from the second run, which generates all needed information. The time used for the first run, however, gives an indication of the time required to process one additional group change of member properties in lieu of using the reanalysis post- processor program. Table 2 illustrates the advantages in computation total time that can be obtained by using the reanalysis proce- dure to process up to four hypothetical property changes for a given bar group for the two structures. The tabula- tions show the time to analyze each modification as a new structure by using NASTRAN only and the time to process the modifications by the reanalysis program. The “NASTRAN only” column does not consider program restarts, because experience does not show significant economies in restarting when the basic structure is modi- fied. Although Table 2 shows substantial advantages for reanalysis post-processing when at least two changes are considered, the benefits would be even more pronounced whenever the first NASTRAN run is eliminated and only the second NASTRAN run (Table 1) is used. Computational time savings are most pronounced when the number of property changes is a relatively small pro- portion of all of the properties because the number of computations depends upon the number of changes. Whenever a large number of members are jointly to be given a small number of changes, the advantages of re- Number 04 ProPeW Structure changes 1 2 3 4 1 2 3 4 I I 1 analysis would tend to diminish. For example, if all mem- bers were to be changed, the time to generate the Us displacements would be equivalent to the generation of the full flexibility matrix. In such cases, it might be most efficient to analyze the modified structures independently. lime, s NASTRAN f post-processor reanalysis MASTRAN only 32 47.7 44 48.0 96 48.3 128 48.6 768 73 8 1536 740 2304 743 3072 746 In addition to the operations described in the foregoing discussion, it has been found convenient and e5cient within the reanalysis program to perform required addi- tional computations in terms of the modified displacements that are developed. For example, the present version of the program contains a subroutine to compute the rms deviations from the best-fitting paraboloid, using the UM displacements for a set of specified nodes (set 6, Fig. 1) on the reflector surface. B. Marcus and M. S. Katow DSlF Engineering Section Field measurements of reflector distortions, using the theodolite angle differ- ences and fixed arc lengths from the vertex of the paraboloid, are based on appar- ent displucements normal to the line of sight. Two computing methods are described which use directions informution from the structural computer anulysis to upgrade the readings in the pathlength errors sense. Comparative rms values of the 1/2 pathlength errors, after a paraboloid best fit, resulting from the field measurements, the anulytical analysis, and the r m equivalences to RF radio star measurements, are overluyed on an rms surface tolerance versus elevation angle chart for the 64-m-diam antenna. Close rms agreements allow designation of an error tolerance of t0.08 mm (0.003 in.) for the field-measured rms values. The state-of-the-art method of measuring distortions of a large ground-based antenna uses an angle-measuring theodolite mounted on the axis of symmetry just above the vertex of the paraboloid. The only other dimension measured is the arc length from the vertex to the target located on the surface of the paraboloid (Fig. 1). Since the theodolite can only measure apparent deflections mea- sured normal to its line of sight, the actual RF pathlength error may not be evaluated within useful precision when the 2 components of the readings are input to the rms computer program (Ref. 1). This anomaly was apparent since the first output of the three vector components from the analytical solutions. One solution proposed by R. Levy (Ref. 2) uses analyt- ically computed three-component distortions read by simulated field readings by the theodolite method. The difference in the two rms values can then be applied as corrections to the field-measurement values. This method, revised to use vector analysis, is described and is now incorporated as subroutine ANAFLD as part of the rms package (Type D). Another solution method, also described in this article, combines the single-dimension field measurement with direction vectors from the analytic solution to determine the correct rms value. Subroutine FIELD modifies the data before its input to the best fitting computation (Type 6). Table 1 defines terms used in the discussion that follows. eos Normally, the surface panels are &st installed and ad- justed close to the center of the adjustment range with the edges faired from one panel to the next one at zenith look. Then, the target mounting holes are drilled to pre- determined arc distances from the vertex of the parabo- loid using a strap gage. The change in arc distances is assumed presently to be negligible for the different ele- vation angle positions. At an elevation angle selected for the “perfect” parabo- loid position (45-deg for the 64-m-diam antenna), each Deflection vector Field reading Instrument angle reading $ Line-of-sight vector Measuring-plane x-axis XY-plane Y-axis Z-axis Zero-plane vector from the original position of the target point to its deflected position the projection of the deflection vector onto the line which passes through the target point and is perpendicular to the line of sight. The unit vec- tor in the direction TI1 has components, denoted Fx, Fp, Fz. Explicitly: if the target point Q has coordinates X, Y, 2, then F - F Y = F z = c o s $ the angle between the line of sight and its pro- jection onto the zero-plane vector from the point (0,0,38 cm) (0,0,15 in.) to the target point plane determined by the Z-axis and the line of sight; field measurements are made in this plane line in XY-plane perpendicular to Y-axis; this line’ is parallel to the elevation axis of an az-el an- tenna plane perpendicular to Z-axis, passing through the vertex line which passes through the paraboloid vertex and points to zenith when elevation angle is 0 deg (to horizon) axis of symmetry of undeflected paraboloid plane perpendicular to Z-axis, passing through the theodolite rotation axes and the datum tar- gets; &is is the reference plane for theodolite measurements -Xsin$ -Ysin$ (XZ + YZ)% ’ - (XZ + YZ)M ’ L. M .+ = +‘ - + = ROTATION CORRECTION R = VERTICAL MOTION Q’T = ACTUAL FIELD READING QT = CORRECTED FIELD READING 2 DEFLECTED MEASURING MEASURING PARABOLO1 D AT 45 deg r DEFLECTED DATUM TARGETS AT 45 deg L DEFLECTED DATUM TARGETS ARC LENGTH HEIGHT .I. e target is sighted with a theodolite located above the ver- tex and on the axis of symmetry, with the zero-plane aligned to the four datum reference targets located in the reflector structure. Two of the datum targets are lo- cated just above the elevation bearings along the X-axis and the other two are in the wheel girder of the 64-m- diam antenna reflector structure along the Y-axis. The angle $ between the target and the zero-plane, calculated from the focal length and arc distances of the paraboloid, is used to set the targets of the panels for the “perfect” paraboloid case (Fig. 1). After the refiector is rotated to another elevation angle, the reading process is repeated for the same target points. There will be a change in the theodolite’s angle $, de- noted as $‘ in Fig. l. Thus, the apparent deflection of target Q will be a vector ’T, which is normal to the line of sight. Due to the change in direction of the gravity load for the different elevation angle positions, the datum targets themselves deflect in a manner (Fig. 1) which effectively: (1) Rotates the measuring paraboloid as determined by the datum plane about the X-axis by angle 4. Since + is very small, we can make the approximation: 4 z sin 4. (2) Offsets the datum plane in th metry axis (Z-axis) by offset irection of the sym- It can thus be concluded that the field measurements, at elevation angles other than the perfect paraboloid angle, are made with respect to a rotated and translated coordinate system. Therefore, correction factors must be applied to obtain the true field readings follows : where the vector which represents the offset motion, and the vector which represents the rotation motion, since (XZ + Y2) 4F z = Figure 2 shows a typical analytically computed distor- with its three components U, in the X, Y, Z coordinate system. When this vector is field read, the theodolite’s field mea- suring plane X’OZ rotates about the symmetric axis OZ of the measuring paraboloid and picks up the change in angle $ (Fig. 1) from point Q to P. Since the angle $ measured by the theodolite to P and P’ is essentially the same, the component P’ in the measuring plane is used in the calculations. Z Z P’, the distortion vector, coordinate system must be transformed to the measuring plane X’OZ (Appendix A). The resulting components are U’, V‘, and W. Since the magnitude of the distortion vector is very small with respect to distance X’, the two lines of sight may be considered to be parallel for vector transformation calculations in the measuring plane. Of course, an optical micrometer used on the theodolite would conform exactly to the above assumption. To upgrade the field-measured deflection se from the analytically comput may be combined (Fig. 3). The direction sense ined by U’ and direction of the fi determine the magnitude of the th ’ is then normaliz or at point Q as that is, projected since the magni- tude of the normal error is a direct function of the path- length error. A further conversion is made for compati- bility to the rms analysis by dividing the normal by the tangent angle to result in a AZ deflection input (Appen- dix C). There has arisen a problem connected with the accu- racy tolerance of the field measurements when P’ is near zero. This method requires more careful rate field readings at these target points. To eliminate the effect on the accur final rms number, any target reading for which ) < 0.05 rad is presently de- leted from the least squares analysis. U- 11 The object of this computation is to simulate the theod- olite method of measuring the analytically computed three-component distortion vectors. Then, the resulting rms output from the best-fit program may be compared to the rms output from field measurements without cor- rections. As described in Section IV with Figs. 2 and 3, the field tends the “transposed to the measurement vec measuring plane” ve tude as well as the Therefore, vector best-fitting program to output the rms figure. computed and input to the In order to satisfy the data requirements of the rms pro- gram coding, the AZ component with W and V set equal to 0. deflection vector QT normal to the line of sight is com- puted by dividing by the cosine of the instrument angle $ (Fig. 4). Instead of assuming that the deflection vector is normal to the line of sight, one can assume that all deflections are Z a = TANGENT .3. I ZERO-PLANE X' 0 Fig. 4. Field measurement eonversions for Type A or B analysis in the line normal to the surface of the reflector (Type B). Using this assumption, QN equals QT divided by the cosine of (a - $) and it follows that QK', the A 2 com- ponent, is computed by dividing by the cosine of the tangent angle a. This results in the Type B rms answers. The different r m s analysis methods for field measure- ments were typed from A to D and applied to the avail- able 64-m-diam antenna data. Types E and F are from analytical computations. Since there was a large change (15,873 to 18,140 kg) (35,000 to 40,000 lb) in the weight of the cassegrain field cone assembly from the monocone to the tricone configuration, the systems were separately evaluated and the answers for reflector structure deffec- tions exclusively are noted in Table 2. At the present time, results from RF tests are available (Ref. 3) for only the monocone configuration; the answers projected from Table 2, with the addition of the surface panel rms distortions, etc., are overlayed on an upgraded curve from page 17 of Ref. 3 and reproduced here as 0.111 I I I I I I I I 0 STAIR ANALYSIS (TYPE E') 0 NASTRAN ANALYSIS (TYPE F') A FIELD MEASUREMENTS (TYPE 6') 0.09 0 IO 20 30 40 50 60 70 80 ELEVATION ANGLE, deg 2.5 2.0 E 1.5 1.0 Fig. 5. RMS surface toleranee vs elevation angle (64-m- diam antenna, monocone eassegrain system; zenith at- tenuation = 0.06 dB) Fig. 5. The upgrade of the curve is a change in the zenith attenuation from 0.036 to 0.060 dB. The conclusions that can be made at the present time (1) The field rms distortion value, using the present theodolite system and Type B or Type (F - D + A) for the 64-m-diam antenna, should be within eO.08 mm (0.003 in.) of the true value. (2) For a practical answer, the Type B analysis method could be used for the present theodolite system. This is the method used to report the field rms dis- tortion described in Ref. 4. (3) The Type C method requires very precise theodo- lite measurements, a condition marginally accom- plished at the present time. (4) There is a definite need to improve the accuracy of the field measurement method, if the method is used for larger antennas. (5) The NASTRAN computer analysis, used for the structural model plates and rigid jointed bars in the tie truss and the rectangular girder, definitely im- proves the fit between the analytic results and field measurements. are: I Trkone cassegrain system I A B C D F bStandard field measurement 0.91 (0.036) 1.45 (0.057) bAssumed normal direction field measuremmt 0.97 (0.038) 1.57 (0.062) bField reads f analytical directions (field subroutine) 1.09 (0.043) 1.63 (0.064) Theoretical f ield reads (ANAFLD subroutine) 0.94 (0.037) 1.40 (0.055) Latest NASTRAN analysis ‘/2 reflector model 4- some rigid joints 1.01 (0.040) 1.52 (0.060) A’ B’ E’ F’ 0.81 (0.032) 0.86 (0.034) 1.01 (0.040) 0.94 (0.037) bStandard field measurement bAssumed normal direction field measurement STAIR analysis ’/4 reflector model f pin joints latest NASTRAN analysis ‘/2 reflector model -k some rigid joints 1.45 (0.057) 1.60 (0.063) 1.90 (0.075) 1.40 (0.055) aSurface panels were set at 45-deg elevatio ... 1. Katow, M. S., and Schmele, L. W., “Antenna Structures: Evaluation Techniques of Reflector Distortions,” in Supporting Research and Advanced Development, Space Programs Summary 37-40, Vol. IV, pp. 176-184. et Propulsion Labora- tory, Pasadena, CaPif., Aug. 31, 1966. Consideration in the Comparison of Measured and rtions,” in The Deep Space Ne pp. 102-107. Jet Propulsion Calif., Jan. 31, 1970. 3. Bathker, D. A, Radio Frequency Performance of a 210-Ft. Ground Antenna: X-Band, Technical Report 32-1417. Jet Propulsion Laboratory, Pasadena, Calif., Dec. 15,1959. 4. Katow, M. S . , “Techniques Used to Evaluate the Performance of the NASAWPL 210-Foot Reflector Structure Under Environmental Loads,” in Structures Tech- nology for Large Radio and Radar Telescope System. Edited by J. W. Mar and H. Liebowitz. The M.I.T. Press, Cambridge, Mass., 1969. 118 With the measuring plane restricted to rotation 0 only about axis OZ (Fig. 2), only X and Y components of deflec- Substituting Eq. (A-1) into Eq. (A-2) results in tion vectors are transformed. ut' = u COS e + v sin 3 vft = vcose - usin~r (A-3) Based on the basic coordinate system, BT = rcos+ = rsint$ where Transformed to the measuring plane, Ut' If (A-1) Also, for target point Q, if Q has components (XQ7 Y,, Z,), Xb = (Xi + Y ; p Yh = 0 z', = z* Y 0 = tan-> - X Also, in this coordinate system, the vector QT/llQTll has components ( F i , F)p, F i ) , where Fk= sin$ F;Y = 0 FG = cos $ e magnitude of the theoretical deflection vector ’ To then compute for the magnitude of the normal error in Fig. 3 is computed as follows. N, when angle Q N P = 90 deg: If the direction vectors of NII = projection of ’ onto the unit normal vector When Nx and N z are unit vector components of the normal, unit vector components of as components (Fk, Fk), the nit vector of QT equal QT when ause we assume that the two IlQNll = To establish compatibility with the rms analysis, the nor- N I I is converted to a A 2 by Transposing, 1 g. 3, the circle product of the two vectors which is equal to where T’ is a unit vector in the direction of U” = -1 X sin$ ’ = l X c o s $ 1 R. A. Paine SFOF/GCF Development Section Frequent reconfiguration of the Space Flight Operations Facility to meet changing mission requirements necessitates complex cable changes throughout the building. This article summarizes a detailed well-defined cabling plan which maintains discipline in the instalktion and removal of cabling. The SFOF experiences frequent reconfigurations both in its physical layout and functional capabilities. These changes are necesqary in order to meet new requirements placed on it by the DSN and unmanned spaceflight programs. Without a detailed well-defined cabling plan, there would be problems with the installation of new cables, rerouting of existing cables, and d.isposition of unused cables. Such conditions would result in overly congested cable drops, cable ducts, and unruly cable lays. This paper discusses the procedures developed to maintain cognizance and control of the cabling in the SFOF. There are cable drops and cable ducts that intercon- nect the floors of the building. The cable drops are cutouts in the floor that give access to the cable ducts, which are vertical enclosures between floors. The cable drops and cable ducts were located, identified and documented. Figure 1 is a typical example of the docu- mentation of the cable drops and cable ducts for the four floors. With the documentation of the cable drops and cable ducts, the vertical cable routes may be determined. The horizontal cable runs, which are under computer flooring, may be determined from the same set of draw- REPRESENTS INTERCONNECTING CABLE FOR LINE PRINTERS 1443 AND 2501 CABLES W20100 THROUGH W20106. CABLES W20130 THROUGH W20140. CABLES W20165 THROUGH W20169. ---- REPRESENTS INTERCONNECTING CABLE FOR 2260 CRT SWITCHER. . . . . . . . . . . . . REPRESENTS INTERCONNECTING CABLE FOR FORMAT REQUEST UNIT. ings (Fig. 1). The drawings are to scale, and the archi- tectural configuration and the equipment layouts of the various ffoors are maintained on this set of drawings. Once the equipment layouts are complete for any given area, the cable routing may be determined. The cable routing will then be documented. Figure 2 is typical of this type of documentation. This, then, shall be the way in which the cables are installed. This method eliminates the congestion in the cable drops, cable ducts, and hori- zontal cable lays. When a system is removed or changed, the cable documentation shall be revised to indicate the modifica- tions or deletions. This makes it possible to control the cable density in the cable drops, cable ducts, and cable lays. k 123 P1 P2 es Figure 3 is representative of a standard cable drawing. As new cable requirements arise, an effort is made to use a cable drawing already in existence. This reduces the number of different types of connectors and wires in a system and also results in a cost savings. In the electromechanical design phase of a system, every effort is made to incorporate the standard cables by selecting interface connectors that will mate with a standard cable and still fulfill the interconnecting requirements. able l ~ v e n t o ~ W,ith the large quantity of cables that are used in the facility and the fact that cables normally do not wear out, it is advantageous to reuse the cables. This is accomplished by maintaining a cable inventory. The cables are identified with a JPL part number. This part number consists of two parts: the basic draw.ing number and the number indicating the length of the cable in feet. The cable will have its part number attached to it by use of a self-adhering label. Having identified the cables and documented their routing, maintaining an “in use” status on the cables can be accomplished. As a data processing system is revised or removed, the cables taken out of service may then be stored for future use. CABLE PART NO. CONSISTS OF DRAWING NO. AND LENGTH IN FEETe.g., XXXXXXX-XXX ELECTRICAL SCHEMATIC Fig. 3. Standard cable Because of the complexity of the SFOF and the con- stant upgrading of its capabilities, it was important that a system for maintaining discipline in the installation and removal of the cabling be developed. With the ability to predetermine cable routes, we can define the length of a cable, limit the density of cable drops and cable ducts, and using the standard cable inventory, determine if a cable is available, or if one will have to be made. R. A. Wells SFOF/GCF Development Section Prior to critical periods of mission processing, comprehensive diagnostic tests of the SFOF central processing system are conducted to detect and correct equipment deficiencies before they can affect the continuity of spaceflight data processing. The article describes the test methods employed in their relutionships to the current dual IBM 360/75 computer configuration. By preparing a test “script” in advance, the hardware checkout process is formalized and the results documented. . I Prior to periods of critical mission data processing, it is worthwhile to conduct a comprehensive series of diagnostic tests to isolate and remedy equipment mal- functions. These diagnostics provide standard criteria and uniform methods for verifying the technical condi- tions of all critical computer-related hardware in the SFOF central procesing system (CPS). In the past, such tests have been performed immediately prior to most launch, midcourse and encounter operations for Mariner, Surveyor, Pioneer and Lunar Orbiter missions. They are also planned in support of the Mariner Mars 1971 mission. It has been the custom to prepare a detailed script for each checkout to reflect the mission-committed hardware configuration, the test durations and the procedural sequences. When completed, this itemized record can be used as the basis for system buyoff and a formal pre- mission transfer to the responsible operational organi- zation. The first two articles in this series (Refs. 1 and 2) have described the various types of diagnostic routines which presently exist for the IBM 360/75-related portion of the CPS. A full set of compatible diagnostics are currently operable under the diagnostic monitor (DIAMON) which supervises their execution under the overall control of the JPL operating system (JPLOS). I CPS diagnostic checkouts serve the following purposes: Verifying performance of newly installed equip- ment. Certifying the condition of previously installed equipment. Exercising hardware elements ind,ividually to ex- pose suspected faults. . IB a Training personnel in a ‘hands-on” environment. Resolving many uncertainties concerning alleged hardware-versus-software problems. (8) Numerous 360/75-driven UTD user devices Uncovering marginal performance. Correcting system deficiencies. Formalizing hardware acceptance. (7) The SFOF time reference interface for GMT dis- plays and CPU timers. throughout the SFOF. There are also 360/75-to-360/75 channels, not pre- sently utilized, and additional peripherals for adminis- trative compuhg section support. These are not part of the committed mission configuration and thus are not discussed here. Figure 1 identifies the 360/75-related elements of the Mark IIIA CPS as they exist in SFOF at the present time (Ref. 3). Not shown are numerous user station devices which are installed in the various mission support areas, the DSN monitor area, and the DSN operations control area on the first floor of the SFOF. In substance, the two 360/75 central processing units (CPUs) may be con- sidered identical and interchangeable, with input/output lines and user terminal and display (UTD) devices switchable between the two. Switching control is vested with the computer chief (or system controller) in the data processing control center (DPCC). Transfer is accom- plished through two DPCC consoles: (1) a user device switching console, and (2) a GCF high-speed data/ wideband data (HSD/WBD) switching console. DPCC personnel are also provided with monitoring and entry devices such as display stations, line printers, and a card reader. b@S Prior to each checkout period, a plan or script is prepared to assure that testing will proceed in an orderly manner, will cover all committed equipment, and will be properly documented. The script: (1) Formalizes the sequence of testing. (2) Gives the required calls (2260 display station requests) for manual entry by the operator. (3) Specifies a suggested duration for each of the tests (number of data blocks, characters, transfers, etc.). (4) Fixes optimal data patterns. (5) Indicates the expected message responses and error printouts. Diagnostic testing involves all of the equipment ele- (6) Identifies variations, or options, for each test. (7) Provides a checkoff column for logging test results. ments shown in F,ig. 1 plus: (1) The communications processor (CP) teletype (TTY) input/output (I/O) interface with GCF. (2) The HSD 1/0 interface with GCF. (3) The WBD input interface with GCF. (8) Acts as a training aid for participating personnel. (9) Is a debriefing med,ium for post-test critiques. (10) Provides a basis for system buyoff and transfer. (4) The Univac 1108A and 1108B I/Q interface with the scientific computing facility (SCF). At the outset of testing, the SFOF time reference and internal CPU timer registers are usually checked first as they are fundamental to all subsequent system testing. Since the test director usually operates from the data processing control center (Fig. l) , DPCC devices are gen- erally tested next. Following this, other diagnostics may be initiated in parallel for concurrent execution. (The (5) The Control Data 3100 output interface for UTB DTV displays. (6) The mission display board (MDB) interface to the UTD Vigicon projectors. II II II II II present diagnostic implementation requires total dedica- tion of the system elements being tested). All DIAMON-related diagnostics are resident under JPLOS and are available for use by SFOF operations personnel as required. In brief, these scheduled pre-mission checkouts are conducted periodically from published scripts. Diagnostic requests are entered via 22601 display station keyboards, DIAMON messages are displayed on 2260 CRTs, and a hard-copy log is provided on the designated line printer(s). Full-scale tests, such as described in this article, are scheduled upon request to assure optimum system per- formance in support of JPL’s mission support commit- ments. 1Also known as manual entry devices ( MED ) . 1. Wells, R. A., “Diagnostics for the SFOF Mark IIIA Central Processing System: Standalone Acceptance and Maintenance Routines,” in The Deep Space Network, Space Programs Summary 37-65, Vol. 11, pp. 97-99. Jet Propulsion Laboratory, Pasadena, Calif., Sept. 30, 1970. 2. Wells, R. A., “Diagnostics for the SFOF Mark IIIA Central Processing System: 360/75 On-Line Test Routines,” in The Deep Space Network, Technical Re- port 32-1526, Vol. I, pp. 103-106. Jet Propulsion Laboratory, Pasadena, Calif., Feb. 15, 1971. 3. Stiver, R. A., “Mark IIIA IBM 360/75 Computer Configuration,” in The Deep Space Network, Space Programs Summary 37-66, Vol. 11, pp. 71-75. Jet Propulsion Laboratory, Pasadena, Calif., Nov. 30, 1970. R. G. Hanselman SFOFIGCF Development Section This article covers the reconfiguration of the Goldstone Deep Space Communi- cations Complex area microwave terminals at the Mars Deep Space Station and the Goldstone area communications terminal for the transmission of dual SO-kbit digital data streams between the two locations. This article describes the reconfiguration of the Goldstone DSCC microwave terminals required for the transmission of dual 5O-kbit/s digital data streams be- tween DSS 14 and the Goldstone area comm terminal (ACT) located adjacent to DSS 12. This article is an amplification of two earlier articles (Refs. 1 and 2). equirements for a 5O-kbit/s data transmission capa- bility of the GCF were derived from the GCF functional design for 1971-1972 (Ref. 3). The GCF functional design specses two 5O-kbit/s data streams between the SFQF and DSS 14, one stream being a backup to the other. The current microwave transmission capability between DSS 14 and the Goldstone area comm terminal is entirely JPL owned and operated. That portion of the overall DSS 14-to-SFQF transmission capability existing between the Goldstone area comm terminal and the SFOF is owned and maintained by Western Union and leased to JPL. The problem of how to best use the available micro- wave baseband spectrum between DSS 14 and the area comm terminal for dual 50-kbit streams was thus a portion of the development effort. There are five duplex microwave channels between DSS 14 and the area comm terminal. Each of these chan- nels has a 10-MHz baseband bandwidth. The equipment manufactured by Collins Radio Corp. of Dallas, Texas, is an off-the-shelf system and is designed primarily for inter- city telephony or broadcast quality television. This par- ticular group of equipment has been installed at the Goldstone DSCC since 1966 and is used for numerous functions such as: (1) Multiple-mission support, wherein the tenna is used to track a spacecraft and the resultant DSS 14 data is forwarded to data reduction equip- ment at another station via the microwave channell. (2) GCF voice circuits between the two stations. (3) GCF teletype circuits and high-speed data circuits between the two stations. Included as a portion of each terminal is a standard fre- quency division multiplex (also manufactured by Collins Radio). Prior to the requirement of 50-kbit data between DSS 14 and the ACT, this multiplex equipment was used to pass several operational voice channels. The GCF recognized the requirement to transmit the following information between the two stations, utilizing as few microwave channels as possible and still be guaranteed of high reliability: 9 Channels of voice 2 Channels of 4800-bit/s high-speed data 6 Channels of 75-bit/s teletype 2 Channels of 50-kbit/s wideband data 1 Channel of 1 pulse/s intersite timing The frequency division multiplex (Collins MX-106) was originally purchased as a basic 60-channel (voice) system, wired for 24 channels and equipped with 12 channels. In the world of commercial communications, 12 voice channels can be multiplexed into a 48- occupying the basic spectrum of 60 to 108 kHz. Channel modems perform the basic modulation of voice signals in the 0 to 4 kHz portion of the spectrum to 60 to 108 k portion of the spectrum. Four kHz is allocated for each of these modems, the twelve channels thus forming a “group” 48 kHz wide. As earlier stated, only one group initially existed (i.e., 12 voice channels) but the wiring, module space, power, etc., provided for future expansion of the multiplex. The General Electric TDM420 modems used to trans- late the GCF 50-kbit conditioned signal to that portion of the baseband spectrum normally occupied by a group of 12 voice channels, generate and accept a vestigial side- band signal occupying 60 to 108 kHz in the baseband spectrum. I Initially it was planned to place the vestigial sideband signal directly on the microwave baseband, thus dedi- cating two 10-MHz-broad channels for carrying the two 48-kEHz 50-kbit signals. Further analysis led to the conclu- sion that the available microwave channels would best be utilized by concentrating as many GCF signals onto one broadbanded channel as practicable. Investigation of the available hardware at each terminal showed the following: An existing baseband combiner which could auto- matically and instantly switch between two micro- wave receivers. A radar multiplex which could be used to pass fast risetime pulses in the 4 to 6.5 M available baseband spectrum (such as 1-pulse/s timing). Resistive splitters allowing a single source to drive two microwave transmitters simultaneously. Available carrier source already installed in the MX-106 multiplex. It was thus determined that the most economical and reliable usage of the microwave would be to stack all known circuits onto two microwave channels and arrange the receivers at each end to feed the GCF equipment via the baseband and combiner mentioned earlier. The net result of the diversity arrangement is one 10-MHz full duplex (both directions simultaneously) diversity video channel carrying the following signals : Secondary 50 kbit 12 (each) voice circuits1 Prime 50 kbit Continuity pilot carrier Time pulse and time pulse verification 456 to 504 kHz o obtain the additional 48-kHz-wide group for the second 50-kbit channel it was only necessary to purchase two each group modulators and demodulators and plug them into prewired mounting positions at each station. Figure 1 shows the multiplex and diversity arrange- ment at both DSS 14 and the ACT. Both stations have 1Each voice circuit can pass either voice, 4800-bps high-speed data, or as many as 18 teletypewriter circuits, c 2 I * N 1 Y m 0 3 4 0 Y 2 + .- A Y 0 v) P i % L Q) I b an identical arrangement, the only difference being the operating frequency of the microwave transmitters and receivers. system, certain rearrangements were made to the exist- ing area microwave between DSS 14 and the ACT to take advantage of the wideband characteristics of the microwave channels. Signals from all GCF systems-voice, high speed, teletype, and wideband-may now be placed on a single diversity channel. Additionally, the GCF has ample bandwidth on this one channel for future expansion. In response to the requirements put forth in the 1971- 1972 GCF functional design for the 1971-1972 wideband 1. McClure, J. P., “GCF Wideband Digital Data System,” in The Deep Space Network, Space Programs Summary 37-65, Vol. 11, pp. 107-110. Jet Propulsion Laboratory, Pasadena, Calif., Sept. 30, 1970. 2. Manselman, R., “Wideband Digital Data System Terminal Configuration,” in The Deep Space Network, Space Programs Summary 37-66, Vol. I 110. Jet Propulsion Laboratory, Pasadena, Calif., Nov. 30, 1970. . P., “Ground Communications Facility Functional Design for 1971- 1972,” in The Deep Space Network, Space Programs Summary 37-66, Vol. II, pp. 99-102. Jet Propulsion Laboratory, Pasadena, Calif., Nov. 30, 1970. R. B. Crow RF Systems Development Section A new frequency generator/distribution subsystem is being developed to meet the increasing complexity of the Deep Space Network Mark I l l data system. The coherent reference generator is an assembly that will accept the primary fre- quency standard from the hydrogen maser (or possible secondary standard from the rubidium, cesium, or remote standards) and furnish required reference fre- quencies for a deep space station. A new frequency generator/distribution subsystem is being developed to meet the increasing complexity of the DSN Mark I11 data system. The coherent reference generator is an assembly that will accept the primary frequency standard from the hy- drogen maser (or possible secondary standard from the rubidium, cesium, or remote standards) and furnish the required reference frequencies for: Receiver assembly (9 each). Programmed exciter assembly (3 each). Subcarrier demodulator assembly (8 each). Ranging demodulator assembly (2 each). Mark 11 frequency/timing system (1 each). The basic design goals that have been stressed are (1) performance, (2) economy, (3) computer control/ monitoring, and (4) minimum physical size. A simplified block diagram of the Mark 111 frequency distribution and the coherent reference generator is shown in Fig. 1. All the inputs to the coherent reference genera- tor will be simultaneously switched when a change in the input frequency standard source is made. A simplified block diagram of the coherent reference generator is shown in Fig. 2. Note that the 1- ence output will have a fail/safe power supply so that there will be no interruption to the station timing system should primary power fail. A list of the preliminary detailed design specifications for the coherent reference generator is shown in Table 1. Be I . a Parameter Phase stability Power output bVSWRin bVSWRout Harmonic distortion Nan-harmonically related spurious OUtDUt Isolation output to output Source at requirement equirement Proposed specification DSN Mark 111 data system development plan [Document 803-1,Val. III,p. ll.C(l)] Charged particle calibration 0.2 m; variation in electrical phase path = 0.2 m Phase budget for Mark 111 receiver/exciter Receiver 4 f0.06 m which implies Receiver reference (0.1, 1, 10, 10.1,45 MHz) = k33 deg (based an differenced range versus integrated doppler technique) at S-bond -1330 deg at S-band Exciter A 30.08 m which implies Exciter reference (50 MHr) = S_ 440 deg at S-band f44 deg at S-band Accepted DSlF reference level Range = 4- 10 to -k 13 dBm; stability k0.5 db Engineering judgment 1.1:l Engineering judgment 1 .5:1 Engineering judgment 5% Engineering judgment Minimum of 70 dB below nominal output Frequency, MHz Isolation, dB 0.1 k 120 1 sz 120 5 s, 120 10 k 110 45 k90 100 &Environment: time = 12 h; temperature = control roam temperature 2217°C (25°F); power supply variations of 3%; elastic survival for temperature bVSWR = voltage standing-wave ratio. of O + 50°C; dc power variation of f5%. COHERENT REFERENCE GENERATOR INPUT SUBASSEMBLY r -- CESIUM I FREQUEN STANDAR L - - 100 kHz 1 MHz 5 MHz 10 MHz 100 MHz I _____1 _____() - _____I COHERENT REFERENCE GENERATOR AND DlSTRl BUT1 O N SUBASSEMBLY COHERENT REFERENCE GENERATOR I1 I RECEIVER ASSEMBLY S-BAND (4) X-BAND (2) SPARE (1 ) OPEN-LOOP (2) ----- , GENERATOR , I OPERATIONAL (8) 10 MHz RANGING DEMODULATOR ASSEMBLY OPERATIONAL (2) I I I DI STRlBUTlON AMPLIFIER 100 kHz AMPLIFIER AND POWER DIVIDER DI STRl BUTION ( 1 5 ) AMPLIFIER 1 MHz 5 MHz 10MHz (HYDROGEN MASER ONLY) 100 MHr (HYDROGEN MASER ONLY) AMPLIFIER AND POWER DIVIDER 1.1-MHz SYNTHESIZER x2 SYNTHESIZER X10 SYNTHESIZER AMPLIFIER AND 45-MHz SYNTHESIZER 100MHz I 50MHz I r I AMPLIFIER AND POWER DIVIDER AND + 2 t These specifications were developed to assure that the fre- quency distribution and synthesis of the coherent refer- ence generator offer reference frequencies that allow the receiver/exciter subsystem to meet design requirements as dictated in the Mark I11 Data System Development Plan (Document 803-1, Vol. 111). A typical distribution amplifier that d be a basic building block in the coherent reference generator is shown in Fig. 3. Broadband circuits with internal feed- back will be used to meet the phase and amplitude stability requirement. No intentional filtering or auto- matic gain control will be attempted in the distribution amplifier since these characteristics have already been defined either in the frequency standard or the refer- ence generator. Preliminary design of the coherent reference generator was started in the middle of January 1971. Present plans call for an engineering mode1 to be completed in fiscal 1972. OUTPlJT 2 I OUTPUT N ISOLATION AMPLIFIER JP% T ICA h 9 35 1. Butcher DSIF Operafions Secfion A hardware-software system is described that is capable of selecting the one from among as many as six incoming data streams which is best by an externully programmed criterion and switching it automatically to the Deep Space Instrm- mentation Facility (DSIF) Telemetry System. The system has been implemented at the Cape Kennedy Compatibility Test Station to provide the best spacecraft telemetry stream to the DSIF Telemetry System during the near-earth phase of a tracking mission, when as many as six Air Force Eastern Test Range stations are receiving spacecraft telemetry. The requirement of the Tracking and Data System for near-earth telemetry support is an important function in the unmanned spacecraft launch support. The require- ment is placed upon the Air Force Eastern Test Range and Kennedy Space Center stations to provide spacecraft telemetry in a serial bit stream to DSS 71 after launch until DSN acquisition. The spacecraft telemetry serial bit stream is received at DSS 71 via 202 data modems. It is processed at DSS 71 by the DSIF Telemetry System and transmitted to the SFOF in real-time. There are as many as six near-earth stations receiving and, subsequently, transmitting telemetry data to DSS 71 during the launch phase. The automatic switching unit at DSS 71 provides a means of selecting the best available data and performs automatic switching to select the station data source to output the data to the DSIF Telemetry System for process- ing. This is to be accomplished with a minimum loss of spacecraft data during the time of launch until loss of signal by the last near-earth supporting station. This data is to be processed at DSS 71 and transmitted to the SP in real-time via the GGF high-speed data system. I The automatic switching unit consists of an XDS CF-16 system controller and an interface buffer implemented with JPL Hi-Re1 digital modules as utilized in the DS Telemetry System. The @F-16 system controller is a pack- aged unit consisting of digital logic and stored programs with an 8k core memory. Peripheral equipment includes an ASR-33 teletype with paper tape reader and punch for input/output, and an F -1200 magnetic tape recorder for recording all incoming data lines. A block diagram of the automatic switching unit is shown in Fig. 1. The interface buffer logic consists of 15 digital Hi-Re1 circuit modules mounted in a single chassis for input, out- put, and data switching to the CF-16 system controller. The telemetry data input from the 202 data modems is 26-V amplitude. All six inputs are input to negative-to- positive modules for uni-polar output clamped at 5 V. All six lines are then fed to the CF-16 controller to be used for frame sync and line selection when in the auto- matic mode. Each input data line also is routed through another negative-to-positive circuit to the line selection logic, which has the capacity of selecting one data line whose data is sent to the DSIF Telemetry System for processing. The data selection can be achieved by either manual or automatic modes. The manual or automatic mode is selectable by a toggle switch on the front panel. When the manual mode is selected, it enables the use of six front panel pushbutton switches to select the desired data source. When the automatic mode is selected, the CF-16 is utilized under program control to select the data source. 202 DATA MODEMS 1 . The output data is also present on an interrupt line and is used for bit sync detection. A timing signal of 1 kHz is supplied to the CF-16 on an interrupt line. A lift-off pulse or first motion pulse is sup- plied to the CF-16 to provide time from lift-off for the automatic operation. As presently implemented, the system has a capabiIity of monitoring six input data lines and selecting one line for output data. Criteria for the selection of a data line are data quality, priority of the data, and tracking predict times for each data source. Data quality is determined by checking for a frame sync pattern in the received data. Counts are kept of the number of good and bad frame syncs for each data line, and therefore a current frame sync status of each line is maintained. Data line priorities are used to help determine which source to select if two or more lines have a good frame sync. Tracking predict times are used to select a data source if no lines have a recognizable frame sync pattern. Predict times are in seconds from first motion to acquisi- tion of signal (AOS) and loss of signal (LQS) for each station and are counted with a 1-pps clock in the software. ____QI SELECTED DATA OUTPUT TO DSIF TELEMETRY SYSTEM NEAR-EARTH TRACKING STATIONS b, - b p =TIME IN rns FROM MID BIT TO MID BIT bg - b4 = TIME IN ms FROM LEADING EDGE In the polling loop, the sync status for each line is checked. If only one frame sync is found, the processor will select the data line corresponding to the frame sync. If that line happens to be the one already selected, the re-selection does not affect anything. If more than one frame sync is found, the processor checks the priority of bl bz b3 b4 TO MID BIT The software for the system is written entirely in sym- bolic language and is divided into three sections: a main routhe, and two interruptable subroutines. The main rou- tine handles all inputs/outputs and, in addition, performs the line switching for the system. The 1-kHz interrugt subroutine controls all the timing functions for the system and maintains bit sync and frame sync status for each of the six input lines. By determining middle bit time, the data interrupt subroutine is used to help maintain bit synchronization for the 1-kHz subroutine (Fig. 2). routine The sequence of operations begins by first setting up the interface logic to accept data on line 1. This would normally be pre-launch and launch data. Then the proces- sor enters an input routine whereby pertinent constants are accepted into storage through the ASR-33 teletype input. These constants are: (1) Bit rate of expected data. (2) Lead time in seconds before AOS to begin looking (3) Number of sequential good sync codes to recognize before setting the frame sync flag. (4) Number of sequential bad syncs to have in order to reset the frame sync flag. (5) AQS times for each data line. (6) LOS times for each data line. (7) Priority number for the line. The processor can handle bit rates of either 8% or 33% bps, and is initialized for 33% bps. If 836 bps is entered for the bit rate, the millisecond counts for middle bit time and total bit time are changed accordingly. for data. After the input routine is completed, the next step in operations is to initialize the message typeout routine and the count for the 1-pps timing. Then the two interrupt routines are enabled and the processor will go into a poll- ing loop to check sync status, AO§, and LO§ for each data line. the lines. A line with frame sync and a priority of one takes precedence, but if no line had a priority of one, the first line found with frame sync status would be the selected line. If no lines had a frame sync, the AOS and LOS times would be examined for current time (X) to be in the range AOS(N) < X < LQS(N) where N is the line number. The first data source to meet this condition would be the selected line. After scanning the status of all the data lines, the next step in the polling loop is a check for launch vehicle first motion. This event, when it occurs, signifies the start of the 1-pps timing clock. The main routine outputs a mes- sage to show that first motion has occurred. Until the time of first motion, only a line with an AOS time of zero (normally the preselected line) could be selected and the processor would ignore all other lines. After sensing for first motion, the main routine next enters a loop where it looks for a change in frame sync status on any line. If a data line has had a change of sync status, a message giving the new sync condition, the line number, and the current 1-pps time value will be typed out. The 1-pps time value is unfortunately typed out as a hexadecimal number instead of a more readable decimal number. This is the result of a time constraint placed upon the main routine in that if it took the time to con- vert to decimal before typing out, the portion which has to make the line selection could not be executed often enough. When this loop has been completed for all lines, the main routine will return to the line selection where the cycle begins again. 1s the execution of this rou- tine so that the 1-pps count, data sampling, data bit sync, and total bit count per frame are all controlled by this timing reference. The subroutine first checks for a first motion flag to have been set. If the first motion has been sensed, a count of 1000 is decremented to generate the 1-pps clock. If first motion had not occurred, the 1-pps clock would be bypassed. If a 1-pps count is generated, the AOS and OS times are decremented for each line so that these counts may be used by the main routine for line selec- tion. No frame syncs wilI be set for any line not having AOS 4 T < LOS. For each 1-kHz timing pulse, the subroutine decre- ments a count which determines the appropriate time to sample a data bit. It represents the time in milliseconds from middle bit time to middle bit time or from leading edge to middle bit time reference (Fig. 2). To aid in bit synchronization, this count is controlled by the data inter- rupt subroutine. If the count for mid bit has reached zero, the routine will then sample the data lines simultaneously by input- ing in parallel and then performing a parallel-to-serial conversion so that the input block is separated into sep- arate lines. Then each stored word of 16 bits correspond- ing to each data line is shifted down and merged with the new data bit for each line. After doing this segment, the subroutine will exit, and on the next 1-kHz count, it performs the portion of the subroutine which checks for frame sync. The reason for the two segments is that the total time required to execute the subroutine is greater than the 1-ms interval between interrupt pulses. With the next 1-kHz pulse after the sampling pulse, the routine enters the segment to check for frame sync. First, it decrements a count representing the total number of bits in a frame for each line and then looks to see if there is frame sync for each line. If there is, the next check is to see if it is time to look for frame sync again. If it is, the routine then compares the stored input data word against the known frame sync pattern both normal and inverted. If it is a good code, the count for bad syncs is reset. If it is not a good code, the number of bad frame syncs for the line is incremented and the number of good counts is reset. The number of good or bad counts con- sidered to be in or out of lock is a keyboard entry in the main routine. If it were not time to look for the frame sync code, the routine would exit and await the next cycle. If the frame sync flag were not set for the line at the time to look for frame sync, the routine would then check AOS and LOS times. This would be the initial condition for any line before it had good data. If the current time (T) corresponded to AOS < T < LOS for that h e , then the routine would check the sync codes. If a sync code is matched with the data word, the number of good frame syncs is incremented. The proper number of consecu- tive good frame syncs would set the frame sync flag for that line. Each data line has its own frame sync loop and good and bad count status so that input data lines with a time delay of a data bit time or more between corresponding data bits may be handled simultaneously. This routine is executed by an interrupt pulse caused by the positive-going edge of a data bit of the data source selected. The subroutine functions as a bit synchronizer by resetting a count which represents the time in milli- seconds from leading edge of a data bit to middle bit time (Fig. 2). The priority of this interrupt is higher than that of the 1-kHz interrupt, thus always giving a correct mid- bit count for data sampling. In the event of a power failure, two high-priority inter- rupt locations are available to access the power fail/safe routine; one for power down, the other for power up. When entered from a power down interrupt, the routine saves the contents of the program accessible registers and the program counter register, stores the current selected line number, and then halts the processor, When power comes back on, the power up interrupt again accesses the subroutine, all registers are restored, the line is again selected, and the program returns to the point at which power failed. D. G. Tustin DSN Engineering and Operations Office The Deep Space Network dkcrepancy reporting subsystem and the results of the first year’s dkcrepancy reporting operation using a computerized data man- agement system are briefly described. The general problems encountered and the steps being taken to solve them are discussed. The discrepancy reporting subsystem (DRS), as a part of the DSN Operations Control System, is responsible for collecting and cataloging all documented reports of oper- ational discrepancies. These include hardware and soft- ware failures, as well as procedural problems that occur during any scheduled operation. The cataloged infor- mation must contain a description of the discrepancy, including (1) the facility(s), system(s), and equipment(s) involved, (2) the corrective action taken, (3) the person- nel involved, and (4) the time of occurrences. This infor- mation should be cataloged in such a way as to facilitate rapid retrieval based on varied selection criteria. Approximately one year ago, the DRS data files were placed in a computerized data management system using a prototype program. This system has since proved its worth many times over. It allows for the rapid retrieval of data in endless formats, using practically any available data parameter or parameters as selection criteria, and can be accomplished with a minimum of programming effort. Although this new capability represented a major im- provement in the DRS, several old, as well as some new deficiencies, were present. There are several new param- eters very necessary for report sorting that were not included in the computerized data base, and a definite ability to correlate discrepancy reports (DRs) with the network schedule is required. In addition, the forms used for reporting a discrepancy are not fully compatible with the computerized data base and are time-consuming. Efforts are now underway to correct these known de- ficiencies and to make several other needed improvements. The data base is being expanded to include the needed missing data parameters. It is also being restructured in a manner that will facilitate the use of the DSN schedul- ing data base as a reference file during computer runs. Also, the reporting forms are undergoing a major revision. Working with the various facilities, the DR forms are being reviewed and redesigned in an effort to make them more comprehensive and easier to complete. The method of reviewing corrective action taken, and s, is also being investigated, and several methods of involving the facility and system engineers in this process are being considered. These improvements should all be completed and placed in operation prior to the Mariner in May 1971. W. Kizner DSN Engineering and Operations Office Optimal frame synchronization algorithms are developed which will reject bad data as well as provide high probabilities for obtaining correct frame synchro- nization with data whkh has an error rate consistent with project requirements. The exact analysis to obtain these probabilities is outlined. The amount of com- putation to obtain these quantities may be very large, hence easily computed approximations are also given. The aim of this study is to develop optimal frame syn- chronization algorithms under a very general set of as- sumptions. The definition of optimality can be speciiied to a large extent by the user. The problem of frame syn- chronization may be posed in the following way. Initially, data is received starting from some unknown part of the frame. The beginning of the frame is determined by rec- ognizing a sync word, or a binary sequence of given length which is not likely to be confused with a string of information bits. When further identifications of sync words occur at intervals equal to integer multiples of a length of a frame, then a final determination can be made that frame synchronization has been attained. The figure of merit of an algorithm depends on how reliably good data is frame-synched, and how few errors are made in labeling as synched data which is not prop- erly synched or data which has a large number of errors. Another is the computation time involved. Here, the user can specify his own figure of merit consistent with his goals. One optimal property of the present technique is that if the quality of the data should be seriously degraded after some point, then this algorithm insures that the good frames up to this point are kept and the bad frames after and including this point are rejected. This is in contrast to other algorithms which have a time lag. It also con- trasts to the use of the computed signal-to-noise ratio, which also lags. Definitions of terms used are given in Table 1. es The method consists of three states: (1) Search mode. Initially, the information train is ex- amined one bit at a time in a serial shift register, N bits in length, where 87 is the number of bits in the sync word. When the contents of the shift regis- ter match the sync word, allowing up to E, error bits, the provisional lock mode is entered. A pattern with up to E , errors is called good. EO E, N N , N , pb P C P L P s q L Type I error rype I1 error maximum number of bad patterns allowed in group number of allowable bit errors in a pattern number of bits in sync word number of bits in frame number of patterns in group, or number of representations of sync word that are used to determine mode of algorithm probability of a bit error probability that a sequence of N bits of the information train is confused as a pattern when starting in the wrong position probability of a bad pattern (one which has more than E, errors) probability that a group of Ng sequences of N bits is mistaken for the condition that the system is in lock when starting in the wrong position probability of a good pattern good data which should be in sync but is declared out of sync locking in at the wrong position in NF or fewer tries of consecutive positions (2) Provisional lock mode. The contents of the shift reg- ister are examined only once every N , bits, where N , is the length of the frame. Next, a sequence of patterns, or group, is tested as follows: NB patterns at a time are examined. Thus, N , - 1 new patterns are considered. If N , - E, or more good pat- terns are observed, lock mode is entered. If not, search mode is entered again. (3 ) Lock mode. Here, groups of N , patterns, each spaced N , bits apart, are examined. If N , - E, or more good patterns are observed, the lock mode is continued. Here, a new pattern would be added to the group and the oldest one eliminated. If more than E , bad patterns are observed, the lock mode is discontinued and the search mode is entered again at the approximate location of the next sync word. The frames up to the last good sync word are accepted and the others are rejected. In the model used here, it is assumed that no bits are created or destroyed in error. Hence, the normal assump- tion is that loss of lock is due to bad data. One type of error, called type I, is to declare good data as out of sync or bad. Another kind of error is to lock on data in the wrong position, which we call a type I1 error. Still another kind of error is to include as good data that data which has a high error rate. It will turn out that it is possible to calculate the probability of a type I error accurately, but the cost of computation may be high. Hence, a simple approximate formula is given here. It is not possible to calculate the probability of a type I1 error with any accuracy without a detailed knowledge of the kind of data transmitted, since this error consists of the possibility of confusing this data or part of this data with a sync word. However, some approximate optimistic esti- mates will be given. Now to the calculation of the different types of errors: the criterion for labeling a pattern as good or bad may be generalized from what has been said before to take into account dependence of errors within a pattern. However, it is assumed that the probability that a pattern is bad pL can be calculated, and that the probability that any pat- tern is good or bad is independent of the state of other patterns. A heuristic method of computing the probability of a type I error can now be given. It consists of finding the probability of observing a group of N , patterns with more than E, errors, or 8, prob (type I) = 1 - 22 B (Ng; i , pL) (1) i = O where B (n; k, p ) is the probability of having k ‘csuccesses” in n Bernoulli trials with probability p for “success,” and is the binomial coefficient. A closer examination of the reasoning reveals many flaws. For instance, part of a bad group may be usable. In addition, the algorithm may reject some good groups in the process of losing lock and reacquiring it. Later, an exact treatment will be given. frames that are lost when going out of lock before the process starts again is calculated. To find the expectation of the number of frames lost before lock is achieved, one uses some results from the Consider next the probability Of a type error, Or the probability Of achieving lock at the wrong position. We that the Probability that any particular Pattern Of theory of Markov chains. A state is defined so as to specify the sequence of good and bad in the group. The initial probability of each state is easily found, given W bits is 2-N. If E , errors are allowed, then the probability of confusion of a pattern on a single try when out of sync p c is the probability of a bad pattern pL. The distribution of succeeding states is found by setting up the transition matrix, or the matrix which specifies the probability of (3) i = O The probability of a type I1 error on a single try when out of sync is the probabiliv that the first pattern in the group is confused as good (to gain provisional lock) and that at least No - 1 - E , in N o - 1 are also taken as good, or No-1 p , = P c 2 B (No - 1; ~ P c ) (4) i =No -1 - E g The probability of a type I1 error in N F tries is the prob- ability of a type I1 error in at least one of the N F - 1 bad positions with the correct position rejected. Hence, The method for obtaining the exact estimate of the probability of a type I error will now be outlined. First, we recognize that loss of lock is a recurrent event. This means roughly that the initial situation is restored again after each loss of lock, or that the experiment starts all over again. Next, the experiment is divided into two phases: gaining and losing lock. The expectation of the number of frames that are lost before gaining lock and the number of frames that are accepted when in lock are calculated. Lastly, the expectation of the number of the transition from state i to state j , given state i. States which are in lock are defined as absorbing, and the expec- tation of the time for absorption, or the expected number of frames lost, is calculated. The process is then reversed. Starting with a known distribution of states in lock, the expected number of frames that are accepted (before going out of lock) is calculated. Lastly, a correction is made for some additional frames lost. ilit For uncoded data, the probability of a good pattern qL is easily computed if it is assumed that the bit errors are independent. EP qL = 1 - p L = Z B ( N ; i , p b ) (6) i = O where pb is the probability of a bit error. For coded data, the situation is different. For block coded data, for instance, one would like to try to deter- mine the number of code words in error, rather than the number of bits in error in a sync word, because the num- ber of wrong code words is a much better indicator of the status of the communications system. Also, although errors in code words are independent errors, the resulting bits are not. The problem of defining bad patterns for coded data is being pursued at present. e eai A few numerical results (Tables 2 and 3) will now be given to show the validity of the approximation for type I error and the possibility of obtaining algorithms which meet the objectives stated. First, a few results are given for establishing how accurate the approximation Eq. (1) is for the probability of type error. These results and the others listed suggest that the approximation Eq. (1) T = 1. Poramefer Probability of o bod pottern Expected number of consecutive frames lost Expected number of consecutive Probability of type I error Approximation for probability fromes in lock of type I error, using Eq. (1) Probability of type II error pe = 0.005 pa = 0.05 0.2513 X lo-' 0.1709 2.25 3.09 40,08 1 18.09 0.5622 X lo4 0.1456 0.6287 X lov4 0.2045 0.8061 X 0.9014 X 10"' is good to within a factor of 2 and is increasingly accurate as the probability gets smaller. Another conclusion that we can draw is that it is pos- sible to meet the requirements of having small type I and I1 errors for bit error rates within project require- ments and having a large type I error when the bit error 1 Parameter Probability of o bod pattern Expected number of consecutive fromes lost Expected number of consecutive fromes in lock Probability of type I error Approximotion for probability of type I error, using Eq. (1) Probability of type II error p b = 0.005 0.2514 X lo-' 2.17 26,925 0.8061 X lo-' 0.1316 X lo-' 0.1709 4.09 17.25 0.1916 0.3422 rate exceeds this amount by, say a factor of 8. If NB = 15, E, = 4, E, = 0, p b = 0.005, and the other quantities are unchanged from those used in Tables 2 and 3, then Eq. (1) yields 0.0032 for the approximate type I error. If pb = 0.04, then Eq. (1) becomes 0.89. In other words, there is a good chance that the poor data will be rejected. For this case, the probability of a type I1 error is essentially zero. 1. A. Miccio DSN Engineering and Operations Office The Traceability and Reporting Program is a combination of three programs designed to coordinate and disseminute infomution to meet the needs of research- ers, analysts, and managers for information concerning the DSN mission data record for a current and/or past mission. I t also serves as a monitor or as an accounting device by providing status information relative to the generation of System, Master, and Experimenter Data Records. The program additionally func- tions as an index to the mission data captured on magnetic tape and microfilm retained in the DSN Operational Data Control Center. The Traceability and Reporting Program (TRP) is part of the DSN Operations and Analysis function; it is main- tained and operated by the DSN Operational Data Con- trol Center (ODC). The DSN ODC acts as the recipient point for DSN products, i.e., the System Data R), Master Data Record (MDR), and the Experimenter Data Record (EDR). The ODC utilizes magnetic tape and microfilm as its principal information retention media; these media are also used for user ref- erence and retrieval. The TRP supplements those reten- tion media and functions as an information source to the DSN and its users, providing information concerning the mission data record. Past mission operations have indicated a need for a system to provide information concerning the status and state of operational data once it has passed through real- time monitor and processing functions. The analyst or investigator of past missions, in order to determine what data had been received, its physical state (magnetic tape, disc, etc.), and its quantity and quality, would have to pursue many avenues to acquire the necessary informa- tion. The monitor and data processing functions normally exercise direct influence over operational data from time of acquisition to 24 h after termination of a tracking JBL T b sequence. The TRP begins building its information base for a tracking period during that same period and dis- plays status information within hours after the end of the tracking sequence. The information base is continually updated until all information concerning a data sequence has been acquired and fed into the TRP. The process of information acquisition encompasses the period from end of track up to 2 yr after the end of the mission. The basic capability of the TRP is the coordination and abstraction of information from multiple sources into one source. The TRP delivers to the user decision making, analysis, and visibility support, covering a time period from near-real-time operational interest to long-term research analysis. nction The Traceability and Reporting Program, in three for- mats, provides the DSN, its users, and investigators infor- mation concerning the scientific and engineering data collected through the course of mission operations. The program provides information to effect the reconstruction or upgrading of a data record sequence for a past data day in terms of current demands. In terms of the Information Sciences, the concept of TRP is oriented toward making more efficient use of docu- mented data and its sources generated in the mission data record sequence. The TRP applies discipline to the extraction of infor- mation from the real-time data stream. In essence, the TRP is based upon an eclectic principle of specifically choosing selected information elements from the data stream. The selection criteria for specific element extraction are predetermined by analyzing the linguistic composi- tion of an investigator’s or user’s subject matter with which he is concerned. The elements extracted from the data stream form the basic linguistic units which lend organization and/or structure to the investigator’s subject matter. The program has the flexibility to permit revision of its data base in light of newly reconstructed and/or up- graded data. Through its various elements, the TRP dis- plays information for each mission in chronological sequence under each source for each data day. (A data day is defined as Goldstone set to Australia rise.) Specifically, under format A, or the Basic User Infor- mation Catalog, all participating sources (ranging from DSIF stations to DSN system analysis areas) are repre- sented. All types of material generated by each source are listed in alphabetical and chronological sequence. Within the TRP, a mission sequence number is as- signed to the spacecraft, source, and GMT combination. Since this combination is fixed, the TRP can list all infor- mation relative to that specific combination whenever it becomes available, thus representing a complete ac- count of information for each source for a specific time period. The Traceability and Reporting Program generates three reports: a Basic User Information Catalog, an SDR/ MDR/EDR Status Report, and a Projected Sequence Report. The information content in each report is in rela- tion to the nature of a user’s request. In format A, or the basic catalog, the design criteria assumed a user would lack sufficient detailed information and therefore provides as complete as possible a representation of all available information concerning the data records for any given data day or portion thereof. Figure 1 represents a sample page of the user catalog. It provides the user such information as mission, space- craft identification, originating source for any data listed, GMT, data day, data processing classification (normal- critical, represented as 1-4), and data quality in actual percent. For example, in a configuration such as Pioneer VZZI, DSS 12, GMT day 018, all information received by ODC is represented on one page. To the user, this cata- log can, by any number of edit parameters, isolate a sequence of data for a specific station or for an entire data day. In the SDR/MDR/EDR Status Report, or format B, detailed information is displayed pertaining to the status and state of each individual data record (see Fig. 2). This report will display data record status by spacecraft, GMT, percentage of good data, serial number, Original Data Record availability, reuse or degauss cycle, and data record availability, either magnetic tape or disc. The , while generating specific reports, continu- ally updates its data base, allowing greater research and analysis of past operations. The program operates simul- taneously on two levels: in operational or near-real-time and in an archival mode, defined as 48 h after the fact. a. P I *. ' . . , ; :! c i.P h I 5 age of user catalog DSN DATA RECORD STATUS REPORT STATION 14 s c DATA DAY AOS 1 81 262 194854 21 1 TRACK I NG TELEMETRY COMMAND MON I1 SYSTEM xxx DATA RECORD DISC TAPE DISC TAPE DISC TC ODR SDR MDR 14762 13765 264 14699 DATA QUALITY 14762 90.5 13621 95.5 14699 77,9 ODR SDR MDR @ re By operating the program on both levels, any past mis- sion report may be entered into the data base and further complete the information bank for any particular mission phase. This process allows researchers, analysts, etc., to review a project and be able to reconstruct, via the infor- mation catalogs, entire phases of a mission 2 mo to 2 yr past. The Traceability and Reporting Program attempts to provide as broad a base as possible to allow for user inquiries that lack specific request information. It pro- vides the user information concerning the status, state, quantity, and quality of the DSN data record in near- real-time and archival modes. The program is new and future design planning is presently working toward real- time access and display of data rec d information, and eventual data base integration with SN Operations and Analysis programs. F. E. Bond, Jr. SFOF/GCF Operations Section This article describes the overall teletype configuration that has been developed to support the ground communications requirements established for the Mariner Mars 1971 missions. Primary emphasis is placed on the worldwide distribution of mission traffic formatted in the teletype discipline, routed through the communi- cations switching facilities, and provided to various analysis and control centers. Since the early Ranger missions, space flight projects supported by the DSN have used the teletype discipline of data transmission to send low bit-rate data between the various locations of the tracking network during all phases of each mission. In addition to this primary use of teletype, special multiple-distribution networks were developed within the Space Flight Operations Facility (SFOF) to use teletype as a means of internally display- ing not only the data received from the tracking stations but also the formatted telemetry readouts of data proc- essing computers receiving input data from higher bit- rate disciplines of communications data transfer. Although the Mariner Mars 1971 Project will make extensive use of high bit-rate techniques, such as the high-speed and wide-band systems, to effect transfer of the majority of their data, their use of teletype will be greater than any mission previously supported by the DSN. Current communications planning for missions to follow Mariner Mars 1971 will place far greater emphasis on transmitting data using multiplexed high bit-rate techniques between the tracking stations and the SFOF. This, in addition to the planned use of specialized readout devices within the SFOF to display the outputs of the data processing computers rather than the use of teletype equipment, indicates that the teletype system developed for Mariner Mars 1971 may well represent the most complex and extensive application of teletype that the DSN has or will develop and employ. In view of this, the following article has been prepared to document the overall teletype system to be used by the Mariner Mars 1971 Project. For purposes of clarity, certain omissions have been made intentionally, such as the use of teletype prior to the launch of the spacecraft for purposes of mission simulations, but in general the following information reflects the total use of teletype by Mariner Mars 1971 from the launch phase throughout the entire mission. 1. Many separate activities located throughout the world, including tracking stations, control centers, launch activ- ities, and communications switching centers, have re- quirements for the transfer of data or exchange of information between themselves using the teletype discipline in order to support the Mariner Mars 1971 missions. Many different types of teletype data will be generated by these activities destined in rare cases for a single recipient but far more often addressed to a multiplicity of receiving stations. Sufficient teletype cir- cuits interconnecting these supporting activities (see Fig. 1) have been provided through the resources of the NASA Communications Network (NASCQM) in con- junction with those provided by the DSN Ground Com- munications Facility (DSN GCF). The teletype circuits routed to each supporting activity, regardless of its location, must always be interconnected to one of the four communications switching centers. Each of these centers has the required capability to receive input tele- type traffic from the activities it is designated to support and to switch and forward such traffic to the desired recipients. The Canberra Switching Center supports DSSs 41 and 42 and the Madrid Switching Center sup- ports DSSs 51 and 62 as well as the activities located at Tananarive (TAN) and Canary Island (CYI). The NASCQM Primary Switching Center at Greenbelt, Mary- land supports the activities at Goddard Space Flight Center at Cape Kennedy, the MSFN tracking station at Ascension Island (ACN), and the activities at Ber- muda (BDA) and USNS Vanguard (VAN). The West Coast Switching Center (WCSC) at the Jet Propulsion Laboratory supports the DSN tracking stations at Gold- stone, California and provides a central communications switching center for all internal switching requirements of the SFOF. The quantity of teletype traf€ic, together with the complexity inherent in multiple recipients for such traffic, creates real-time switching functions at each of the switching centers which are performed by spe- cially programmed computers, termed Commprocessors, that are programmed to read and react to addressing information furnished by the transmitting activity. Each teletype circuit, interconnecting the various net- work locations to the Commprocessors, is assigned a teletype address, termed a routing indicator, consisting of four letters used by the originators of teletype traffic to indicate the desired distribution of their traffic. Table 1 lists the routing indicators assigned to the various activ- ities supporting the Mariner Mars 1971 missions. In some cases, a particular routing indicator may be assigned to two circuits at a given activity, where incoming trafEc will be routed to the second circuit by the Commproces- sor in the event that the first circuit is busy with traffic sent at an earlier time. All switching elements of the teletype communications system carrying teletype traffic use these routing indicators, first, to direct traffic to the appropriate switching center using the first letter of the Area California Florida Atlantic Ocean Australia Maryland Spain Africa Activity DSS 12 DSS 14 SFOF Operations Control Net Control wcsc DSS 71 RTCS Bldg A 0 Merritt Island (MILA) MSFN-ACN DSS 4 1 DSS 4 2 MSFN-OC Goddard Real, Time Systen (GRTS) NASCOM DSS 6 2 DSS 51 TAN Location Goldstone Goldstone Pasadena Pasadena Pasadena Pasadena Cape Kennedy Cape Kennedy Cape Kennedy Cape Kennedy Ascension Island Bermuda Canary Island USNS Vanguard Woomera Tidbinbilla Green belt Green belt Greenbelt Ce breros Johannesburg Tonanarive "Must be used in conjunction with preambles. Routing indicators JECO, JELA, JEXC JMAR, JMLA, JMXA JSFOa JOCC JTNC, JNCC, JNDC, JNEC JJPL, JCNF GKEN, GKXE GKAP GKYA GMlL GACN, GSXE GBDA L c l l GVAN AOMJ, AOLA, AOXM? ANBE, ANLA, ANXB GCEN, GUNV GCDF, GDCS GSTS LCEB, LCLA, LCXE LJOB, LJLA, LJXO LTAN routing indicator and, secondly, to direct traffic to the circuits assigned to the recipient as determined and iden- tified by the remaining letters of the routing indicator. Actual All teletype traffic is divided into messages, usually 15 min in duration. Each message is divided into three parts: the header, which contains among other things the addressing information (routing indicator); the body, which contains the data to be transferred; and the end- of-message, which instructs the Commprocessor to dis- connect all circuits established by the header. Simulated In the special case of the SFOF, incoming teletype messages are quite often simultaneously furnished to many areas and internal recipients. As a result, the addressing and the routing to achieve the overall tele- type data 00w becomes rather complex. Two separate methods of message addressing have been developed to simplify these routing problems, both basically controlled by the information provided within the messages them- selves by the originators. The first of these methods, referred to as direct addressing, is used in those cases where the originator of teletype messages knows the desired distribution of his traffic and can therefore ad- dress his messages using the routing indicators assigned to the recipients. This is the standard method of addres- sing teletype traffic and is used throughout the majority of the NASCOM network. In the majority of cases, such as the transmission of tracking data to the SFOF from various tracking stations, the originator is not and can- not be aware of the real-time distribution of his data within the SFOF and thus cannot directly address his traffic to all recipients involved. In these cases, the second method of addressing is used, termed indirect addressing, where the originator of teletype messages addresses his traffic to a pseudo routing indicator, JSFO, which routes the traffic to the WCSC Commprocessor serving the SFOF, and additionally includes information in each message that identifies the source of the data, the space- craft, and the type of data being sent. These three items of information, collectively, are termed a preamble, which is used by the Commprocessor to route incoming messages to any desired set of multiple recipients in the SFOF. The first part of the preamble consists of two numbers (station identifier), which indicate the source of the data. A listing of all station identifiers used in support of Mariner Mars 1971 is given in Table 2. The second part of the preamble consists of two numbers (spacecraft identifier) which indicate the spacecraft to which the data applies. A listing of Mariner Mars 1971 spacecraft identifiers is given in Table 3. The third part of the preamble consists of two numbers (message data Station DSS 12 DSS 14 SFOF DSS 41 DSS 42 DSS 51 DSS 62 DSS 71 RTCS MSFN-ACN identifiers 12 14 20 41 42 51 62 71 70 75 bocasion Goldstone, Calif. Goldstone, Calif. Pasadena, Calif. Woomera, Australia Tidbinbillo, Australia Johannesburg, South Africa Cebreros, Spain Cape Kennedy, Florida Cope Kennedy, Florida Ascension Island tifiers type identifier), which indicate the type of data being transmitted. As an illustration, when DSS 12, having a station identifier of 12 (obtained from Table 2), sends a message pertaining to Mission A, having a spacecraft identifier of 84 (obtained from Table 3), containing track- ing information with identifier 40 (obtained from Table 4), the preamble 12/84/40 would be entered after the routing indicator JSFO. Upon receipt of this message at the WCSC, it would be automatically routed to the data processing computers (for computation), to teleprinters (for page copy), and by means of teletype-television converters to television monitors (for viewing). The man- ner in which this is accomplished is discussed later in this article. Use of the assigned routing indicators and preambles in conjunction with the message switching Commproc- essors and the worldwide network of teletype circuits will collectively provide the necessary teletype capa- bilities required to properly support the Mariner Mars 1971 Project. These capabilities, as provided, are neces- sary, but not, in themselves, sufficient to insure proper 00w of teletype traffic, unless they are used in accordance Ta essagef data type 10 20 21 22 23 24 28 40 42 43 Name Operations Control OPS-x Trajectory RTCS predicts RTCS predicts RTCS predicts Conference Tracking SFOF predicts SFOF predicts ~~ Description Consists of computer formatted sequence of events and real- time schedules as well as manually formatted status and launch messages generated at the SFOF intended for trans- mission to tracking stations. Consists of manually formatted messages sent between the SFOF Net Control and track- ing stations containing infor- mation and instructions relative to a spacecraft tracking pass. Consists of inter-range vector, state vector, /-matrix, orbital elements and liftoff data gen- erated by either SFOF or RTCS intended for transmission to tracking stations. Consists of predicts generated by Air Force Eastern Test Range (AFETR) RTCS intended for transmission to tracking station at Ascension Island and fur- nished to SFOF and MSFN-OC. Consists of predicts generated by AFETR RTCS intended for trans- mission to DSS 51 and furnished to SFOF. Consists of predicts generated by AFETR RTCS intended for trans- mission to DSS 62 and furnished to SFOF; generated only when requested by SFOF in reo1 time. Consists of manually prepared real-time messages between SFOF and DSN tracking sta- tions; used only when normal voice circuits become inoperative. Consists of trocking dato gener- ated by al l tracking stations (including Ascension and excluding DSS 71) intended for transmission to SFOF. Consists of first set of predicts generated by SFOF intended for transmission to applicable tracking stations. Consists of second set of predicts generated by SFOF intended esragef data type 44 45 46 47 50 51 55 70 Name SFOF predicts Pseudo residuals Pseudo residuals Pseudo residuals Telemetry Telemetry Telemetry Command Formats Description for transmission to applicable tracking stations. Consists of third set of predicts generoted by SFOF intended for transmission to applicable tracking stations. Consists of first set of pseudo residuals generated by SFOF intended for SFOF internal distribution. Consists of second set of pseudo residuals generated by SFOF intended for SFOF internal distribution. Consists of third set of pseudo residuals generated by SFOF for SFOF internal distribution. Consists of 8%-bi t / r engineering telemetry data generated by the tracking stations intended for transmission to SFOF in event prime method of telem- etry transmission (high-speed data) becomes inoperative. Consists of 33%-bit/s engineering telemetry data generated by tracking stations intended for transmission to SFOF in event prime method of telemetry transmission (high-speed data) becomes inoperative. Consists of 8%-bit/s engineering telemetry data and edited 50-bit/s science telemetry data generated by tracking stations intended for transmission to SFOF in event prime method of telemetry transmission (high-speed data) becomes inoperative. Consists of spacecraft commands generated by SFOF intended for transmission to tracking stations, and of verification of receipt of these commands generated by tracking sfations intended for transmission to SFOF. Consists of various types of telemetry formats generated by SFOF (360-75) for SFOF internal distribution only. with preplanned instructions as developed and provided by the DSN GCF to all locations transmitting teletype data during the launch and subsequent phases of the Mariner Mars 1971 missions. Table 5 is a worldwide teletype data flow plan that reflects the overall planning furnished to participating locations. Those locations transmitting teletype data are listed in the left column of this table together with the types of data to be trans- mitted by them on each routing indicator assigned to each transmitting circuit. The remaining part of this table illustrates receipt of the transmitted data by all other locations and indicates not only the actikity receiv- ing the data but the routing indicator of that activity to be used by the transmitting location for the transfer of his traffic. All traffic intended for locations other than the SFOF is directly addressed to the recipient, while traffic intended for the SFOF may be either indirectly addressed, by use of the routing indicator JSFO and the preamble as previously discussed, or directly addressed as indicated. Thus, Table 5 reflects both the source and the recipient of all teletype data for the Mariner Mars 1971 missions except for the detailed distribution of teletype data within the SFOF and the backup teletype capabilities, both of which are discussed later. eletype Message ~ w ~ t c ~ i ~ ~ The computer programs residing in the Commproc- essors, shown in Fig. 1, have been designed so as to automatically route any message, properly addressed, from any location in the network to any other location. This technique, termed message switching, is used throughout the teletype system and may be roughly compared to direct dialing techniques used over com- mercial telephone networks in that the user provides the addressing information to equipment capable of interpreting such addressing and establishing the re- quired circuit interconnections to connect the originator to the addressed location. Thus the routing indicator of the teletype system is analogous to the telephone number of the commercial telephone network. Continuing this analogy, the first letter of the routing indicator, either A, 6, L, or J for each of the four switching centers, may be compared with the telephone area codes, since the Commprocessor at each switching center provides a central switching point in the various geographical areas where tracking stations are concentrated. These stations transmit and receive teletype data from the switching centers at 100 worddmin, which roughly corresponds to 50 bits/s. At each switching center, data are forwarded to other switching centers in blocks via 2400-bit/s data lines. By inserting a routing indicator in the header of a message, a transmitting location instructs the receiving Commprocessor to route the message to the desired recipient over the teletype circuit corresponding to the chosen routing indicator. For the duration of this mes- sage, this circuit is not available to teletype traffic trans- mitted from other sources directed to the same routing indicator. Such traffic is held at the Commprocessor and automatically forwarded when the circuit is released upon completion of transmission of the previous message. Release of the circuit is made by the Commprocessor upon receipt of the end-of-message indicator sent by the transmitting location at the end of his message. If pro- visions were not otherwise made, non-transmission or non-receipt of this end-of-message indicator would result in permanent denial of the circuit to all subsequent traffic addressed to it. To prevent this from occurring, the Commprocessors are programmed to count the time intervals experienced between successive teletype char- acters as received from all transmitting locations. Should a time interval greater than 16 s occur, the Commproces- sor interprets this as an end-of-message indicator and releases the circuit. Normally, transmitting locations are capable of formatting their output teletype traffic such that 16-s pauses between successive characters do not occur; thus, this timeout feature does not interrupt traffic. Certain types of data formats, however, are such that pauses between characters greater than 16 s will occur and additional provisions have been made to accom- modate these exceptions. In these cases, the transmitting locations enter the code AA in the header of the mes- sages and the receiving Commprocessor automatically substitutes a 30-min timeout in place of the 16-s limit. Should a transmitting location format traffic having pauses between characters greater than 30 min, then they must transmit an end-of-message indicator subse- quent to such a break and a new header prior to resuming traffic transmission. In addition to having the same capabilities as the other Commprocessors, the Commprocessor at the WCSC is specially programmed to multiple route received mes- sages to many internal SFOF locations depending upon the preamble assigned to each message. Upon receipt of all messages addressed to JSFO, the WCSC Commproc- essor reads the preamble and by referring to a previously entered preamble table, which cross-references preambles to multiple internal routing indicators, receives instruc- tions to route such traffic to all such addresses. Since these preamble tables can be changed in real time by the operators of the WCSC Commprocessor, the capa- bility exists to internally route incoming messages in any 15 E Y 5 0 Y x t; Lu x z Y x Y Y x Y Y x Y Y x Y s -l d 0 r! A m x z 4 U x Y T z e T 4 2 T 5 Y T 5 Y -l z 2 -l U E U U f 7 E U U Z f -l I- 7 U U f T E U U f E -3 U U f -l f T 9 $ 2 T N e 9 9 2 9 N -l N T 9 2 T N Y P Y P Y P v) n 9 2 9 v) n 9 2 9 2 9 0 Y) v) . 0 - 0 " N N U > l ? v) 0 v) . 0 Y) v) . 0 - 0 - 0 N N * > h v) 0 v) . -5 0 v) N v) c n Z Z 4 4 > 5 0 - - > > Y Y Z Z !2 4 2 m YI YI x Y Y ~ Q Q Y X Y Y Y Y 0 0 0 0 0 0 4 4 4 2 4 % 2 2 2 2 2 X Y Y X Y 0 0 0 0 0 f IO L T 1 1 T m ter roeessous manner desired by Mariner Mars 1971 personnel. Figure 2 is a photograph of the WCSC Commprocessor. Two Univac 490 computers are used to provide necessary redundancy. It would perhaps be informative to illustrate, by example, the end-to-end flow of a specific type of data from the originator multiple addressed to the recipients, illustrating the use of routing indicators, using both direct and indirect addressing techniques. Table 5 re- flects that DSS 51 in South Africa must send spacecraft tracking data to the SFOF in Pasadena and to the Real- Time Computing System (RTCS) in Florida. At the SFOF, these data must be supplied not only to the a n t r a l Processing System (CPS) but to users in page copy form as well as visually presented on television moni- tors. The time sequence of this data flow is shown in Table 6. During the launch and subsequent phases of the Mariner Mars 1971 mission, personnel located within the SFQF must receive, monitor, and analyze various types of teletype data. Such streams of traffic may be sent to the SFQF from external locations, or sent from the SFQF to external locations or generated within the SFQF in- tended solely for distribution to internal areas. Each operating position having requirements for page copy of teletype data is equipped with one or more teleprint- ers, each of which is driven by the WCSC Commproc- essor since all streams of teletype traffic, regardless of origin or destination, must be routed through this computer. Three methods of distribution from the Commprocessor to user teleprinter equipment have been developed. The first method interconnects 36 teletype output channels to numerous selector boxes capable of selecting any one of the channels for connection to an 4 P k BEC L T bocation DSS 51 ~~ Madrid Switching Center NASCOM Primary Switching Center RTCS West Coast Switching Center Central Processing System SFOF Mariner Mars 1971 users b. Multiple addresses TTY messages to GKAP and JSFO adding preamble 51 184140. c. Transmits TTY messages to Madrid Switching Center using circuit LJLA at 1 00-wordlmin rate. a. Receives TTY messages from DSS 51. b. Multiplexes messages and transmits to NASCOM Primary Switching Center at 2400 bitsls. ing Center. b. Demultiplexes messages. c. Reads routing indicator GKAP and transmits messages to RTCS at Cape Kennedy at 100- word/min rate. d. Reads routing indicator JSFO and multi- plexes messages to WCSC at 2400-bitls rate. a. Receives messages from NASCOM Primary Switching Center. a. Receives messages from NASCOM Primary Switching Center. b. Demultiplexes messages. c. Reads routing indicator JSFO which instructs Commprocessor to read 51 184140 preamble. d. Accesses preamble table and determines local routing indicators (JDPS and JBWS) to which traffic i s to be routed. e. Transmits messages to JDPS at 40.8 kbitsls. f. Transmits messages to JBWS ot 100 words/ min. This provides data to GCF TTY BUS for page copy and to associated TTY-TV can- verter for TV viewing. This provldes data io CPS. a. Receives messages from WCSC. a. Selects messages by depressing button 10 on TTY selector box of GCF TTY BUS (JBWS) for page copy printout. on TV selector switch for visual presentation. b. Selects messages by depressing button 10 associated teleprinter. This 36-channel bus arrangement is termed the GCF ' ITY BUS and is illustrated in Fig. 3. Teletype traffic fed to any of these channels may consist of any of the various types of data except telemetry. The second method interconnects 26 teletype output channels to numerous selector boxes capable of selecting any one of the channels for connection to an associated teleprinter in a manner similar to the first method. This 26-channel bus arrangement is termed the TLM TTY BUS and is illustrated in Fig. 4. The h t 12 channels of this bus are allocated for displaying teletype telem- etry data incoming from tracking stations (data types 50, 51, 55, and 60). The remaining 14 channels of this bus are allocated for displaying teletype telemetry for- mats generated by the Central Processing System of the SFOF (360-75 computers). These 14 channels on the bus are not sufficient to display all of the CPS formats. As a result, a third method of distribution has been developed which consists of 14 teleprinters directly connected (no selector box) to the Commprocessor, located within the Mariner Mars 1971 operational areas. Figure 5 shows all three of these methods for internal distribution. It should be noted that 24 channels of the GCF TTY BUS and 12 channels of the TLM TTY BUS are connected to TTY-TV converters which provide visual display via television monitors of the traffic carried on these channels. Two additional TTY-TV converters are so configured as to permit Mariner Mars 1971 operations personnel to drive them with any desired output teletype format generated by the 360-75 computers so that these outputs may be viewed on television monitors capable of selecting the video outputs of these two converters. To illiistrate, by example, the manner of distributing teletype date incoming to the SFOF, when the tracking station DSS 12 is tracking the first Mariner Mars 1971 spacecraft and desires to send tracking data to the SFOF, such traffic will be indirectly addressed by DSS 12 to JSFO followed by the preamble 12/74/40. The selection of the identifiers in this preamble follows the outlines given previously. Table 5 indicates that this indirectly addressed traffic when received at the WCSC Comm- processor will be routed to JDPS (the Central Processing System) and to JBSS on the GCF TTY BUS. Figure 5 reflects that JBSS may be selected for page copy printout by depressing the sixth button on the GCF TTY BUS selector box and may be viewed on television monitors. To illustrate, again by example, the manner of distribut- ing teletype data outgoing from the SFOF, when a message is sent pertaining to the fhst Mariner Mars 1971 spacecraft, containing instructions to DSS 14, such traffic would be directly addressed to JMXA and indirectly addressed to JSF followed by the preamble 20/74/20. Table 5 indicates that the JMXA routing indicator per- mits the WCSC Commprocessor to route the traffic to DSS 14 and the JSFO routing indicator together with the preamble is used to internally route the traffic to JBOS on the GCF TTY BUS, to JZED in the Network Analysis Team (NAT) Area, and to JTNC in the DSIF Control h e a . Figure 5 reflects that JBOS may be selected for page copy printout by depressing the second button on the GCF TTY BUS selector box and may be viewed on television monitors. The teletype system used for Mariner Mars 1971 mes- sage traffic has the capability to receive, transmit, and print out on page copy a multiplicity of symbols which can be divided into three categories: 26 symbols repre- senting the alphabetical characters from A to Z (upper case and lower case), 10 symbols representing the num- bers from 0 to 9, and 14 symbols representing punctua- tion and other meanings as follows: Symbol Meaning ~ ~~ Dash, hyphen, or minus sign Number Period, full stop, or decimal point Slash Question mark Semicolon Quotation or ditto marks Colon Dollar sign Apostrophe Comma Open parenthesis Ampersand Close parenthesis. Many receiving teleprinters in the SFOF are equipped with standard type-boxes which, upon receipt of the teletype codes assigned to all of the possible symbols, convert these codes and print out the corresponding symbols. Transmission of plain-text Mariner Mars 1971 teletype messages uses all of these symbols in the normal conduct of routine operations traffic. Certain types of Mariner Mars 1971 teletype messages require that re- ceiving teleprinters have the capability of printing 11 special symbols not normally available in the standard type-boxes. As a result, special modified type-boxes have been developed and installed in appropriate teleprinters 158 I 0 F n -a -rJ C I ) -4 s) -4 -c) -c) -c3 159 which delete certain standard symbols and replace them with the special symbols listed below: Standard symbol ? 7 I! $ I 9 Open & ( ) Converted to + * - 0 1 2 3 4 5 6 7 - - - - - - - - Pronounced Plus Asterisk Equal Bar zero Bar one Bar two Bar three Bar four Bar five Bar six Bar seven Meaning Addition Alarm Equal Ten Eleven Twelve Thirteen Fourteen Fifteen Sixteen Seventeen Thus, a transmitting station desiring to transmit the bar one symbol, meaning eleven, would transmit the teletype code corresponding to the $ and the modified type-boxes on the receiving teleprinters would print the symbol 1. Occasions will arise during the Mariner Mars 1971 mis- sions where any of the recipients of teletype messages in the SFOF may require that certain messages be retransmitted to them a second time. To eliminate the obvious disadvantages of requiring the originator of such traffic to retransmit these messages, provisions have been made in the WCSC Commprocessor that permit the operators of this computer, upon request for such retransmission, to recall the desired messages from the Commprocessor and route these recalls to any teletype channel in the SFOF. It is possible, of course, to route recalled messages to the teletype channel to which the original messages were transmitted, but if this were done, real-time traffic addressed to this channel would be interrupted or delayed by the recalled traffic. To pre- vent this, two special teletype recall channels, JCSR and JCMR, have been established with associated tele- printers located throughout the SFOF to which recall traffic is routed by the operators of the Commprocessor, thus producing page copy for the requesting user. These two channels are illustrated in Fig. 1. Teleprinters con- nected to channel JCSR are equipped with standard teletype type-boxes while those connected to channel JCMR are equipped with Mariner Mars 1971 modified teletype type-boxes in order to display special Mariner Mars 1971 formats. It is entirely possible during a lengthy tracking pass that streams of teletype traffic sent to the SFOF by several tracking stations may arrive at the WCSC Comm- processor at a higher total bit rate than the teleprinter on the output channel can operate. This is especially true in those cases where a particular type of data com- ing from several tracking stations is routed to a single- output teletype channel using stacking techniques. As a result, incoming messages may be queued in the Comm- processor awaiting available output time for transmis- sion and, as a result, the data displayed on the output teleprinter will be delayed. To overcome the obvious disadvantages of this, user personnel in the SFOF may request that the operators of the Commprocessor initiate input instructions to this computer which result in the skipping of one or all messages on queue and routing the latest incoming message to the output channel tele- printer, thus providing real-time data to using personnel. All messages skipped in this manner are not automatically forwarded to the user but may be recalled using the methods previously described for recalled traffic. Within the operational areas of the SFOF, numerous consoles and positions have been equipped with tele- vision monitors, each having an associated selector box permitting the user to select many channels of television displays. These television channels display various types of data including equipment status, codgurations, mis- sion times, area surveillance, and 360-75 computer out- puts. In addition, 38 different channels of both incoming and outgoing teletype traffic may be viewed on these television monitors. Figure 5 identifies these 38 channels, each of which is derived from an output channel of the WCSC Commprocessor and fed to 38 teletype tele- printers individually viewed by the same number of television cameras as shown in Fig. 6. The outputs of these cameras are provided to the selector boxes of each television monitor. Thus, project personnel may select any incoming or outgoing teletype channel for direct viewing on any of numerous television monitors. Figure 7 illustrates a typical Mariner Mars 1971 operations con- sole showing two television monitors with their asso- ciated selector boxes, and Fig. 8 illustrates a typical desk type television monitor. It should be noted that the first 24 of these teletype-television channels are associated with all types of teletype traffic, such as predicts and tracking data, except telemetry, while the remaining 14 channels are associated with telemetry and commands only. Of these 14 channels, 12 are assigned for viewing 1 . II d b S E e incoming raw teletype telemetry from the tracking sta- tions while the remaining 2 channels are user selectable for viewing any of the teletype-formatted telemetry out- puts of the 360-75 computers as derived from the telem- etry data furnished to these computers by the inbound high-speed data circuits. e its The overall design and mission configuration of the entire teletype system provided for the Mariner Mars 1971 Project, by the DSN GCF and NASCOM, requires that all agencies and stations transmitting or receiving teletype traffic be directly connected to one of the four Commprocessors at one of the switching centers. Certain streams of teletype data may pass only through a single Commprocessor, such as traffic originating at DSS 12 at Goldstone, California destined for transmission to the SFOF, while other streams may pass through as many as three Commprocessors, such as traffic originated by DSS 51 in South Africa destined for transmission to the SFOF. In all cases, however, since all teletype traffic must pass through at least one Commprocessor, a failure of any one of the four during critical phases of the Mariner Mars 1971 mission could result in non-receipt at the SFOF of data that could have critical importance to mission personnel. NASCQM, in conjunction with the DSN GCF, has developed limited capabilities which, when activated, will permit certain selected streams of teletype traffic inbound to the SFOF from any of nine supporting loca- tions to be bypassed around the Commprocessors and provided in raw form to mission personnel as illustrated by Fig. 9. Since the interface between any two of the Commprocessors is dependent upon the proper operation of both, then a failure experienced by one necessitates that all Commprocessors in the teletype stream must be bypassed from the originating station to display devices in the SFOF. Although nine different locations are sup- ported by these bypass capabilities, only combinations of two stations can be used simultaneously and combina- tions of DSS 41 with DSS 42 and DSS 51 and DSS 62 are not possible. These restrictions are a result of the availability of only a few discrete teletype circuits to the overseas switching centers and only two display channels in the SFOF. Upon receipt of the two bypass teletype streams, the WCSC connects these to output channels 23 and 24 on the GCF TTY BUS, which may be selected by mission personnel for page copy printout or for television viewing. During the two launches scheduled for Mariner Mars 1971, certain portions of the total bypass capability illus- trated in Fig. 9 will be pre-scheduled and activated by communications personnel. Certain streams of teletype traffic, such as that originated by the MSFN tracking stations at Ascension Island, will be transmitted to the SFOF not only by the normal Commprocessor route but also by the bypass circuits activated for this purpose until loss of signal by the tracking stations. In the event of a Commprocessor failure resulting in interruption of the normal streams, the data arriving via the bypass circuits will immediately be available for use. During subsequent cruise phases of the mission, activation of bypass circuits to DSN tracking stations will be made only as required by either spacecraft maneuvers or space- craft emergencies when receipt of teletype data is vital to operations. Since the prime method of telemetry trans- mission is over the high-speed data system and since this system is independent of Conimprocessor failures, it is not anticipated that these bypass circuits will be used for transmission of telemetry data. A considerable amount of advanced planning has been accomplished for all types of communications support for the two Mariner Mars 1971 missions, not only by personnel of the Mariner Mars 1971 Project and the DSN GCF but by elements of the DSIF, SFOF, MSFN, and NASCOM as well. As a result, the teletype system de- veloped for the support of the Mariner Mars 1971 mis- sions undoubtedly represents the highest development of this communications discipline used for any mission supported by the DSN. J L r - - - - - - - - - - 1 I I I r- - - - I I I I L. - 1 7 \ \ \ I I ) I r 1 r I I _1 I I I -_I 1 I A. Bryan and G. Osborn DSIF Operations Section A simple and inexpensive upper bound technique for the measurement of DSIF effective radiated power is described. Test results verify the theoretical model and imply that DSIF uplink stability can satisfy Pioneer F/G attitude control requ3ement.s. 1. The spin-stabilized Pioneer F/G spacecraft will utilize an automatic attitude control system (CONSCAN) based upon RF conical scanning techniques. The uplink RF signal radiated by the DSIF is the reference for the spacecraft attitude control system. The CONSCAN proc- essor cannot distinguish between RF amplitude varia- tions due to spacecraft aiming errors and those due to fluctuations in the DSIF effective radiated power (ERP). This usage of the uplink signal as a beacon required the DSIF to reevaluate its ERP variations. The variations are due to two uncorrelated sources, uplink antenna pointing errors and power variations in the transmitter. Instability in the transmitter power was monitored with a crystal detector and, with the aid of the frequency translator, the DSIF S-band receiver. The antenna point- ing error was evaluated from the ground receiver auto- matic gain control (AGC) while tracking ALSEP 1, the lunar scientific package. Instability in the uplink due to antenna pointing errors is assumed to be nearly the same as in the downlink signal. ERP variations detected by the synchronous AGC detector are recorded on digital tape and analyzed off- line. In addition to frequency analysis (i.e., a power spectrum), a chart recorder provides a time domain record for comparison. s Software development and validation of the use of the receiver AGC to measure power spectra was done at DSS 71. The spectrum is obtained from the squared magnitude of the discrete Fourier transform in an ensemble averaging mode, wherein many spectra are averaged to reduce the variance of the final estimate. The Fourier transform has a dynamic range of approxi- mately 80 dB, and is adequate for analysis of very small stationary and random signals so long as the thermal noise is not too great. Figure 1 is the program output for 0.406% rms (0.1 dB peak-to-peak) sinusoidal carrier modulation at 0.08 1 .o I I SCALE FACTOR x SIGMA = 0.11 19 MEAN = -129.82 I I I : --4 P PREDICTED? = -47.8 d B pc 0.5 0 I I I 0.0 0.08 0.16 0.24 0.32 0.40 0.48 FREQUENCY, Hz k Hz. The signal was generated by modulating the station test transmitter, and was detected from the receiver AGC. To estimate the sideband power P,, it is noted that the instantaneous power P(t ) can be expressed as P ( t ) = P,(1 + 2 m(t)) with < m (t) > = 0 and < m2(t)> << 1. The ratio of side- band to carrier power is just p E - = < m2(t) > p , For the test case, < m2 (t) > (4.06 X 10-3)2 = 1.65 X or -47.8 dB. To compare this value to the data graphed in Fig. 1, the program output must be inter- preted. The program provides the power density spectrum as a function of frequency (at 0.004 Hz intervals) in the form of the coefficients Co is the carrier term, which is normalized to one. The program returns coefficients representing positive fre- quencies only, while an equal amount of power is con- tained in negative frequencies. Each coefficient other than C, must therefore be doubled to get the total power. In order to improve the spectral response, a gaussian time window truncated at the three sigma points was used, but this technique smears the power in the C, term. This effect requires the C, term to be multiplied by a constant of 2.39I or P , = 2.39 C, The gaussian window does not affect the amplitude of a flat spectrum, and has little effect on the power contained in a window spanning many coefficients. The power contained in such a window (with no strong signals near the end points) is then Pt is the thermal noise power. When Eq. (1) is applied to the test data of the 0.1 dB modulation, the sideband power is -47.5 dB, which compares favorably with the -47.8 dB computed above. A prediction of thermal noise power with respect to carrier power, considering that the receiver coherent detector enhances carrier power by a factor of two relative to thermal noise, is The signal level in dBmW is obtained from the receiver AGC, k is Boltzmann's constant, T is the system tempera- ture, and A F is the bandwidth. Note that the predicted thermal noise level agrees closely with the observed level in Fig. 1. S Using the validated test techniques, pre1,iminary tests of the antenna pointing error were performed at DSS 12. This station tracked ALSEP 1 in both autotrack and APS modes. Figure 2 is a block diagram of the hardware configuration. Results of the APS track are shown in Fig. 3. Note how easily detectable is the 1.6-Hz ALSEP IDetermined from the transform of a constant. I CALI BRATI ON TEST - ANALYSIS C = AUTOMAT Fig. 2. ~ t ~ ~ i o n configur FREQUENCY. Hz ig . in e) frame rate as an amplitude modulation. Also clearly evident (at 1.2 Hz) is the effect of the maser amplitude modulation. The thermal noise level is consistent with the predicted thermal noise level. Within the CONSCAN frequency response band, Eq. (1) yields P J P , = 1.28 X which is equivalent to 0.0098 dB rms amplitude variation. There is no evidence of a periodic component in this region. Tests were performed at DSS 11 on the 20-kW trans- mitter operated at 250 W, 1 kW, and 20 kW. Operation at 250 W was required in order to satisfy an uplink power requirement of the Pioneer F and G spacecraft during initial maneuvers after launch. Essentially the same tech- nique is used as in the antenna pointing error measure- ment, as shown in Fig. 4. Figure 5 shows the results of these three tests. TZle 20-kW transmitter operating at 250 W revealed the most instability, as expected. Note that the result, 0.0035 dB rms in the CONSCAN frequency band of 0.04 to 0.12 Hz, is strictly an upper bound as the receiver noise is also included in this measurement. To obtain a spectrum of the transmitter amplitude instability without the receiver noise, a crystal detector operating in its linear range was used to detect power variations at 1 and 20 kW. With the assumption of no correlation between the antenna pointing error and transmitter instability at 250 W, a vector sum result is 0.0135 dB rms. This figure is to be compared to a project suggested value for the DSIF ERF' variation of 0.0355 dB rms centered at the maximum response of the CONSCAN attitude control system. A relatively simple and inexpensive upper bound2 measurement technique for ERP amplitude variations has been developed and verified. Preliminary tests imply that the DSIF uplink amplitude stability can satisfy Pioneer F/G attitude control requirements. The tech- nique will be implemented at all stations supporting Pioneers F and G, as RF amplitude stability tests will probably be required periodically, as well as following maintenance of subsystems that affect antenna pointing or transmitter stability. 2The real part of the cross power spectrum of two receiver outputs would also provide the desired power spectrum, and would reduce the receiver contribution to the measurement. 147 TEST DIGITAL TAPE TRANSLATOR PLOTTER TELEMETRY COMMAND - PROCESSOR OR DIGITAL INSTRUMENTATION SUBSYSTEM ~ A NALY SI S - I I I T"DC\I,DITCD I U" 'G V 10-6 10-7 10-E 10-5 lo-'( 10-1 FREQUENCY, Hz T J. M. Urech DSlF Operations Section This article documents the continuation of the study described in JPL Space Programs Summary 37-63, Vol. 11, pp . 116-120, “Telemetry Improvement Pro- posal for the 85-Foot Antenna Network.” Four methods of improving telemetry bit error rate performunce by combining data with common time tags from two stations are described in the referenced article. This article discusses the method of a posteriori combination of processed telemetry data. The theory is presented and the results of a scheduled test with a Pioneer spacecraft are shown to be in good agreement. The computer program description is included as an appendix. This article is a follow-up to the original report (Ref. 1) which presented method 4, “processed data combination,” as a potential telemetry improvement procedure. The method suggested the sending of the two data streams via teletype or high-speed data line (already processed by the telemetry and command processor) from the different stations to a central computer located at either of the stations or at the SFO . This computer will compare both information streams and select the best data. It is obvious that the process is only applicable to projects using some kind of error detection technique (such as parity error for Pioneer) which will allow the computer to detect erroneous words in either channel and select the corresponding good word from the other channel. Thus, the probability of having an error after combination is the joint probability of having the same word in error in both channels. This, due to the inde- pendent noise contributions, is equal to the product of error probabilities for each channel, The processed data combination method is not affected by the time delay between the two stations, since the computer may employ a high-capacity telemetry buffer. This method may also be used by any pair of stations sharing a common view period ( SS 11/12, DSS 41/42, DSS 51/61/62). Although the method is ideally applicable to two stations with the same configuration (two ground opera- tional equipment [CQE] or two multiple-mission telem- etry [MMT] stations), the computer program has been developed at DSS 62 for its peculiar configuration- DSS 61 using GQE and DSS 62 the MMT. This config- uration further complicates the computer program as it intends to make the formats used by GQE (DQI-5021-QP) and by MMT (DQI-5020-QP) compatible. The program description is given in the appendkl In the case of uncoded telemetry for the Pioneer Project, we may use the following definitions and approx- imations : P E L = probability for a bit level in error (NRZ-L) PEM = 2PEL (1 - PEL) N 2PEL if PEL is very small, PEN being the probability for a bit error (NRZ-M) P , = probability of a word in parity error, where P,, 1: 8PEL (1 - PEL), and if PEL < lo-, then P , N 8P,, Pu = probability of having an undetected word in error, where N 28PiL N 0.437PiL, if P E L < lo-' This is computed for the case of having two bit levels in error per word, although any even number of bit levels in error per word will be undetected by the parity bit; however, the probability of having more than two errors is much smaller. For the general development of a combining program, two different criteria may be selected: irsf Criterio A word tagged in error is outputted only when that word contains a detected error in both channels. In the case of two corresponding words being different (due to a word with an undetected error), select as the output word the one coming from the channel with the most favorable signal-to-noise ratio (SNR), which also has the highest probability of being correct. Then, the theoretical improvement may be easily com- puted by making a few valid approximations of the normal operating values of SNR (the exact calculations lThe assembly language listing is available by request to author. would be much longer, but for our purpose would not add any useful information). If the subscripts 1 and 2 are used for each channel, under the assumption that channel 1 has a greater SNR, we will have: obability of an output word with a detected error (due to independent noise contribution from both channels) P L = P,, 0 P,, as Pw < 1 this always means an improvement. 2. Probability of an undetected word in error at the output P; = P,,, e Pr2 + P,, Pr1 + P,. , (1 - PlI'Z) = P,, + P,, *P, , - , which means a small increment in the undetected word error rate. These results are shown in Figs. 1 and 2 for the particular case of two channels having the same SNR and therefore, P& = P& and P , . = p,. (1 + p,,.) . Second Criterion A word tagged in error is outputted when both chan- nels contain a detected error, and also if two correspond- ing words are different. Then, using the same notation as before, we will have: robability of having an output word with a de- tected error robability of having an undetected word error at the output P; = P,,- , e Pr, + Pur, e Pc-, These results are also shown in Figs. 1 and 2 for com- parison with previous ones and for two channels having the same SNR, in which case: and P:' = 2P,a P , tu 3 a 0 2 4 6 8 10 ST/NO, d B rd error for a single of ~o~binu~jon By examining both sets of curves (Figs. 1 and 2), one can see that the first procedure yields an almost constant improuement of 3 dB (for the operating SNR range) if the detected word error rate is considered. However, the undetected word error rate is not improved with respect to only one channel; furthermore, this rate slightly in- creases for very low signal-to-noise ratios. For the second criterion, the results are different. The detected word error rate shows an improvement smaller than 3 dB, but still greater than 2 dB in the normal operating range; also, the undetected word error rate is smaller having an equivalent improvement of 1 to 1.5 dB. Based on the above theory, the selection of the criteria to be employed should be made by the Pioneer Project, which should specify the relative importance of decreas- ing the detected word error rate and the undetected word error rate. Nevertheless, the first criterion has been se- lected at DSS 62, since this computer program is not intended to be an exhaustive and definite one, but rather has been developed mainly for feasibility demonstration and testing. The “merging” program was checked out internally at the station, and a DSN scheduled test was conducted on day 260 from 16:OO Z to 18:OO Z at DSS 61/62. The re- sults were satisfactory and in close agreement & h the basic theory, although the statistics obtained for such a short period (1.5 h) cannot be very accurate. The estima- tion obtained was: Percentage of error (avg) Station DSS 62 (MMT) . IB This shows a telemetry improvement close to the theo- etry combination is considered useful to the DSN, the demonstration program will be modified in order to have more capabilities, such as: GOE/GOE combination and/ or MMT/MMT combination, or to use the second cri- terion if required. retical 3 dB. Further tests will be conducted when both stations are available at the same time; but if this method of telem- 1. Urech, J. M., “Telemetry Improvement Proposal for the 85-ft Antenna Net- work,” in The Deep Space Network, Space Programs Summary 37-63, Vol. 11, pp. 116-120. Jet Propulsion Laboratory, Pasadena, Calif., May 31, 1970. 17 T Demonstration program for Pioneer science data merging or Miguel A. Urech, DSS 62. o n ~ i ~ u r ~ ~ i o ~ Any TCP Phase 11-C. urpose Program has been written to demonstrate Method Num- ber 4, Processed Data Combination of “Telemetry Im- provement Proposal for the 85-ft Net” by Jose M. Urech, DSS 62, dated 11 March, 1970. rogr peralors None. Program requires locations 00200 to 00202 and 00300 to 22037. Yimin N/A. Source uage SDS Meta-Symbol. 1. The purpose of the program is to improve telemetry by means of combining two data streams already pro- cessed by TCP. The two data streams used for this combination are: (1) Science TTY output of program DOI-5021-QP at a GOE station, and (2) Science TTY output of program DQI-5020-0P at a MMT station. These two streams are first fed into the “merging com- puter,” through R1 and RO of the communications buffer, where they are compared word by word and the com- bined data outputted through T3 of the buffer with the same format as used by DOI-5021-QP. The program has two modes of operation: In this mode the two data streams are compared word by word employing the following criteria: (1) If both words are the same and different from dollar signs ($$), anyone of the two is outputted and no error is counted. (2) If each word is dollar signs ($$), these signs are outputted and the error counter is incremented by one. (3) If the two words are different, and one of them is dollar signs, the word not being the dollar signs is outputted and no error is counted. (4) If the two words are different and neither is dollar signs, the word coming from the stream with the most favorable SNR is outputted, and no error is counted. The program does not calculate the SNR of the two data streams, but this can be introduced by an input through typewriter. The above-mentioned criteria apply only to words 4 to 16 and 19 to 32 of the data frame. Words 3, 20, and 21 (except in the case of extended frame count for word no. 3) are parity checked in both streams, and the fol- lowing criteria are then used for comparison: (1) If the two words are the same and have correct parity, anyone of them is outputted and no error is counted. (2) If the two words are the same and both have in- correct parity, any one of them is outputted and the error counter is incremented. (3) If the two words are different and only one has correct parity, the one with the correct parity is outputted and no error is counted. (4) If the two words are different and both have cor- rect parity, the word from the prime stream (that of most favorable SNR) is outputted and no error is counted. (5) If the two words are different and both have in- correct parity, the word from prime stream is out- putted and the error counter is incremented. This mode of operation is used upon operator request or when one of the two data streams is not available for comparing in the merging mode (i.e., out-of-lock condi- tions, which mean that no science TTY data is available from the GOE station, nor is there frame synchroniza- tion data from MMT). In this mode of operation the program’s only function is to count errors in the available data stream as it copies each word. The program determines that the DOI-5021-OP data stream is out-of-lock by means of the time tags on each frame. When the time difference between one frame and the next is not the expected one ( ~ 2 s), the program assumes that there is missing data caused by an out-of- lock condition in the GOE computer. This means that the program will not go into a “count error only mode” until this data stream is back in lock, and time gap is determined. The out-of-lock condition in the DOI-5020-OP data stream is easily detected by means of non-frame syn- chronization data. The other most important features of the program are: The program outputs percentage of error (PE) messages every 10 min through the T T Y and also on the console typewriter. The PE is calculated by the simple algorithm: number of words in error = number of words processed The program monitors ground receiver AGC and SPE, either from the TTY data coming from the COE station, or directly through the analog-to- digital converter at the MMT station. The program copies any command messages ap- pearing in the TTY data from GOE station. Ensure that the TCP-11-C computer is configured follows: (1) MMT TTY data stream is a patched to “RO.” (2) GOE T T Y data stream is patched to “Rl.” as (3) Batteries are on “T3,” which is the SCP used for output. Ensure that the following patches are made on the interrupt patch panel: BLUE 6 TO GREEN 5 (100 pps) ORANGE 21 TO GREEN 6 (TTY OUT) BLUE 11 TO GREEN 7 (1 pps) BLUE 19 TO BLUE 20 (ADC CONVERT) eraiional Procedures Load program from paper tape with “standard fill” procedure. Once loaded, the program requests initialization param- - - (mainly header information) which are the rollow- STATION ID = Type the two digit number of the station. Example: STATION ID = 62$ SPACECRAFT ID = Type the two digit S/C number Example: SPACECRAFT ID = 20$ DAY OF MONTH = Type the two digit day of month Example: DAY OF MONTH = 09$ BIT RATE IS = Type one of the two legal entries, either 08 or 16. Example: BIT RATE IS = OS$ CHANNEL DESIGNATOR = Type the three-letter designator. Example: CHANNEL DESIGNATOR = GEB$ MESSAGE COUNT = Type a three-digit number for the first header Example: MESSAGE COUNT = 003$ CHANNEL INDICATOR = Type a one-letter channel indicator Example: CHANNEL INDICATOR = B$ PRECEDENCE = Type a two-letter precedence code Example: PRECEDENCE = SS$ ROUTING 1= Type a four-letter routing indicator or a 0 (zero) if it is not required. Examples : ROUTING 1 = JSFO$ ROUTING 1 = 0 $ ROUTING 2 = Same as item 9. ROUTING 3 = Same as item 10 Thus, up to three routing indicators may be re- quested . STATION INDICATOR = Type the four-letter station indicator. Example: STATION INDICATOR = LCEB$ TYPE 1 IF MMT IS PRIME, OR 0 IF 502 PRIME Type 0 or 1 to indicate which stream has the most favorable SNR. TYPE 1 TO MONITOR MMT AGC, or 0 FOR 5021. Type 0 or 1, depending on which stream (GOE station or MMT station) is desired to monitor the ground receiver AGC and SPE. As soon as the last input is entered the program types : PROGRAM READY DOY HHMMSS This indicates that the program is ready and will try to go to the “merging mode” unless forced into a “count error only mode” by breakpoint setting. Breakpoint 2 Set: Go to “count error only mode” with GOE data. Breakpoint 3 Set: Go to “count error only mode” with MMT data. Breakpoint 4 Set: Request new Nascom header. (1) R01/, RQ2/, and R03/ are used to request, change, or delete a routing indicator. Examples : R02/ JAER$ R02/0$ (to delete) (2) MCT/, followed by a three digit number, is used to change the Nascom header sequence number. Example: MCT/008$ (3) EOM/$ Put an EOM on the TTY line and terminate the program. (4) AGC/ Type either 0 or 1, depending on which AGC and SPE is desired to monitor. (5) RSP/$ In the event of garbled characters appearing in the GOE data stream, synchronization of this stream could be lost for comparison purposes. Under these circumstances this message would serve to resyn- chronize the stream. (6) RSM/$ Same as item (5) but for the MMT data stream. (7) BRT/ Is used to change the nominal bit rate, either 08 or 16. (8) MMT/$ Is used to indicate that the prime stream is now that of the MMT station. (9) PNN/$ Same as item (8), but GOE station is the prime. (10) STR/YYYYY/XXXXXXXX$ Breakpoint 1 Set: Requests typewriter input in the case Used to alter the contents of a memory location of operator wanting to make any real-time requests. (YYYYY). (11) EXM/XXXXX$ tion and has typed at least ten frames. This is to avoid wrong time tag on the frames. Used to display the contents of a memory location (XXXXX) on the console typewriter. A sample printout is shown in Fig. A-1. Since storage b d e r s are capable of holding only 50 data frames (about 10 min of data at 16 bps), any out- of-lock condition detected on one of the two streams and lasting more than 10 min will cause some data from the other stream to be lost. Therefore, the operator should set either B.P. 2 or 3 to force a “count error only mode.” The program should not be initiated in merging mode until the MMT stream has achieved frame synchroniza- * * * PN SCIENCE DATA MERGING PROGRAM * * * STATION ID = 62$ SPACECRAFT ID = 2$$ DAY OF MONTH = 18$ BIT RATE IS = @8$ CHANNEL DESIGNATOR = CEB$ MESSAGE COUNT = @l$ CHANNEL INDICATOR = B$ PRECEDENCE = SS$ ROUTING 1 = JSFO$ ROUTING 2 = $$ ROUTING 3 = JACR$ STATION INDICATOR = LCEB$ TYPE 1 IF MMT IS PRIME OR $ IF 5021 IS 1 $ TYPE 1 TO MONITOR MMT AGC OR $ FOR 5$21 1$ PROGRAM READY 261 114833 261 1150q0 PE 0000 261 120000 PE 0658 261 160000 PE 0016 261 161000 PE 0344 261 162000 PE 0049 RSP/$ 261 162612 RSW$ 261 162754 261 163000 PE 0076 261 164000 PE 0015 - 7 . n F. D. Mclaughlin DSlF Engineering Secrion The status of construction of two 64-m-diameter steerable paraboloid tracking antennas being installed near Canberra, Australia and Madrid, Spain is presented for the January/Februay 1971 reporting period. The antennas are being con- structed by the Collins Radio Company, Dallus, Texas, under TPL contract. Two existing 26-m-diameter antenna stations at over- seas locations are being upgraded to 64-m antenna sta- tions. This effort includes the construction of two 64-m antenna instruments and requisite support facilities. (See Refs. 1 and 2 for earlier reports on the upgrading effort.) A fixed-price contract was effected with the Collins Radio Company, Dallas, Texas, on August 14, 1969. The contractual period for construction of the Spanish antenna is 40 mo and that for the Australian instrument is 31 mo. The contract is presently in its eighteenth month. Figure 1 shows a cross section of the 64-m antenna and identifies the major components. Figure 2 lists the sub- contractors to the Collins Radio Company who are re- sponsible for the fabrication of these components and gives the status of the components through the January/ February reporting period. As a structural support element, the Australian antenna pedestal (Fig. 3) is essentially complete. However, the architectural and engineering portions of the pedestal interior and placement of facility equipment within the pedestal have yet to be completed. Atop the pedestal, the azimuth radial bearing runner and the azimuth bull gear have been installed and the hydrostatic thrust bearing runner has been positioned in @ SUBREF 3 COMPONENT SUBCONTRACTOR I (FIG. 1) 8 G3 HYDROSTATIC BEARING RUNNER PADS HYDRAULICS AZIMUTH GEARS BULL GEAR REDUCERS AZIMUTH RADIAL BEARING RUNNER TRUCKS CABLE WRAP INSTRUMENT TOWER/WIND SHIELD ALIDADE STRUCTURE BUI LDI N G ELEVATION GEARS BULL GEAR REDUCERS ELEVATION BEARING MASTER EQUATORIAL ROOM TIPPING ASSEMBLY ELEVATION WHEEL BACKUP STRUCTURE REFLECTOR PANELS QUADRI POD SERVOb TRIC ONE^ ELECTRONICS HYDRAULICS WESTERN GEAR WESTERN GEAR HYDRANAMICS PHILADELPHIA GEAR PHILADELPHIA GEAR WESTIN GHOUSE WESTERN GEAR CAPITAL WESTWARD PRECISION FABRICATORS PRECISION FABRICATORS PRECISION FABRICATORS PHILADELPHIA GEAR PHILADELPHIA GEAR NATIONAL STEEL 8. SHIP BUI LDlN G LECKEMBY CO. COEUR D'ALENES COEUR D'ALENES RADIATION SYSTEMS INC PREC IS1 ON FABRICATORS COLLINS HYDRANAMICS 'SEPARATE CONTRACT WITH PHILCO FORD; FOLLOWS COLLINS EFFORl b~~~ ILLUSTRATED SCHEDULED COMPLETED 0 FABRlCATl ON A DELIVERY TO SITE V INSTALL 0 TEST A A 1970 AUSTRALIAN SITE v o v o A V O 0 A V 0 0 v o A V 0 V v o V O A V 0 c1A V 0 V 0 c 1 A V A V 0 A V O 0 A V c7 A V 0 c 1 A V 0 A V 0 '971 1972 I SPANISH SITE or co preparation for final alignment and grouting. The remain- der of the runner sole plates, the last items required for completion of installation of this critical path element, was received at the site on February 16. Grouting is expected to begin near the end of this reporting period. During the next reporting period, the hydrostatic bearing runner grouting will be completed and erection of the alidade structure will begin. !I. Construction at the Spanish site began on June 22,1970. Figure 4 shows the status of the pedestal construction. Lift number 8 was completed on February 10, 1971. Figure 5 shows an overall view of the construction site. During excavation, granite outcroppings were encoun- tered which required considerable blasting and slowed progress. The site construction, however, is well ahead n~emma pedestal at of schedule. LIFT NO. LIFT NO. LIFT NO. LIFT NO. LIFT NO. LIFT NO. LIFT NO. .LIFT NO. 14 m 2 w A COMPLETED LIFT NO. 9 LIFT NO. 5 LIFT NO. 3 :::: ig. 5. View of Spanis site 1. Casperson, R. D., and Lord, W. W., “Overseas 210-ft-diam Antenna Project,” in The Deep Space Network, Space Programs Summary 37-65, Vol. 11, pp. 1 5 4 158. Jet Propulsion Laboratory, roll, G., and Kushner, E., “BSS 61/63 Facility Modifications ,” in The Deep Space Network, Space Programs Summary asadena, Calif., Nov. asadena, Calif., Sept. 30, 1970. 37-66, Vol. 11, pp. 154158. Jet Propulsion Laboratory, 30, 1970. ‘g NASA - JPL - Coml., LA., Calif.

The deep space network, volume 2 Progress report, Jan. - Feb. 1971

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