Author name used in this publication: K. W. E. ChengAuthor name used in this publication: D. SutantoVersion of RecordPublishe

Chan, HL

Cheng, KWE

Sutanto, D

Author name used in this publication: K. W. E. ChengAuthor name used in this publication: D. SutantoVersion of RecordPublishedVoR allowe

PolyU Institutional Repository

IEEE 1999 International Conference on Power Electronics and Drive Systems, PEDS'99, July 1999, Hong Kong. A Novel Square-Wave Converter with Bidirectional Power Flow H.L. Chan', K.W.E. Cheng' and D. Sutanto3 The Hong Kong Polytechnic University, Hung Horn, Hong Kong. Fax: (852) 2330-1544 Email: (1) eehlchan@inet.polyu.edu.hk (2) eeecheng@polyu.edu.hk (3) eesutant@polyu.edu.hk Abstract - A novel ZVS phase-shift controlled bidirectional DC- DC convertor iis proposed. It possesses the conventional features of phase-shifed convertors, including constant switching frequency, zero-voltage switching (ZVS) and voltage controlling by phase-shift. However, the new convertor has been improved by developing a tii-directional power flow capability and applying synchronous rectification, hence the on-state voltage drop of the active devices are small. Therefore it is recommended for use in electric vehicle (EV). In this paper, the principle of circuit operation of thle proposed convertor was described in detail with mathematical calculations, and the experimental results were also given to verify the concept. I. INTRODUCTION In this paper, a novel bidirectional DC-DC convertor is proposed [l], which is an excellent candidate for high voltage and high power applications [2]-[5]. The newly developed bi- directional feature is especially useful for regenerative braking in EV, where the mechanical energy is converted into electrical energy by traction motor and then fed back into the batteries. The converter employs the phase shift technique for voltage control; utilizes parasitic capacitance of active switches and leakage inductance of transformer for resonant switching; does not require a large filter inductor and applies the synchronous rectification [B]-[7] for bidirectional power flow. The bidirectional phase-shifted DC-DC convertor is comprised of an inverter bridge (Ql-Q4) and a convertor bridge (M1-M4), which are connected with a high fiequency power transformer. The 50% duty-ratio gate signals applied to the convertor bridge are synchronized with those applied to the left-leg of inverter bridge. Figure 1 shows the topology of the proposed convertor. b e  1 : Bi-Directional Phase-Shift DC-DC Converter All the active switches in both the inverter and convertor bridges are isdentical, each of them consists of a MOSFET, a body diode and a capacitor. The capacitors, C1-C4, represent the sum of stray capacitance of the MOSFET and an additional resonant capacitor. L l k  is the total leakage inductance of the transformer's primary and secondary windings. C,, and CO,,, are the input and output filter capacitors. 11. OPERATIONS OF THE CIRCUIT The operations of the proposed converter are different when the converter operates under light and heavy load conditions. Figure 2 shows the waveforms of the converter under the condition of light load, where the primary current of transformer is only sufficient to switch on Q3 (e4 at negative cycle) under zero voltage. On the other hand, Q2 (Ql at negative cycle) is turned on in a manner of hard switching, because the light load results in small primary current, which will flow in a reverse direction through the body diode of Ql (e2 at negative cycle) after the energy stored in the leakage inductance of transformer is empty. Figure 2: Circuit Oueration under Light Load Condition [Slam- I Un I UnTlrsniillon Figure 3: Circuit ODeration under Heavv Load Condition 0-780 3-5 76 9-819 9/$10.00 0 1 9 9 9IEEE 966 Authorized licensed use limited to: Hong Kong Polytechnic University. Downloaded on July 15, 2009 at 00:05 from IEEE Xplore.  Restrictions apply.Under the heavy load condition, the converter can thus operate under ZVS for all active switches as shown in Figure 3, which also illustrates the complete cycle of operation for the proposed converter. V,,, to V,,, are gate signals applied to the switches Ql to Q4. In addition, Vg,l is also used to drive MI and M4, Vgy2 to drive M2 and M3, such that the converter bridge can operate in a manner of synchronous rectification. The synchronous rectification allows the converter to have low on- state loss and ease of control at the transistors MI to M4. The positive half cycle is the same as the negative half cycle except the signs of voltage and current are reversed, so only the positive half cycle of circuit operation is explained here. The principle of circuit operation can be divided into the following five modes as shown in Figure 4a-e. Mode 1: Ql and Q4 are on. Source voltage V,, is applied across the primary winding of transformer, and the secondary winding of transformer is clamped at the level of output voltage V,,,, because MI and M4 are on. As energy has been stored in primary and secondary leakage inductances of transformer during the previous mode of operation, then it is returned to the source. U Figure 4a Mode 1 of Operations Mode 2: The difference between Mode 1 and Mode 2 is that the primary current is positive, or the source is charging up the leakage inductances of transformer with the same slope. In this mode of operation, the source is supplying energy to the load. Figure 4b: Mode 2 of Operations Mode 3: Ql remains on, but Q4 is being turned off. Energy stored in leakage inductances is charging the parasitic capacitance of Q4 (i.e. C4) and discharging that of Q3 (i.e. C3). D3 will conduct once the voltage across C3 is zero. Subsequently, Q3 is turned on under ZVS. It is clear that a dead time between turn-off of Q4 and turn-on of Q3 is requisite for Q3 to achieve the ZVS. Mode 4: Ql and Q3 are on. Energy in leakage inductances continues to deliver to the load, and the large output filter capacitor maintains the secondary side of transformer at voltage level of V,,(. I T Mode 5: Q3 remains on, but Ql is being turned off. CI is being charged up and C2 is being discharged simultaneously by the energy stored in leakage inductances. Once voltage across C2 reaches zero, D2 conducts and Q2 switches on under ZVS condition. Also a dead time between the turn-off of Ql and turn-on of Q2 is required for ZVS of Q2. *:L M, Figure 4e: Mode 5 of Operations 111. MATHEMATICAL CALCULATIONS In this section, we will calculate the requisite dead times for the operations of modes 3 and 5, and the DC characteristics of the converter as well. Some parameters are used, namely D = phase shift, n = N1/N2 and T = switching period. In addition, calculations were made in accordance with an assumption that the C,,,,, is sufficiently large to hold the K,,, at nearly constant. Left-Leg ZVS Transition: ( t j  < t I t6 ) It takes place in mode 5 operation, where Ql is being turned off and Q2 is being turned on. To reset the duration of time t5 < t I t6, we use 0 < t I &-, and the equivalent circuit of this transition is shown in Figure 5. T 2 I ! 1 v"' * 4 U Figure 4d: Mode 4 of Operations I I Figure 5 :  Left-Leg ZVS Transition The initial and final states of circuit elements are given as: I -  I Figure 4c: Mode 3 of operations 967 Authorized licensed use limited to: Hong Kong Polytechnic University. Downloaded on July 15, 2009 at 00:05 from IEEE Xplore.  Restrictions apply.The circuit equations can be obtained by KCL as follows: (1) di(t) ~ I I  -k Y.wt + Llk - - vc:2 ( t )  dt M' Figure 6:  Eauivalent Circuit of Right-Lee ZVS Transition The initial and final states of circuit elements are given as: T 2 T 2 i(& -) = i(t3) = Z4. v(.3 (6, -) = 0 .  ' ( 9 4  ( 6 3  -1 = V,n  3 vC.3 (0) = V,n  7 V(.4 (0) = 0,  T 2 i(0) = i(t3) = 4,  dv,,, ( t )  i ,  ( t )  = C', -. dt Then, solving (1) to (3), and taking Laplace transforms with the initial values of i(O-) and vcl(0-), we obtain (4) The circuit equations can be obtained by KCL as follows: ( 5 )  . c 2  (9) dv,:, ( t )  i4 ( t )  = C4 -. dt + Cxfm,. For optimal design, simultaneously the voltages across Ql and Q2 should be risen from OV to V,, and VI, to OV respectively afer  time t = 6' T / 2 , which is the minimum dead time for left-leg ZVS transition. To evaluate S, T I 2  , we take the derivative of (5) with respect to t and get: Then, solving (8) to ( 1  0), and taking Laplace transforms with the initial values of i(O-) and vc4(03, we obtain sin(wrIghtt) + I3 CoS(Wr&tt) 2 ( l  '1 Y n  - n t u t  i ( t )  = I I where wnght = ~ , Zrfght = \j_i.L and c r , g h ,  = c3 By comparing vc4(t) of (12) with vCl(t)  of (5) and i(t) of (1 1 )  with i(t) of (4), we find that & T / 2 for vc4(t) resonating from OV to VI, is much shorter than & T / 2  for vCl(t)  resonating from OV to VI, on the grounds of (VI, - n V,,,) > -n V,,, and Z3 > Z,. In other words, the dead time 6; T /  2 for right-leg ZVS transition is much shorter than the dead time &,TI2 for left-leg ZVS transition. Thus, (1 1) becomes \ I '  Llk 'right Cright + C 4  -t Cxfmr. However, to ensure the current flowing in the leakage inductance of transformer is large enough to achieve the ZVS at left-leg, v ( . ~  (6,T 12) must be larger or equal to V,,. Then, we substitute (6)  into (5) and evaluate Z5 that is the limit for the converter to achieve ZVS. --(7) Right-Leg: ZVS Transition: (t3 < t 2 t4 ) This transiti-on occurs in mode 3 operation, where Q3 is being turned on and Q4 is being turned off. To reset the duration of time t3 < t 4 tLl, we use 0 < t 4 & TI  2 , and Figure 6 shows the equivalent circuit of the operation. I L I 968 Authorized licensed use limited to: Hong Kong Polytechnic University. Downloaded on July 15, 2009 at 00:05 from IEEE Xplore.  Restrictions apply.Since I ,  is always larger than 15, and if the condition of (7) is fklfilled, then the ZVS at the right-leg will be definitely accomplished. And (12) becomes Therefore, the minimum dead time for right-leg ZVS Y n  ‘right transition is - 1 3  DC Characteristics re-arrange the circuit operations into three stages as follows: To derive the DC characteristics of the converter, we T 2 “On” Stage: (tl < t I t 4 )  or (t4 - t l  = D - ) Since modes 1 and 2 are identical when they are considered under circuit theory, and the primary current changes in mode 3 is small (i.e. I3 c I4 as indicated in (13)), so this stage includes modes 1, 2 and 3. In addition, we also neglect the dead time & - as it is insignificant when compared with D. T 2 T “Off’ Stage: (t4 < t I t5) or t5 - t4 = (1 - D - 6,) - 2 In this stage, the energy stored in leakage inductance of the transformer is freewheeling to drive the load, it operates in mode 4. (1 - D - 6 , ) ~  =I, -I4 Llk - n v o u t  where Is = i(ts). T “Transition” Stage: (t5 < t I t6) or t6 - ts = 65 - 2 This stage is the changeover of the “Off’ stage to the “On” stage, it is actually the mode 5 of operation. di - V,, - y2YIut = L, - dt where I6 = i(t6) = - i ( t l )  = - II. We solve (16) and (17), qn6,T + (1 - D)nVo,,T = I, +I, . 2Llk (17) The input energy W,, to the converter during a half cycle is given by: T D- 1 W,, = J i ( t ) .  vn dt 0 T 4 3 w,, = (I1 +I,)Y,D- Then, substitute (1 8) into (1 9), and we get And the output energy W,,, of the converter for a half cycle is obtained by: w =-.- T Vout2 OUI 2 RLoad Assumed that WO,, = W,,, :. I( FJ -nkD(1- D ( F )  - k6, D = 0 1, (20) I J , and this is the voltage conversion ratio of where k = -the converter. ‘Load’ 4Llk Control Region Equation (20) indicates that V,),,, is not directly proportional to the phase shift D, and it rather shows a second-order relationship between Vo,lI and D. In order to define a linear control region of phase shift for the converter, we then calculate a phase shift D,, for the maximum output voltage. Thus, differentiate V,,, by D to obtain the maximum output voltage and get the DmW: Therefore, the output voltage reaches its maximum when phase shift = Dmw. However, equation (7) sets a constraint for I, to exceed a minimum current in order for the converter to achieve ZVS, then we evaluate Is in term of phase shift. 969 Authorized licensed use limited to: Hong Kong Polytechnic University. Downloaded on July 15, 2009 at 00:05 from IEEE Xplore.  Restrictions apply.First, we solve (1 5 ) ,  (1 6) and (1 7) together, and get Then, put (22) into (7), and assumed that D > 6, and 1 > 265, This is the minimum phase shift for the converter to achieve zvs. By utilising the equations (20), (21) and (23), we can then design the proposed converter with a linear control region where the Z\'S is ensured. Figure 7 demonstrates the typical DC characte:ristics and control region of the proposed converter. DC Characteriscs and Control Region In this cordrol region, wth hard switching. In this control region, to the phase shift. In this control region, the converter operates with ZVS and linearly with phase shift. I 0 0  0 1  (12 0 3  0 4  0 5  0 6  0 7  0 8  0 9  I Figure 7 DC Characteristics and Control Region of the Prooosed Converter IV. EXPERIMENTS There were two experiments carried out to test the performance of the proposed convertor's efficiency and reverse mode of operation. Experiment A A prototype of the proposed convertor was built up with the specifications: switching frequency of 1 OOkHz, input voltage of 70V and output voltage of 30V. The circuit components included: all of MOSFETs (el-Q4 and MI-M4) are IRF530N, total leakage inductance (Llk) of transformer's primary and secondary windings is 2 1 .SpH, equivalent resonant capacitance (Cl-C4) is 3.:2nF, input and output capacitance (Ci, and CO,,) are 5.4pF and 14.8pF respectively. Moreover, the power transformer was constructed using a bobbin of ETD49 and Ferrite core material of 3C85, its turn number of primary winding ( N I )  and secondary winding (N2) were 70 and 40 respectively. Under the load range from 60W to 120W, the conversion efficiencies were measured and plotted in Figure 8. It is obvious that the overall efficiency of the proposed convertor is more than 91% over the specific load range. 711 8 0  ?,I 1110 ou,pt P".rr(W, Fiwre 8: Conversion Efficiencies The finding in Figure 9 and Figure 10 can be used to account for such high efficiency. Zero Voltage Switching of MOSFET I -80 ' Time (us) I I I I -  -15 J I Time (us) Figure 9: Zero-Voltage Switching of MOSFET Qi Zem-Voltage Switching: DninSourre Voltage vs Dmin Cumnt I 15 I ,of 6o I 20 I IO . I I I '  Dram Cumnt (A) Figure 10: V-I of MOSFET 0, It was clearly shown that the drain-source voltage of Q4 has been resonated to zero before the gate-source signal is applied. With the aim of this switching method, the switching losses of MOSFETs are minimized, and the overall efficiency of the convertor is maximized. The other MOSFET switching waveforms are similar. Experiment B bidirectional power flow of the convertor. Another setup was used to test the performance of 970 Authorized licensed use limited to: Hong Kong Polytechnic University. Downloaded on July 15, 2009 at 00:05 from IEEE Xplore.  Restrictions apply.t=0.5s Bidirectional -” -. .+ 8 40. 2 2 0 -  5 0 >0 ,  V. CONCLUSION 1 T 7 0 v e  T Figure 11: Testing Setup for Reverse Power Flow As illustrated in Figure 11, the convertor frrst ran in forward mode with V,,, = 7QV, I,,, = l.O8A, K,:,,,= 36V and I,,,, = 1.95A. Then, at t = 0.5s, an external voltage source of 45V was applied momentarily to the output of the convertor to force reverse power flow. Consequently, the output current and input current flowed in reverse direction simultaneously with amplitude of 1.9A and 1.1A respectively, however, the input voltage almost kept constant at 70V either in forward and reverse operations. Figure 12 shows the changeover of operations. I 100 t l  ‘ I  02 0 4  0 6  0 8  Time (s) I - 1  5 J I Time (s) 3 11 A novel bidirectional phase-shift controlled DC-DC convertor was proposed. The converter uses leakage inductance of the transformer and parasitic capacitance of active switches to perform resonant switching. All the active switching devices are switched at zero voltage switching. The proposed convertor, if employed in EV, is capable of not only transferring energy fiom EV batteries to powertrain for motoring, but also returning energy back into batteries in case of regenerative braking. Moreover, by making use of a simple drive circuit for the convertor bridge, it can gain the benefit of synchronous rectification. ACKNOWLEDGEMENTS This project is supported and fimded by The Hong Kong Polytechnic University, and the project account code is G- V305. 1. 2. 3. 4. 5. 6. 7. REFERENCES Chan H.L., Cheng K.W.E. and Sutanto D., “Bidirectional Phase-Shifted DC-DC Converter”, IEE Electronics Letters, 1’‘ April 1999 Vol. 35 No.7 Forsyth A.J., Evans P.D., Cheng K.W.E. and Ai-Mothafar M.R.D., “Operation Limits Of Power Converters For High Power Ion Engine Control”, AIDAA/AIAA/DGLWJSASS 22nd Int ‘I Electric Propulsion Con$, 1991. Forsyth A.J., Evans P.D., Ai-Mothafar M.R.D. and Cheng K. W.E., “A Comparison Of Phase-Shift Controlled Resonant And Square-Wave Converters For High Power Ion Engine Control”, Proc. Ofthe European Space Power ConJ, 199 1 .  Vlatkovic V., Ridley, Lee F. C. and Cho B. H., “Design Consideration For High Voltage, High Power, Full Bridge, Zero Voltage Switched PWM Converter”, Proc. of the IEEE APEC., 1990, pp. 275-284. Cho J. G., Sabate J. A., Hua G. and Lee F. C., “Zero-Voltage And Zero-Current-Switching Full Bridge PWM Converter For High-Power Applications”, IEEE Trans. On Power Electronics, Vol. 11 ,  No. 4, July, Kazimierczuk M.K., Mescher M.J. and Prenger R.M., “Class D Current- Driven Transformer Center-Tapped Controllable Synchronous Rectifier”, IEEE Trans. on Circuits and Systems-I: Fundamental Theory and Applications, Vol. 43, No. 8, August 1996, pp. 670-678. Yamashita N., Murakami N. and Yachi T., “Conduction Power Loss in MOSFET Synchronous Rectifier with Parallel-Connected Schottky Barrier Diode”, IEEE Tram. On Power Electronics, Vol. 13, No. 4, pp. 523-524. 1996. pp. 622-628. July, 1998, pp. 667-673. 3 J  1 Time (s) Figure 12 Changeover from forward to reverse Dower flow omrations 971 Authorized licensed use limited to: Hong Kong Polytechnic University. Downloaded on July 15, 2009 at 00:05 from IEEE Xplore.  Restrictions apply.

A novel square-wave converter with bidirectional power flow

Abstract

Similar works

Full text

Available Versions

PolyU Institutional Repository