This paper presents novel implementations of bandpass filters in multilayered microstrip technology. The structures presented implement transversal filter topologies of order three. In order to allow for easy implementation of the required couplings, multilayered broadside configurations are proposed. We demonstrate that with the proposed configurations, both dualband and quasi-elliptic responses can be easily synthesized, by only changing the sign of certain entries in the coupling matrix. To easily adjust the signs of the couplings, novel resonators are proposed, including simple half-wavelength transmission line resonators, meander line resonators and short-circuited resonators. Experimental validation for one prototype is presented, demonstrating the validity and usefulness of the proposed configurations.The authors thank Ministerio de Educación y Ciencia of
Spain, which has supported this work under Grant TEC2007-
67630-C03/TCM

Cañete Rebenaque, David

Gómez Díaz, Juan Sebastián

Martínez Mendoza, Mónica

Pascual García, Juan

Álvarez Melcón , Alejandro

English

Repositorio Digital de la Universidad Politécnica de Cartagena

Novel Implementations for Microstrip ResonatorFilters in Transversal TopologyDavid Cañete-Rebenaque 1, Mónica Martı́nez-Mendoza 1, Juan Pascual-Garcı́a 1,Juan Sebastián Gómez-Diaz 1, Alejandro Álvarez-Melcón 11Technical University of Cartagena,Campus Muralla del Mar-Antigones, Cartagena, E-30202, Spaine-mail: david.canete@upct.esAbstract—This paper presents novel implementations of band-pass filters in multilayered microstrip technology. The structurespresented implement transversal filter topologies of order three.In order to allow for easy implementation of the requiredcouplings, multilayered broadside configurations are proposed.We demonstrate that with the proposed configurations, bothdualband and quasi-elliptic responses can be easily synthe-sized, by only changing the sign of certain entries in thecoupling matrix. To easily adjust the signs of the couplings,novel resonators are proposed, including simple half-wavelengthtransmission line resonators, meander line resonators and short-circuited resonators. Experimental validation for one prototypeis presented, demonstrating the validity and usefulness of theproposed configurations.I. INTRODUCTIONThe interest in the development of new filter configurationswith enhanced performances is motivated by the deploymentof new communication systems with increased bandwidthallocations. One of the important issues that must be takeninto consideration for new applications, is the selectivity of thefilter, and the ability to reject unwanted spurious signals thatmay interfere in the crowded electromagnetic spectrum. Also,the increasing difficulties in the selection of communicationchannels, has fostered research in new multiband filters [1].One important research line for filter design in the last yearshas been the development of transversal coupling topologies.This has been motivated by the availability of an exactsynthesis technique of the generalized coupling matrix, afterthe work presented in [2]. In this topology, the input/outputports are coupled to all of the resonators of the filter, whileall inter-resonator couplings are zero. One of the advantagesof this topology is that it provides high flexibility to place amaximum number of transmission zeros in the insertion lossresponse of the filter. A wide class of frequency responsescan be synthesized, ranging from multiband to quasi-ellipticresponses. Normally, all these frequency responses can beeasily implemented by adjusting the signs of certain elementsof the coupling matrix.On the contrary, a big disadvantage of the pure transver-sal topology, is that it is difficult to implement in practicemany coupling paths from the input/output ports to all theresonators of the structure, still maintaining all other inter-resonator couplings to zero. Due to these difficulties, practicaltransversal implementations have been limited up to now tosimple structures of order two [3].In this contribution we propose novel microtrip filtersimplementing the pure transversal coupling topology. Forthe first time, third order transversal filters are implementedusing printed resonators. To increase the number of couplingpaths from the input/output ports to the resonators, broadsidecouplings are used in multilayer arrangements. To easily adjustthe signs of the couplings, three different compact printedresonators are combined in the transversal filters, namelybasic half wavelength printed line resonators, meander lineresonators and central short-circuited stubs. We show that bycombining appropriately all these types of resonators, severaluseful transfer functions can be effectively implemented. Inthis work we will show as an example the design of a multi-band transversal filter and of a quasi-elliptic filter. Measuredresults of one prototype are presented to show the validity ofthe new structures.S L1 32S L312a)                                         b)Fig. 1. Coupling schemes. (a) Single trisection. (b) Transversal third ordertopology.II. NOVEL MICROSTRIP TOPOLOGIESThe coupling and routing scheme of the third-order transver-sal filter, that will be designed in this paper, is shown inFig. 1(b). The dark circles represent resonators, the source andthe load are shown as empty circles, and the solid lines repre-sent coupling between nodes. For comparison, in Fig. 1(a) wepresent a typical topology of a third order filter containing onetrisection [4]. As it can be seen, in the transversal topology allresonators are directly coupled to the input/output ports, whileall other inter-resonator couplings are zero. The purpose of thissection is to introduce new microstrip structures to implementthis transversal coupling scheme.The first step in the design process of a filter is thecalculation of the coupling matrix that synthesizes the desiredfrequency response. It should be noted that for the topologies978-2-87487-011-8 © 2009 EuMA 29 September - 1 October 2009, Rome, ItalyProceedings of the 39th European Microwave Conference775presented in Fig. 1(b), a maximum of two transmission zerosat finite frequencies can be implemented. This can be inferredby the application of the minimum path rule [2]. For the puretransversal topology shown in Fig. 1(b), the coupling matrixcan be easily calculated using the procedure presented in [2].For multiband frequency responses, the procedure describedin [5] can instead be used.Once the coupling matrix has been computed, the next stepis to synthesize the required couplings with a suitable topology[6]. In the next subsections we present several examplesusing novel microstrip structures. These structures containdifferent kind of printed line resonators, and side-couplingsare combined with broad-side couplings, when necessary, toprovide for the required coupling paths. Two strategies areused to adjust the sign of the couplings. The first is to use over-sized resonators (meander line resonators) working at higherorder resonances. The second is to use short-circuited or open-ended terminations, as appropriate, to adjust the correspondingsigns.A. Transversal filter with dual-band operation and two trans-mission zerosThe first example corresponds to a pure transversal topologyas shown in Fig. 1(b), implementing a dual bandpass response.This topology can implement up to two transmission zerosat finite frequencies. By adjusting the signs of the couplings,different responses can be obtained, ranging from quasi-ellipticresponses to dual-band responses. In the case of a dual bandresponse, a transmission zero can be placed in-between thetwo passbands to increase rejection. The other transmissionzero can be used to increase rejection at the other side of oneof the passbands.For the synthesis of the coupling matrix, the techniquepresented in [5], for dual-band transfer functions, has beenused. The filter contains two bands, with a total bandwidthof 30 MHz (lower passband) and 100 MHz (upper passband),and return loss of 15 dB. The coupling matrix obtained usingthe procedure described in [2], [5] is the following:MA =⎛⎜⎜⎜⎜⎝0 −0.357 0.421 −0.563 0−0.357 1.596 0 0 0.3570.421 0 −1.099 0 0.421−0.563 0 0 0.098 0.5630 0.357 0.421 0.563 0⎞⎟⎟⎟⎟⎠(1)For the practical implementation of this filter we select amicrostrip structure with two dielectric substrates broadsidecoupled, as shown in Fig. 2. In the first substrate we printtwo resonators, together with the input/output lines. In thesubstrate on top we print the remaining resonator. In this waywe can obtain a pure transversal topology of order three, witha very compact microstrip structure. We can observe in the(MA) coupling matrix (1), that the couplings of two resonatorschange in sign from the input to the output port. This can beobtained by using simple (λ/2) printed line resonators. Onthe contrary, the signs of the couplings for the third resonatormust be the same. To avoid the sign change, a printed line                                                                                                                                                                                                                                                                                                                                                                                                              H1H1H0εrεrεoFig. 2. 3D-view of the proposed filter containing two broadside coupleddielectric substrates.resonator can be operated at the (λ) resonance. This can bedone easily by introducing a meander line resonator, as shownin Fig. 3.                                                                            wgll1 l2l3l4l5l6l7Fig. 3. Layout of a third order transversal microstrip filter, implementingthe coupling scheme of Fig. 1(b).In Fig. 3 we show the basic topology used for the imple-mentation of the filter in microstrip technology. The dottedresonator is broadside coupled to the input/output ports. Todo this, the resonator is printed in the dielectric placed on top.The planar structure shown in Fig. 2 and Fig. 3 can beanalyzed with the commercial software Agilent-ADS c©. InFig. 4 we present the results obtained from the analysis ofthe (MA) coupling matrix, together with the results obtainedfrom the analysis of the multilayered structure, after the designprocess. The dimensions obtained for this topology are shownin Table I. We can observe very good agreement between thetwo responses.TABLE IDIMENSIONS OF THE DESIGNED DUAL-BANDPASS FILTER SHOWN INFIG. 3.Dimension Value (in mm.) Dimension Value (in mm.)w 1.85 l1 20.8g 0.3 l2 1.5H0 5.0 l3 13.2H1 1.27 l4 1.5εr 6.15 l5 26.0Lin = Lout 30.0 l6 51l 5.0 l7 547761.1 1.2 1.3 1.4 1.5 1.6 1.7 1.8−70−60−50−40−30−20−10010Frequency (GHz)Mag. (dB)S21 Full−waveS11 Full−waveS21 MatrixS11 MatrixFig. 4. Fullwave simulations and coupling matrix response for the dualbandpass filter shown in Fig. 3.B. Transversal filter with pseudo-elliptic frequency responseIt is interesting to observe that the same configuration asintroduced in the previous subsection can also be used tosynthesize a single-band transfer function with quasi-ellipticresponse. Two transmission zeros can be placed on both sidesof the passband for maximum selectivity, by changing the signof one of the couplings.To illustrate the concept, we now design a single-bandpassfilter with quasi-elliptic response and a bandwidth of 110MHz. The two transmission zeros are located at 1348 MHz and1504 MHz, and the ripple level is 15 dB. Since the topologycorresponds to a transversal filter of order three, the synthesisprocedure reported in [2] can be directly used to obtain thecoupling matrix:MB =⎛⎜⎜⎜⎜⎝0 0.352 0.421 −0.756 00.352 1.190 0 0 0.3520.421 0 −1.099 0 0.421−0.756 0 0 0.130 0.7600 0.352 0.421 0.756 0⎞⎟⎟⎟⎟⎠(2)We can see that the coupling matrix is very similar to the(MA) matrix previously computed. Only one of the resonatorschanges the sign of the coupling to the input port (comparesecond column of matrices MB and MA). Also, the thirdresonator has to be adjusted (in coupling and length) to recoverthe desired response (see fourth column of the matrix).Note that the additional sign change can be implementedin one of the (λ/2) resonators, by placing a short-circuit inthe central point of the line. This short-circuit will force aneven symmetry of the electric field in the resonator, so the signwill not change anymore from input to the output port. In thisway we can easily implement the same sign for the couplingelements MS1 = ML1 = 0.352. In Fig. 5 it is shown the newlayout for the proposed microstrip configuration.Again, the dotted resonator is printed on the layer on top.The only modification of this structure with respect to thelayout presented in Fig. 3, is the via-hole introduced at theVia−hole                                                               H0H1H1wwwwgglin loutl1l3l4l5l6l7l8εrεrFig. 5. Layout of the third order transversal filter with quasi-elliptic response.The sign of one coupling is changed by introducing a short-circuit in thecentral point of one resonator.central point of this resonator, used to implement the requiredshort-circuit. Also, the other (λ/2) resonator has been adjustedby slightly shortening its length. It is important to point outthat the meander resonator has not been changed, as indicatedby the third column of matrix (MB), which is identical to thematrix (MA). The final dimensions for this filter are given inTable II.TABLE IIDIMENSIONS OF THE DESIGNED PSEUDOELLIPTIC FILTER SHOWN INFIG. 5.Dimension Value (in mm.) Dimension Value (in mm.)w 1.85 l2 1.5g 0.3 l3 13.2H0 2.0 l4 1.5εr 6.15 l5 26.0Lin = Lout 30.0 l6 49.6l 5.0 l7 45.4l1 20.8 l8 1.5In Fig. 6 we present the results obtained from the analysisof the (MB) coupling matrix (2), together with full wavesimulations of the structure shown in Fig. 5.1.15 1.2 1.25 1.3 1.35 1.4 1.45 1.5 1.55 1.6 1.65−60−50−40−30−20−100Frequency (GHz)Mag. (dB)Resonator 1Resonator 2Resonator 3S21 Full−waveS11 Full−waveS21 MatrixS11 MatrixFig. 6. Fullwave simulation and coupling matrix response of the pseudo-elliptic transversal filter shown in Fig. 5.We can observe good agreement between both responses.Only at the beginning of the passband, the full-wave simula-tion shows a slight deviation. It is believed that this is due toa spurious radiation of the via-hole at these frequencies.777III. RESULTS AND EXPERIMENTAL IMPLEMENTATIONFor a practical validation of the filter designs presented inthis paper, several prototypes have been manufactured andtested in microstrip technology. In this contribution we willreport the results obtained for the dual-bandpass filter. Thebasic layout of the fabricated filter is shown in Fig. 3. Thesubstrate selected for manufacturing is an RT-Duroid/6006,with a relative permittivity εr = 6.15, and thickness H1 =1.27 mm.1.2 1.3 1.4 1.5 1.6 1.7−45−40−35−30−25−20−15−10−50Frequency [GHz]Scattering ParamtersS21 measuredS11 measuredS21 simulatedS11 simulatedFig. 7. Comparison between simulated results and measured results for thefilter shown in Fig. 3.In Fig. 7 we present the results obtained for the filter shownin Fig. 3, when losses are included in the simulations. Both,dielectric losses (tan(δ) = 0.002) and losses in the printedmetalizations (σ = 4 · 107Ω−1/m), are considered. Measuredresults are also included in Fig. 7 for comparison. We canobserve good agreement between measured and predictedresponses, for both passbands of the filter. The first passbandshows an insertion loss of 1.3 dB and a return loss of 15 dB,while the upper passband has a maximum insertion loss of 1dB and return loss of 17 dB.The manufactured prototype is shown in Fig. 8. The basicstructure consists of two different dielectric substrates sep-arated by a layer of air, providing the required broadsidecoupling to the input/output ports. The substrates are properlyaligned to allow for a precise adjustment of this broadsidecoupling (H0 separation in Fig. 5). The lower dielectric sub-strate of the filter (coupling matrix MA, (1)) contains theinput/output ports and two of the resonators (one of typehalf wavelength and one of type meander line), and it canbe seen at the top of the photo. The meander line resonatorimplements the third column of the coupling matrix MA, whilethe half wavelength resonator implements the fourth column.The bottom of the photo shows the upper dielectric substrate,containing the third resonator. This substrate contains the basicprinted line resonator, also of type half wavelength, needed toimplement the second column of the coupling matrix MA (1).Fig. 8. Photo of the manufactured prototype.IV. CONCLUSIONSTwo third-order novel microstrip bandpass filters havebeen designed. The coupling schemes chosen correspond totransversal third-order filters, exhibiting two transmission ze-ros, and have been implemented for the first time in microstriptechnology. It is shown that several frequency responses canbe implemented, such as dual-band or pseudo-elliptic. Severalstrategies to adjust the signs of the couplings have beenproposed. This includes the use of printed resonators ofdifferent lengths, and the introduction of short-circuits whenappropriate. Measured results are presented for one prototype.The obtained measurements show good agreement with respectto theoretical predictions, showing the validity of the proposedstructures.V. ACKNOWLEDGMENTSThe authors thank Ministerio de Educacion y Ciencia ofSpain, which has supported this work under Grant TEC2007-67630-C03/TCM, and to Miss Attieh Shahvarpour of theÉcole Polytechnique de Montréal for her collaboration inmanufacturing and testing the prototype.REFERENCES[1] G. Macchiarella and S. Tamiazzo, “Design techniques for dual-passbandfilters,” IEEE Transactions on Microwave Theory and Techniques, vol. 53,no. 11, pp. 3265–3271, Nobember 2005.[2] R. J. Cameron, “Advanced coupling matrix synthesis techniques formicrowave filters,” IEEE Transactions on Microwave Theory and Tech-niques, vol. 51, no. 1, pp. 1–10, January 2003.[3] D. C. Rebenaque, F. Q. Pereira, J. P. Garcia, A. A. Melcon, andM. Guglielmi, “Two compact configurations for implementing transmis-sion zeros in microstrip filters,” IEEE Microwave and Wireless Compo-nents Letters, vol. 14, no. 10, pp. 475–477, October 2004.[4] J. T. M. Garay, “Synthesis of physically asymmetrical n-trisection filterswith transmission zeros at n differente real frequencies,” ElectronicsLetters, vol. 35, pp. 226–227, February 1999.[5] K. R. M. Moktaari, J. Bornemann and S. Amari, “Coupling-matrix designof dual and triple passband filters,” IEEE Transactions on MicrowaveTheory and Techniques, vol. 54, no. 11, pp. 3940–3946, November 2006.[6] D. Swanson and G. Macchiarella, “Microwave filter design by synthesisand optimization,” IEEE Microwave Magazine, pp. 57–69, April 2007.778

Novel implementations for microstrip resonator filters in transversal topology

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Novel implementations for microstrip resonator filters in transversal topology

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