D'Hulst, R

Driesen, J

Mitcheson, P D

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CMOS Buck-Boost Power processing circuitry forpowerMEMS generatorsR. D’hulst1∗, P. D. Mitcheson2, J. Driesen11K.U.Leuven - ESAT/ELECTAKasteelpark Arenberg 10, B-3001 Heverlee (Belgium)2Department of Electrical and Electronic EngineeringImperial College London, Exhibition Road, South Kensington, London, SW7 2AZAbstractThis contribution describes a power extraction circuit designed to interface to a vibrational energy harvester.Vibrational energy harvesters produce an AC voltage and therefore a power processing circuit is needed to convert theAC voltage into a DC voltage of an appropriate level for supplying a low power load. The power conversion circuitrymust also present the correct electrical impedance to the generator to maximize its power output. The buck-boosttopology is chosen for the extraction circuit, its operating point in to discontinuous mode. A reservoir capacitor ispresent to store the extracted energy. An implementation of the design was simulated using a standard 80 V CMOSprocess, and efficiencies of over 60% are demonstrated. With suitable control, the input impedance of the circuit isindeed resistive as well as variable and thus well suited as a load for powerMEMS generators.Keywords: Energy Harvesting, Power Processing circuit, Buck-Boost converter, integrated CMOS1 - INTRODUCTIONThe current trends towards miniaturization, portabilityand ambient intelligent networks require new means toproduce energy in a decentralized fashion. ‘Energyharvesters’ offer a solution to this problem. These aredevices able to convert ambient energy in the environmentinto useful energy to power, for example, an autonomoussensor. New MEMS techniques are used for fabrication ofthese micro power generators [1, 2]. The output power ofsuch devices is very limited, ranging from milliwatts downto only a few microwatts. In this paper a power extractioncircuit designed to interface to the electrostatic vibrationalenergy harvester described in [2] is presented, although thecircuit may also be applicable to the other vibration-drivengenerator types (piezoelectric and electromagnetic) [3, 4].All three types can be modeled as mass-spring-dampersystems. When there is relative motion between themass and the generator housing work is done against thedamper and this represents the energy generated.Vibrational energy harvesters produce an AC voltage andtherefore a power processing circuit is needed to convertthe AC voltage into a DC voltage of an appropriatelevel for supplying a low power load. The power density∗Contact author: Tel: +32 16 32 10 20, Fax: +32 16 32 19 85,email: reinhilde.dhulst@esat.kuleuven.beof an energy harvester (at resonance) increases withthe amplitude of the relative motion between the proofmass and generator housing and therefore the powerconversion circuitry must also present the correct electricalimpedance to the generator to maximize the travel of theproof mass [5].The input specifications for the power processing circuitare taken as a sinusoidal input voltage with a frequencyof 1000 Hz. This is typical for the harvester devicein [4] at an energy extraction rate of around 100 µW. Aproposed overall system block diagram is shown in Fig. 1.The first power processing stage must maximize energyextraction from the scavenger mechanics and store it inthe reservoir capacitor. The reservoir is present due tothe intermittent nature of many vibration sources meaningthat the operating duty cycle of the scavenger may besignificantly less than 1. The final regulation stage ensuresa high quality supply for the load.2 - BUCK-BOOST CONVERTERThe buck-boost topology [6], shown in Fig. 2, is chosen asthe extraction circuit for two main reasons: its simplicityand its ability to step-up and down the voltage. Stepup and down is important because as the mechanicaloperating conditions of the generator change, the voltageFigure 1 - Energy scavenger system level diagram.Figure 2 - Basic Buck-Boost AC-DC switch-mode powerconverter.at the input to the extraction circuit may fluctuate aboveand below the voltage on the storage capacitor. Theoperating point of the buck-boost switching converter isset to discontinuous mode, in order to reduce switchinglosses, and to keep the inductance as small as possible.This may allow integration of the inductor on chip andwill at the very least ensure that external componentsizes are minimized [7]. A high voltage is allowed on theoutput capacitor, to ensure optimal energy capture fromthe generator at all times, while energy usage by the loadis variable. The input current of the buck-boost converteris shown in Fig. 3. The on-time (Ton = DTs) of the switchis kept constant, as to assure a ‘resistive’ input impedanceas is required by the scavenger. This load impedanceneeds to be variable to be able to optimize energycollection at different harvesting conditions. The inputimpedance of the processing circuit can be easily adaptedby changing Ton of the main power transistor. Moreover,the input current drawn from the harvester is sinusoidal,this significantly increases energy yield compared to theuse of a simple diode rectifier according to [7].The peak input current can be approximated by eq. (1), ifTs  1fh , with Ts =1fsthe switching period of the switch,and fh the input vibration frequency of the scavenger. Theoutput voltage amplitude of the harvester is given by VI ,and the inductance is given by L.Figure 3 - Input current of the buck-boost converter indiscontinuous conduction mode.ipeak(t) =VI |sin(2πfht)|DLfs(1)The input current of the converter, averaged over oneswitching period is given by eq. (2).iavg(t) =VI |sin(2πfht)|D22Lfs(2)The average input power of the converter is then givenby eq. (3), this equation is used to calculate Ton, given acertain input power.Pin = 2fh∫ 12fh0V 2I |sin2(2πfht)|D22Lfs=D2V 2I4Lfs(3)The input resistance is given by eq. (4).Rin =2LfsD2(4)A high switching frequency should be used if a smallinductance is desired, to allow for a fully integrated circuit.At very high fs, Ton decreases, but Ton should at least behigher than the sum of rise and fall times of the usedswitch, to allow it to turn fully on. The fall time of thecurrent through the inductance is proportional to ipeak,and inversely proportional to the output voltage. To allowfor a minimal current fall time at high output voltages, theswitching frequency cannot be too high. An inductanceof 10 µH is used further on, a PCB-integrated inductorwith this inductance seems to be feasible [8]. A switchingfrequency of 100 kHz is chosen, this gives a theoreticalTon of about one order larger than the reported sum ofthe transistor rise and fall times, with an input voltagerange between 1 and 15 V, and an input power range of10-100 µW. Figure 4 shows how the Ton of the switchvaries with varying input power and voltage amplitude, itcan be seen that a low input voltage has an advantageouseffect on Ton. The highest achievable output voltage,allowing for an acceptable current fall time is then about20 V.Figure 4 - Ton versus Pin and VI .3 - IMPLEMENTATION3.1 - Component sizingThe design has been implemented and simulated using astandard 80 V CMOS process. High-voltage diodes andhigh-voltage transistors are available in this technology.The size of the power transistor was optimized byminimizing gate-charge losses as well as conduction andswitching losses. Because conduction times of thetransistor are very short, it is hard and even irrelevantto make a distinction between conduction and switchinglosses. As can be seen from Fig. 5, simply optimizing forconduction losses is not sufficient, gate charge losses aresignificant, so a compromise has to be made. Still, thegate-charge losses amount to almost one third of the totalpower transistor losses at the optimal transistor size. Adecrease of the gate-charge losses is obtained by purposelyunder driving the gate with a low turn-on voltage. Thethreshold voltage of the high-voltage transistors lies at0.7 V, if a gate drive voltage of 1 V is used in steadof the commonly used 3.3 V, the total loss due to thepower transistor can be reduced with 25% at the optimaltransistor size, as shown in Fig. 6. Diode losses, includingreverse recovery losses, are minimal if the largest availablediode area is used.A diode rectifier is implemented for initial rectification,chosen for its simplicity. Possibly a fully synchronousrectifier is more efficient, to be investigated in future work.Figure 5 - Conduction loss, gate-charge loss and totalpower loss during half a vibration period, versus thetransistor area.Figure 6 - Total energy loss versus transistor area, atdifferent turn-on voltages.3.2 - Synchronous rectificationTo avoid the power loss due to the forward voltage dropover the diode, a design with synchronous rectification hasbeen implemented. The same conclusions for the sizingof the synchronous rectifying transistor can be drawnconcerning gate charge losses and conduction losses of themain switch transistor, versus transistor size. Therefore,both the main switching transistor and the rectifyingtransistor are chosen to be equal in size and are driven witha gate-drive voltage of 1 V. Simulations show that the totalsynchronous rectifier losses are substantially less than thediode losses, if the same input power is drawn from thesource. Figure 7 shows the distribution of the energylosses of the different components during one scavengervibration period. ‘Eds1’ and ‘Egc1’ are the drain-sourceand the gate-charge losses of the main switch transistor,‘Edio’ are the diode losses, ‘Eds2’ and ‘Egc2’ are the lossesin the synchronous rectifier and ‘Eout’ is the fractionof the input energy going to the output capacitor. Noparasitic losses in the inductor or the output capacitorare taken into account, neither are the losses due tothe gate-drive circuit and other control units. With theuse of synchronous rectification, simulations show thatan efficiency of more than 60% can be achieved whileconverting 100 µW. Efficiency varies greatly with changingoutput voltage: if the output voltage is low, the efficiencyis low, due to the long conduction times of the diode orthe synchronous rectifier transistor. The efficiency gain ofsynchronous rectification with respect to the use of a diodeis higher, because conduction losses of the synchronousrectifier-transistor are lower, see Fig. 7. At higher outputvoltages the efficiency gain decreases, conduction timesbecome very short, explaining the smaller difference inefficiency between synchronous rectification and the useof a diode. The efficiency decreases at very high outputvoltages: a very large inductor current fall-rate combinedwith a finite diode reversing time, causes the current tobecome negative for a very short time. This reversedcurrent flows back to the generator through the parasiticdiode of the main transistor, explaining the lower energyoutput at the highest voltage in Fig. 7, this effect has tobe avoided. The size of the output capacitor is determinedby the amount of energy needed to be buffered. Thisamount depends on the requirements of the load and onthe duty-cycle of the harvester. An optimal capacitorsize can be derived according to the load and harvestercharacteristics, to ensure that the output voltage of theconverter at most times, is the voltage where the converterworks at optimal efficiency.Figure 7 - Distribution of the losses of the differentcomponents in the buck-boost converter with and withoutsynchronous rectification, at different output voltages.4 - CONCLUSIONIn this contribution a proposal is made for a powerprocessing circuit for vibrational energy harvesters. Thecircuit has to perform an AC-DC conversion as well aspresent the correct electrical impedance to the energyharvester to maximise its power output. This is achievedby a buck-boost topology, working in discontinuousconduction mode. The design has been implemented in a80 V CMOS technology, and proves to have a conversionefficiency of more than 60%. The input impedanceis resistive and easily adaptable to different operatingconditions.ACKNOWLEDGEMENTSResearch presented in this paper is financed by the IWT(Institute for Promotion and Innovation by Science andTechnology in Flanders) under the project SBO 030288“PowerMEMS”.REFERENCES[1] A. Holmes, G. Hong, K. Pullen, and K. Buffard, “Axial-flowmicroturbine with electromagnetic generator: design,CFD simulation, and prototype demonstration,” in MicroElectro Mechanical Systems, 2004. 17th IEEE InternationalConference on. (MEMS), 2004, pp. 568–571.[2] T. Sterken, P. Fiorini, K. Baert, R. Puers, andG. Borghs, “An electret-based electrostatic µ-generator,”in TRANSDUCERS, Solid-State Sensors, Actuators andMicrosystems, 12th International Conference on, 2003,vol. 2, 2003, pp. 1291–1294.[3] N. Ching, H. Wong, W. Li, P. Leong, and Z. Wen,“A laser-micromachined multi-modal resonating powertransducer for wireless sensing systems,” Sensors andActuators A, vol. 97-98, pp. 685–690, 2002.[4] A. Schmitz, T. Sterken, M. Renaud, P. Fiorini, R. Puers,and C. Van Hoof, “Piezoelectric scavengers in memstechnology: Fabrication and simulation,” in The 5thInternational Workshop on Micro and Nanotechnology forPower Generation and Energy Conversion Applications(PowerMEMS 2005), University of Tokyo, Japan,Nov.28-30 2005, pp. 61–64.[5] P.D.Mitcheson, T.C.Green, and E.M.Yeatman, “Powerprocessing circuits for mems inertial energy scavengers,”in Proc. Design, Test, Integration and Packaging ofMEMS/MOEMS Symposium, Stresa, Italy, Apr. 2006, pp.332–337.[6] N. Mohan, T. M. Undeland, and W. P. Robbins, PowerElectronics: Converters, Applications, and Design. Thirdedition. John Wiley & Sons, Inc., 2003.[7] G. K. Ottman, H. F. Hofmann, and G. A. Lesieutre,“Optimized piezoelectric energy harvesting circuit usingstep-down converter in discontinuous conduction mode,”IEEE Transactions on Power Electronics, vol. 18, pp.696–703, Mar. 2003.[8] M. Ludwig, M. Duffy, T. O’Donnell, P. McCloskey,and S. Mathuna, “PCB integrated inductors for lowpower DC/DC converter,” IEEE Transactions on PowerElectronics, vol. 18, no. 4, pp. 937–945, July 2003.

CMOS Buck-Boost Power processing circuitry for powerMEMS generators

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