The measurement theory is described and a test method to perform phase noise verification using off-the-shelf components and instruments is presented. The measurement technique described consists of a double-balanced mixer used as phase detector, followed by a low noise amplifier. An FFT spectrum analyzer is then used to view the modulation components. A simple calibration procedure is outlined that ensures accurate measurements. A block diagram of the configuration is presented as well as actual phase noise data from the 400 kW, 2.115 GHz (S-band) klystron transmitter

Bhanji, A.

Boss, P.

Hoppe, D.

English

NASA Technical Reports Server

TDA Progress Report 42-88
N87-17946
October-December 1986
Phase Noise Measurements of the 400-kW,
2.115-GHz (S-Band) Transmitter
P. Boss, D. Hoppe, and A. Bhanji
Radio Frequency and MicrowaveSubsystemsSection
This article describes the measurement theory and presents a test method to perform
phase noise verification using off-the-shelf components and instruments. The measure-
ment technique described consists of a double-balanced mixer used as phase detector,
followed by a low noise amplifier. An FFT spectrum analyzer is then used to view the
modulation components. A simple calibration procedure is outlined that ensures accurate
measurements. A block diagram of the configuration is presented as well as actual phase
noise data from the 400-kW, 2. ] 15-GHz {S-band) klystron transmitter.
I. Introduction
Various papers (Refs. 1 and 2) contain outlines of methods
to measure transmitter phase stability. This article describes
the implementation of the phase detector method, with par-
ticular emphasis on close-in phase noise measurement. Practi-
cal problems encountered during such a measurement are also
described. Simple, accurate, and easily reproducible measure-
ments are necessary so that improved transmitter designs can
evolve.
Unwanted modulation and noise in the transmitted RF
signal are caused by a variety ot sources - specifically, shot
noise in the klystron, fiuctuations in the cathode beam power,
RF drive power, and heater supply voltages, cooling water
temperature fluctuations, focus supply fluctuations, and
mechanically induced vibrations from cooling fans and water
pumps. These pertubations will in most cases produce both
angle and amplitude modulation. However, in klystron ampli-
hers, near the carrier, angle modulauon _pnase nmse) pre-
dominates.
The most significant source of phase error is generally the
cathode voltage applied to the klystron. As the cathode volt-
age increases and decreases, the corresponding increase and
decrease in electron beam velocity create a decrease and
increase in the apparent electrical length of the tube. This
results in a higher quantity of electrons being emitted, which
in turn produces an increase in amplitude of the RF output.
Generally, as stated earlier, any amplitude disturbance is an
order of magnitude less significant than the phase change and
can therefore be ignored.
Another factor that introduces apparent change in phase
length of the linear beam tube is the focusing magnet. As the
electron beam enters the magnetic field, any component of
the magnetic field which is perpendicular to electron velocity
creates a transverse velocity component of the electrons.
Since the energy of the electrons is conserved, any transverse
velocity component that is added will detract from the axial
velocity component, thus causing an apparent increase in the
electrical length of the tube.
17
https://ntrs.nasa.gov/search.jsp?R=19870008513 2020-03-20T12:40:08+00:00Z
TheRF driveamplitudevariationisanotherfactorwhich
inducesphasedelayin theklystronamplifier;it isdependent
onthetubegain(numberof cavities),howclosethetubeis
runto saturation,andtheoperatingfrequencyof thetube.
Sincevariationin drivepoweresultsin extractingmoreor
lessenergyfromtheelectronbeam,therearecorresponding
changesin thebeamvelocitythatinducephasechanges.
Variationsin heatervoltages,whichwill changecathode
temperature,will alsohaveaneffecton thephaserror.
Similarly,variationsin theklystronbodytemperaturewill
affecthelengthof thecircuitandthuswill alsoaffecthe
phaserror.
Finally,onemustconsidersucheffectsasvibrationin the
tube from mechanicalcavitytunermechanismsthat can
physicallymovedueto vibrationorantennamovement.In
addition,physicalvibrationof coaxialcables,couplers,and
similarcomponentsshouldbeconsidered.
II. Noise Spectrum
A frequency domain representation of a CW carrier with
noise sidebands is shown in Fig. 1. The vertical scale repre-
sents the ratio of single sideband (SSB) power in a 1-Hz band-
width, at a frequency fm away from the carrier, to the total
signal power. This quantity is given the symbol £(f), with units
dBc/Hz (dB below carrier in a 1-Hz bandwidth). Noise power
varies directly with measured bandwidth. Thus, noise mea-
sured in a bandwidth other than 1 Hz may be normalized to a
1-Hz bandwidth by subtracting the correction factor given in
Eq. (1).
BW
Correction Factor = 101Oglo _ (1)
where BW is the measured bandwidth in Hz. The power of
discrete deterministic sidebands does not scale with measure-
ment bandwidth; it remains constant.
As stated earlier, the phase noise in a klystron dominates
am noise at frequencies close to the carrier, and to measure
this noise the phase detector method was used.
III. Phase Detector Method
The theory behind this measurement is fairly straightfor-
ward. The basic components are a double balanced mixer used
as a phase detector, a low noise amplifier, and a low frequency
spectrum analyzer. The relationship between the RMS varia-
tion in phase between the L0 and RF ports of the mixer and
the ratio of the power in a single sideband to the power in the
carrier (PssB) is given by:
where ¢ is the difference between the RF and LO ports. Also,
the relationship between the IF voltage (VIF) of the mixer and
q_is given by:
VIF = K cos Qr/2 + ¢ rms)
VIF _--K 4) rms
(2)
where K is the mixer sensitivity and
Vm = A VIF
where A is the LNA gain and
PssB -
2(AK) 2
This is valid only for phase deviations of less than 0.1 radians.
Thus, by measurement of Vrn and knowledge of the charac-
teristics of the circuit (A and K), the noise spectrum can be
determined.
IV. Hardware
A complete block diagram of the measurement system is
shown in Fig. 2. Several elements must be added to the simple
circuit described earlier to allow an accurate measurement.
Circulators to provide a matched impedence, attenuators to
control the power level, and couplers to monitor the power
level should be placed at both the LO and RF ports of the
mixer. A phase shifter to control the offset phase should be
placed in the RF side of the mixer circuit. A bias tee should be
placed at the IF port of the mixer to provide a non-reactive
termination. It was also found necessary to place a 1 kohm
resistor from the output of the bias tee to ground to increase
the low frequency sensitivity of the mixer. In addition, a DC
blocking capacitor was needed to prevent any DC offset from
the mixer from forcing the high gain LNA to saturate. One
should also consider the noise contribution of the test system
elements. When selecting elements such as the driving source
(i.e., the LO signal) and the low noise amplifier, care should
be taken to choose generators and amplifiers that will not
contribute significant noise compared to what is being
measured. An lip 3561 A dynamic signal analyzer was used as
the low frequency spectrum analyzer.
18
V. Calibration
A simple method of calibration was chosen so that it was
not necessary to determine all the parameters of the measure-
ment system. The system was calibrated by injecting a known
phase-modulated signal and measuring the results. This is per-
formed for all modulation frequencies of interest in order to
form a calibration curve for the circuit at the frequencies of
interest. For accurate results, it is important that the signal
modulation be fairly pure phase modulation of a known PSSB
level. In calibrating the circuit, a Pssa ratio of -30 dB was
used with higher order sidebands at -50 dBc or lower. This
level is set with an RF spectrum analyzer and modulating at a
frequency that can be seen. The modulation frequency was
then stepped from 100 kHz down to 0.01 Hz and the sideband
observed with the low frequency FFT analyzer. For system
calibration, the analyzer was used in the narrow-band mode to
read the actual values of the modulation sidebands.
It was found that the major contributor to any fre-
quency roll-off is the high-pass filter formed by the DC
blocking capacitor and the combination of the corre.
sponding circuit element resistances. This can be fairly well
modeled by a simple RC filter whose corner frequency is
about 2 Hz. The response also falls off a little at fre-
quencies higher than 60 Hz, but since the main frequencies
of interest for this measurement are below 20 kHz, this
effect can be neglected. This response will then be used to
correct the measured data.
Vl. Measurement
When setting up the system to make a measurement, care
must be taken to match the calibration conditions as closely
as possible. The power levels seen by the mixer at its RF and
LO ports are especially critical. Values of PLO -- +8 dBm and
PRF = -2 dBm were chosen for this measurement. These
values can be changed but they must be repeated between
calibration and an actual measurement.
The mixer must also be in quadrature for maximum sensi-
tivity to phase variations and minimum sensitivity to ampli-
tude variations. This is determined by monitoring the mixer
IF voltage before the 13(2 blocking capacitor. Because it can
take hours to make a measurement at low frequency spans,
it is desirable to know about any long-term phase drifts. An
attempt was made to continuously monitor this point to check
for very long-term phase drifts, but this led to problems because
the spectrum analyzer was able to detect noise from the instru-
ment used to monitor this point. Quadrature was then checked
before and after the measurement in order to detect unaccept-
able amounts of drift.
The measurement was,made at seven different frequency
spans. The data for each of these spans was then combined to
form one plot. These frequency ranges were:
0-100 kHz
0-10 kHz
0-1 kHz
0-100 Hz
0-10 Hz
0-1 Hz
0-0.1 Hz
The measurement data for each frequency span was read
from the spectrum analyzer into the computer, where any
gain or attenuation effects imposed by the system and deter-
mined in the calibration were weighted out. This provided
results in the form of dBc/Hz. Each frequency span consists
of 400 data points, which provides single-point resolution of
0.25 mHz on the 0.1 Hz frequency span.
When making this measurement, the spectrum analyzer will
automatically range to accommodate the largest noise signal
seen at its input. If there are relatively high-level discrete
lines, the analyzer will choose a range to accommodate these
signals. Since the dynamic range of the analyzer is 80 dB,
low-level random noise cannot be detected when these discrete
lines are present. Although the noise per Hz typically rises as
one approaches the carrier, this effect is overwhelmed by the
low frequency roll-off of the measurement system, and by the
decreasing measurement bandwidth.
To avoid the above problem, which was caused by cathode
voltage power supply lines and other discrete lines, a low-pass
filter was used for spans less than 10 Hz. The filter had a cor-
ner frequency of about 2 Hz, and its effect was corrected for
in the data measured at 1 Hz or less.
Since the spectrum analyzer was capable of storing the time
data as well as the frequency data, the measurement was done
using this "time buffer" capability. The time data can then be
searched for any one-time event such as a step in phase.
Events of this type can cause very deceptive results in the
measured noise level. This method, although helpful, requires
the storing and handling of large amounts of data.
VII. Measured Data
Figure 3 shows the results of the phase noise measurement
made on the class C driver amplifier. It should be noted that
this follows a theoretical 1If noise characteristic. It is also
19
apparent that the driver noise is approximately 20 dB lower
than the noise measured with the klystron in the circuit. This
plot is shown in Fig. 4 The plot of the transmitter phase
noise reveals several discrete spurious lines. Most of these
components are traceable to the cathode voltage noise charac-
teristics, which are shown in Fig. 5. The dominant lines are
at 400 Hz due to the generator frequency, at 800 Hz (the
2nd harmonic), and at 2400 Hz (a product of the 400-Hz
rectification). Also present is an unexpected line at 16 Hz,
which was found to be a product of the cathode voltage sup-
ply's control feedback loop.
When the cathode supply voltage noise spectrum and the
phase noise spectrum are compared, it becomes apparent that
the two are very similar. This implies that the supply voltage
is the major contributor to the noise. This is expected because
the cathode voltage has the highest phase pushing factor of
the affecting parameters. It is also seen that the noise of the
system approaches the noise of the driver amplifier at high
frequencies.
VIII. Conclusions
This article has presented a technique for measuring the
phase noise of RF power amplifiers that is easily reproducible
using off-the-shelf instruments and common electronic compo-
nents. Measurement results are readily interpreted and unam-
biguous. This method will assist in the design evolution process
(improved power supply regulation and long-term stability,
improved temperature control of the cooling water, etc.)
by correlating design changes with overall performance
measurements.
Acknowledgments
The authors acknowledge the assistance of Marlyn Gregg and John Daeges in perform-
ing the measurements and interpreting the data.
References
1. Nocella, P. (Sperry Corporation), "Practical Purity Measurement of Pulse Doppler
Radar Transmitter Amplifiers," IEEE 17th Pulse Power Modulator Symposium, 1986,
pp. 32-38.
2. "Spectrum Analysis Noise Measurements," Hewlett-Packard Application Note 150-4,
Palo Alto, California, April 1974.
2O
dBclHz
-40
-50
"/-60
-70
-80
S -90
DISCRETE
SIDEBANDS
POWER SUPPLY
RELATED OR
MECHANICALLY
I I I L I I
101 102 103 104 105 106
FREQUENCY OFFSET, Hz
Fig. 1. The CW phase noise
HP 8349B
AMPLIFIER
OSCI LLATOR
PLO--J
LNA
SPECTRUM
ANALYZER
I--
I
'I
t
_ h,_---"rr/2
•20 dB
VLO
LO
DSN 2582/TXR-011 TRANSMITTER
DRIVE
CIRCUIT
BRIDGE
IF
VKS 8276
IV
-20 dB
"--I
I
I
I
._!
10dB
h
L PRF
I,VRF
Fig. 2. Phase noise measurement system
RF
21
0.00 L ,
-30.00
' ''1 ' ''I ' ''1 ' ''1 .... I ' ''1 ' ''{
-120.00
-150.00
10-3 10-2 10-1 100 101 102 103 104 105
FREQUENCY
Fig. 3. Class C driver amplifier phase noise
0.00 i ' ' ''1 ] ' F'I ' ' '_1 ' ' ''l ' ' ''1 r ' ''1 ' ' 'rl ' , b,
-30.00 _ S 16 Hz
_" F 400 Hz
-60.00
_" / _800 Hz
-90.00 2400 Hz
-150.001_ ; , ,,t J , *,1 , , ,,I , ,_,1 , ,
10-3 i0-2 i0-1 100 101 102 103 104 105
FREQUENCY
Fig. 4. The 2.115-GHz, 400-kW transmitter phase noise
22
86.0
-64.0
10-3 10-2 10-1 100 101 102 103
FREQUENCY
Fig. 5. Cathode voltagesupply noise spectrum
104 105
23


Phase noise measurements of the 400-kW, 2.115-GHz (S-band) transmitter

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