An advanced coherent demodulation technique is described for land mobile satellite (LMS) communications. The proposed technique features a combined narrow/wind band dual open loop carrier phase estimator, which is effectively able to compensate the fast carrier phase fluctuation by fading with sacrificing a phase slip rate. Also included is the realization of quick carrier and clock reacquisition after shadowing by taking open loop structure. Its bit error rate (BER) performance is superior to that of existing detection schemes, showing a BER of 1 x 10(exp -2) at 6.3 dB E sub b/N sub o over the Rician channel with 10 dB C/M and 200 Hz (1/16 modulation rate) fading pitch f sub d for QPSK. The proposed scheme consists of a fast response carrier recovery and a quick bit timing recovery with an interpolation. An experimental terminal model was developed to evaluate its performance at fading conditions. The results are quite satisfactory, giving prospects for future LMS applications

Tomita, Hideho

Yoshida, Shousei

English

NASA Technical Reports Server

N92-24173 !
A New Coherent Demodulation
Technique for Land-Mobile Satellite
Communications
Shousei Yoshida, Hideho Tomita
C&C Systems Research Labs., NEC Corp.
1-1, Miyazaki 4-chome, Miyamae-ku, Kawasaki-city, 213, JAPAN
PHONE +81-44-856-2255 FAX +81-44-856-2061
ABSTRACT
This paper describes an advanced coherent
demodulation technique for land-mobile satellite
(LMS) communications. The proposed technique
features a combined narrow/wide band dual open
loop carrier phase estimator, which is effectively
able to compensate the fast carrier phase fluctu-
ation by fading without sacrificing a phase slip
rate. Also included is the realization of quick
carrier and clock re-acquisition after shadowing
by taking open loop structure. Its bit error rate
(BER) performance is superior to that of exist-
ing detection schemes, showing a BER of 1x 10-2
at 6.3 dB Eb/No over the Rician channel with
10 dB C/M and 200 Hz (1/16 modulation rate)
fading pitch fa for QPSK. The proposed scheme
consists of a fast response carrier recovery and a
quick bit timing recovery with an interporation.
An experimental terminal modem has been devel-
oped to evaluate its performance at fading con-
ditions. The results are quite satisfactory, giving
prospects for future LMS applications.
1. INTRODUCTION
Satellite based land-mobile communications
have received increasing attention over recent few
years because of its extended service to the rural
area, where terrestrial mobile systems cannot be
incorporated from an economical viewpoint. In
LMS systems, the use of power and spectrally ef-
ficient transmission scheme is intensely requested
to overcome the large space loss, and to conform
the limited available frequency band. To meet
such requirements, coherent detection combined
with forward error correction (FEC) is in corn-
mon practice. However, in the LMS environment,
there exist two serious degrading factors for co-
herent detection. One is multipath fading, which
ranges to 100- 200 Hz at maximum. Since a 4800
bps data rate is sufficient for a current digital
voice transmission technique, good phase coher-
ence cannot be counted on over long bit period,
thus it results in serious BER degradation. The
other is frequent received signal interruption by
obstacles (i.e. shadowing). A phase lock loop
(PLL) used conventionally in carrier and clock
synchronization is very vulnerable to such envi-
ronments, and often loses its track, taking a lot of
time for the recovery. Although the phase track-
ing speed and/or recovery time can be improved
by making loop bandwidth wide, it induces fre-
quent phase slips as a penalty. It is quite difficult
to find a good compromise between these two con-
flicting factors. Authors propose a new coherent
demodulation technique, which is almost able to
remove PLL's deficiency.
2. NEWLY PROPOSED COHERENT
DEMODULATION SCHEME
2.1 Carrier Recovery by Dual Open Loop
Phase Estimator
In order to resolve aforementioned problems,
authors propose a new carrier phase estimator,
which consists of two different bandwidth esti-
mators to compensate the fast phase fluctuation
by fading without increasing phase slips. An
LMS channel is modeled by the mixture of a
strong direct-path signal component and a scat-
tered weak multipath signal component, that is
Rician. Moreover the received signal is contami-
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International Mobile Satellite Conference, Ottawa, t990
https://ntrs.nasa.gov/search.jsp?R=19920014930 2020-03-17T12:27:44+00:00Z
nated with thermal noise, as depicted in Fig. 1.
Q THERMAL
NOISE
RECEIVED
Fig. 1 Received signal vector over LMS channel
Figure 2 shows a block diagram of the pro-
posed carrier recovery by a dual open loop phase
estimator. This block diagram features open loop
architecture used throughout to make free from
acquisition problems. Basic open loop phase es-
timator's performance is previously analyzed in
[1],[2]. A long term frequency offset is assumed
to be corrected by an automatic frequency control
(AFC) circuit. As a first operation, phase modu-
lation is removed by an Mth-law operator, after
that, the signal is routed both to narrow and wide
band phase estimators. The narrow band estima-
tor's bandwidth is designed around some tens Hz
so as not to induce phase slips by thermal noise,
and the estimator's filter is constructed by a finite
impulse response (FIR) filter because of its well-
defined delay time. As a result, almost all the
multipath components as well as thermal noise
are smoothed out, because the filter's bandwidth
is much narrower than assumed maximum fading
pitch 200Hz, and only a direct-path component is
derived. A narrow band estimator's output 0,_(n),
n as an integer, is expressed by
n+(Nl-1)/2
O.(n)= arg[ C.(,, - k)SM(k)]
k=n-(Nl-l)/2
n+(Nfil)/2= SM(k)]
arg[ N----_k=,,-(m-1)/2
(c.(i) = 1/N1 for vi), (1)
where SM(n) is a modulation removed signal, and
C,(i) and N1 are tap coefficients and tap number
of the narrow band estimator respectively.
The fast phase fluctuation due to the multi-
path is estimated by the wide band estimator.
The wide band estimator takes the same con-
figuration as the narrow band estimator except
its bandwidth. A wide band estimator's output
O_(n) is also expressed by
n+(N2-1)/2
= arg[ C (n - k)SM(k)]
k=n-(N2-1)/2
1 n+(N2-1)/e
= arg[_ _ SM(k)]
2 k=n-(N2-1)/2
(C,_(i)=I/N2 for vi), (2)
where C,_(i) and N2 are tap coefficients and tap
number of the wide band estimator respectively.
An impulse response of the respective estimators
is designed rectangular to reduce a computational
load.
A carrier phase is estimated by combining the
narrow and wide band estimator's outputs so that
the wide band estimator's output cannot induce
carrier phase slips. First, the narrow band esti-
mator's output is subtracted from the wide band
estimator's output, and the result is divided by
modulation phase M, thus a fast phase com-
ponent scattered by the multipath is obtained
within +Tr/M radian, for example -t-7r/4 radian
for QPSK. The narrow band estimator's output
is also divided by M, and then its phase range is
extended to +_" radian by quadrant correction. A
recovered carrier phase is obtained by adding the
estimated fast phase component to the quadrant
corrected slow changing phase. Thus, a recovered
carrier phase O_(n) is calculated by using the fol-
lowing equation,
27r.
O,(n) = mod{10,(n) + --_z(n)
+ MmOd[0 (n) - (3)
where i(n) denotes mod{i(n- 1)- sign[0,(n)-
On(n- 1),I],M} for I O,_(n)-On(n- 1)I> r/M
and i(n - 1) otherwise. In this carrier recovery
scheme, a carrier phase slip rate depends only
on the narrow band estimator's performance. Fi-
nally, the recovered phase is converted to a com-
plex form, and a received signal is coherently de-
tected by the complex carrier. Delay time T1 and
International Mobile Satellite Conference, Ottawa, 1990
623
INPUT
PHASE COMBINER
I ..................................
.[_FTR--I ._ On I I--'T1 [ QUADRANT I ,,_
I1  u .rlcoRRECTION 
II( RR°m .. I I Ir .....Jj ,
(WIDE) L ......................... j'
I
I
I
.I 0r
)====_OUTPUT
Fig. 2 Proposed carrier recovery block diagram
T2 in Fig. 2 are determined using baud interval
Tb as follows,
T1- N1- N2Tb' T2- N'- I Tb" (4)
2 2
The BER performance for several detection
schemes including the proposed scheme is evalu-
ated by computer simulation over the Rician fad-
ing channel with carrier-to-multipath ratio (Ri-
clan factor) C/M of 10 dB and fading pitch fd
of 1/16 modulation rate for QPSK. These results
are shown in Fig. 3.
3x10 -1,
• PROPOSED
DUAL OPEN LOOP P.E.
(N,=101, N==7)
10-' O SINGLE OPEN LOOP P.E.
(Nt =101)
A DIFFERENTIAL DETECTION
--- AWGN
FADING
C/M=10dB,
d =MODULATION
Fig. 3 BER characteristic by simulation
The proposed scheme shows the Eb/No improve-
ment of around 1-2 dB compared with conven-
tional coherent detection using a single open loop
phase estimator with narrow bandwidth, and the
Eb/No improvement of around 2 dB over non-
coherent differential detection, at the operating
1 × 10-2-5 x 10-3 BER point derived from the high
coding rate FEC.
Although static BER performance of the pro-
posed scheme is degraded by the wide band es-
timator, its amount is relatively small from the
conventional scheme, and in an LMS environment
its degradation becomes negligible compared with
that by fading.
2.2 Quasi-Open Loop Bit Timing Recovery
In bit synchronization, quick recovery from
frequent shadowing is another requested charac-
teristic in an LMS environment. To meet such
a requirement, authors propose a new bit timing
recovery scheme, which is able to make free from
acquisition problems by taking quasi-open loop
structure. The block diagram is shown in Fig.
4. After root Nyquist filtering, a signal envelope
is detected, and then phase correlation between
the signal envelope and a complex sinusoidal local
reference is calculated. A RaM table is referred
to generate the sinusoidal reference at the input
sample timing. Ts as sample interval, phase cor-
relation Sc(nTs) is expressed by
Sc(nT_) = Ve(nT_)exp(-j27cn/N), (5)
where V,(nT_) is a signal envelope and N is a
sample number per baud interval. The correlated
output is fed to a low-pass filter to smooth out the
phase jitter, and tan -1 operation of the filter's
output is performed. The result represents a bit
timing error against the local reference. In a case
when a slight frequency difference exists between
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International Mobile Satellite Conference, Ottawa, 1990
a modulation clock and the local reference, the
slow bit timing change is observed. To track the
bit timing continuously, a wide band first-order
PLL is introduced. A 27r/N radian phase step size
is sufficient for controlling a digital VCO, because
its request is to specify the nearest sample to real
bit timing from the input N samples. Hence, the
PLL's loop gain K is represented as
K=I/N. (6)
The loop gain K is designed large enough that the
PLL's bandwidth is sufficiently wide compared
with the open loop phase estimator's bandwidth
(low-pass filter's bandwidth). Thus, the whole
response is solely determined by the low-pass ill-
ter's characteristics, showing this loop virtually
open.
_,(PER N SAMPLESINPUT
BAuo _ OTPUT
(Oo_2_/N (rad / Ts)
Fig. 4 Bit timing recovery block diagram
A real sampled value at bit timing is ob-
tained by an interporation. The second-order's
Lagrange's formula is selected as the interpola-
tion scheme to facilitate the calculation. An in-
terpolated signal S,(r) is calculated by using the
following equation,
r_-r s r2+r _
Stir) : 7 (-T,) - (r2-1)S(0) + _S(T,)
= CI(r)S(-T,) + Co(r)S(O) + C_,(r)S(T,)
= _ C__(r)S(nT,)(I r I< ), (V)
TI-_--- 1
where S(0) is the nearest sample to real bit tim-
ing, S(-T,) and S(T,) are its adjacent samples,
and r = TilT, is normalized interporation tim-
ing. This calculation can be implemented by an
FIR filter with time variant three tap coefficients
Ci(r). These coefficients are written in a RUM
table in advance.
3. MODEM IMPLEMENTATION
An experimental terminal modem is devel-
oped to evaluate our proposed scheme. From a
viewpoint of the spectral efficiency, 5 kHz chan-
nel spacing is strongly requested in LMS systems.
Therefore, the modulation rate is determined to
be 3.2 kbaud for QPSK modulation using band-
limit filters with 0.5 roll-off factor. Thus, it en-
ables redundancy for FEC with 3/4 coding rate.
Table 1 shows the modem specifications sum-
mary.
Items Specifications
Modulation scheme 3.2 kbaud QPSK
Band- limit filter Raised cosine roll- off
(roll - off factor=0.5)
Operation mode SCPC continuous mode
5 kHzChannel spacing
Carrier deviation
A / D sampling rate
A / D quantization
Sample number N
FEC
Voice codec
Less than + 3 kHz
606 667 kHz (455 kHzlF x 4 / 3)
8 bits
6 samples per baud interval
K=7, R=3/4, 6 / 7
Punctured convolutional coding
with Vtterbi decoding
4.8 kbps multi- pulse coding
Table 1 Modem specifications summary
Figure 5 shows the demodulator block dia-
gram. A 455 kHz intermediate frequency (IF) sig-
nal is fed to a single A/D converter, and directly
sampled by 4/3x IF clock. Quadrature detection
is performed all by digital circuit using these sam-
ples, and the digitized orthogonal signal is appro-
priately decimated. After that, demodulation is
performed by three NEC's #PD77230 digital sig-
nal processors (DSP) except Viterbi decoding us-
ing a dedicated LSI chip, and the following tasks
are assigned for each processor.
1) Root Nyquist filtering (FIR filter)
2) Bit timing recovery, AFC and carrier recovery
3) Frame synchronization, de-interleaving, Vi-
terbi decoder interface and Voice decoder inter-
face
Since an open loop phase estimator is very
sensitive to a carrier frequency offset, an AFC
circuit [3] is necessitated to eliminate this offset
before the carrier phase estimation. The AFC cir-
cuit is composed of slow and fast tracking parts.
The slow tracking AFC part is requested so as
to calibrate a local oscillator at all times stably,
and the fast tracking AFC part is for the purpose
International Mobile Satellite Conference, Oflawa, 1990
625
455 kHz _ ] DIGITAL ] _ :[INTERPORATOR L,_ -- =t_
IF I I I DETECTORI IF_LTea I
l BIT CARRIER
(TOvco)
I"I FRAME LI DE I _1 VOICE14.8 kbps
SYNCHRONIZERI I INTE RLE-AME R _ INDEECR%DEcRD_IGITAL
VOICE
Fig. 5 Demodulator block diagram
of compensating the fast frequency fluctuation by
Doppler.
A demodulated signal after Viterbi decoding
is sent to a 4.8 kbps voice decoder and converted
to a voice signal in it.
4. PERFORMANCE OBTAINED
The modem has been tested using a radio
channel simulator. The bit error rate and car-
rier phase slip rate are measured in the presence
of AWGN and Rician fading. The BER perfor-
mance is shown in Fig. 6 with Rician factor C/M
of 10 dB and fading pitch fa of 200 Hz (1/16
modulation rate). The deviation from computer
simulation is well within 1 dB at the operating
BER point. To prove the effectiveness of our pro-
posed scheme over the conventional scheme using
a single open loop estimator with narrow band-
width, the performance is measured removing a
wide band estimator. As was expected from sim-
ulation, the proposed scheme showed the Eb/No
improvement of around 1-2 dB over the conven-
tional scheme at the operating 1× 10-2-5× 10-3
BER point. And, a low carrier phase slip rate
of less than 5 x 10 -2 times/sec is realized at the
operating Eb/No of 6 dB with severe Rician fad-
ing, as seen in Fig. 7. The demodulated scatter-
ing diagrams for the proposed and conventional
schemes are depicted in Fig. 8, which shows an
appearance that the fast phase fluctuation by fad-
ing is effectively compensated by a dual open loop
phase estimator.
is presented, which remarkably augments the
demodulator's performance in an LMS environ-
ment. Furthermore, an actual experimental ter-
minal modem has been manufactured to verify
the effectiveness of the proposed idea. The re-
sults are quite satisfactory, achieving good bit er-
ror rate performance and stable synchronization.
This scheme is applicable for different modulation
schemes.
ACKNOWLEDGMENTS
Authors wish to thank Mr. Furuya, Dr. Hi-
rosaki and other members of C&C Systems Re-
search Labs., NEC Corporation, for their helpful
discussions.
REFERENCES
[1] A.J.Viterbi,A.M.Viterbi, "Nonlinear Esti-
mation of PSK-Modulated Carrier Phase with
Application to Burst Digital Transmission,"
IEEE Trans.Inform.Theory, vol.IT-29, pp.543-
551, July 1983
[2] W.Hagmann,J.Habermann, "On the phase
Error Distribution of an Open Loop Phase Esti-
mator," in Proc.ICC'88, June 1988, pp.1031-1037
[3] F.D.Natali, "AFC Tracking Algorithms,"
IEEE Trans.Commun., vol.COM-32, pp.935-947,
Aug. 1984
5. CONCLUSION
A new carrier and bit timing recovery scheme
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International Mobile Satellite Conference, Ottawa, t990
3x10-'-
ld'
10'
_ 10-'!
3x10 -t
• PROPOSED
DUAL OPEN LOOP P.E.
oS,NG'E
_,,, (N'i =101)
_e _ --- AWGN
_,_ _ FADING
"_','_,,_ (C/M=10dB, ld =200Hz)
'_,\x,
\\
II
It
18
Eb/No(clB)
3x10° t PROPOSEDDUAL OPEN LOOP P.E.
10°4 • AWGN (N1 =101, Nz=7)I o FAOING(C/M=12dB, fd=200Hz)& FADING(C/M=IOdB, 1,= =200Hz)
u
10 _
1
10-_ 1
3xlo-'l
o i ; ; ; 1'o
Eb/No (riB)
12
Fig. 6 BER characteristic Fig. 7 Carrier slip rate characteristic
C / M,,,10 dB
fd= 200 Hz
El)/No=DO
C/M=10 dB
fd=200 Hz
E_ No=6 dB
C / M=10 dB
_'d=200 Hz
Bo/No-OO
C / M-10 dB
fa-200 Hz
El3/No=6 dB
CONVENTIONAL SINGLE OPEN LOOP
PHASE ESTIMATOR
PROPOSED DUAL OPEN LOOP
PHASE ESTIMATOR
Fig. 8 Demodulated scattering diagrams
international Mobile Satellite Conference, Ottawa, _990
627


A new coherent demodulation technique for land-mobile satellite communications

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