Tone-aided modulation is one way of combatting the effects of multipath fading and Doppler frequency shifts. A new tone-aided modulation format for M-ary phase-shift keyed signals (MPSK) is discussed. A spectral null for the placement of the tone is created in the center of the MPSK signal by translating the upper sideband upwards in frequency by the same amount. The key element of the system is the algorithm for recombining the data sidebands in the receiver, a function that is performed by a specialized phase-locked loop (PLL). The system structure is discussed and simulation results showing the PLL acquisition performance are presented

Depaolo, Anthony D.

Millar, Gilbert M.

Saulnier, Gary J.

English

NASA Technical Reports Server

N92-24151
A Tone-Aided/Dual Vestigial Sideband
(TA/DVSB) System for Mobile
Satellite Channels
Gary J. Saulnier, Gilbert M. Millar and Anthony D. de Paolo
Electrical, Computer and Systems Engineering
Rensselaer Polytechnic Institute
Troy, New York 12180-3590
Tei:(518)276-2762
Fax:(518)276-6261
ABSTRACT
Tone-aided modulation is one way of
combating the effects of multipath
fading and Doppler frequency shifts.
This paper discusses a new tone-aided
modulation format for M-ary phase-shift
keyed signals (MPSK) . A spectral null
for the placement of the tone is
created in the center of the MPSK
signal by translating the upper
sideband upwards in frequency by one-
half the null width and the lower
sideband downwards in frequency by the
same amount. The key element of the
system is the algorithm for recombining
the data sidebands in the receiver, a
function that is performed by a
specialized phase-locked loop (PLL) .
The system structure is discussed and
simulation results showing the PLL
acquisition performance are presented.
INTRODUCTION
Mobile-satellite channels are
characterized as having Rician
multipath fading and Doppler frequency
shifts. Due to the narrowband nature
of the channels and, in particular, the
magnitude of the expected Doppler
shifts with respect to the channel
bandwidth, the receiver must perform
some type of adaptation to the channel
in order to maintain acceptable
performance. , One approach to
performing this adaptation uses a tone
contained within the signal to
calibrate the receiver to the channel
characteristics. Such tone-aided
modulation has been shown to remove the
irreducible error floor that is
associated with many modulations in the
Rician channel (Rafferty, 1987).
While a number of tone-aided
modulation schemes have been proposed
(Davarian, 1987, Simon, 1986, McGeehan,
1984) each has some drawback which has
detracted from its usefulness. One
problem common to all the tone-aided
schemes is the undesirable bandwidth
expansion of the signal necessary to
accommodate the one or more tones. The
TA/DVSB system discussed in this paper
attempts to minimize the bandwidth
expansion by using a single tone placed
in a spectral null that is only as wide
as is necessary to accommodate the
Doppler spread. Results for a binary
PSK TA/DVSB system have been reported
earlier (Hladik, 1989) and this paper
extends this work to MPSK.
As shown in Figure I, the TA/DVSB
signal consists of a tone inserted
between the frequency-translated upper
and lower vestigial sidebands of a
phase modulated carrier. The modulator
creates this signal by separating the
sidebands and frequency translating the
upper sideband upwards in frequency by
ft and the lower sideband downwards by
the same amount. Although the TA/DVSB
spectrum is similar to that of another
tone-alded system, Transparent-Tone-In
Band (TTIB) (McGeehan, 1984), the
TA/DVSB demodulator uses a different
algorithm for recombining the data
sidebands. A major consequence of the
TA/DVSB recombination algorithm is that
International Mobile Satellite Conference, Ottawa, t990
493
https://ntrs.nasa.gov/search.jsp?R=19920014908 2020-03-17T12:28:06+00:00Z
the vestigial sidebands can be kept as
small as possible (ideally they would
be zero), minimizing the required
transmission bandwidth. The
translation frequency, ft, is selected
to accommodate the expected Doppler
spread while allowing for recovery of
the tone in the receiver. Typically,
ft must equal twice the expected
Doppler shift so that the pilot
recovery filter can extract the tone
without any data sideband energy.
MODULATOR
The TA/DVSB modulator accepts M-ary
symbols and a produces root-Nyquist
MPSK signal whose sidebands have been
separated and frequency translated as
shown in Figure i. The tone is also
inserted in the modulator. A number of
approaches for generating the vestigial
sidebands were considered, including
Weaver's single sideband (SSB)
modulator and the phase-shift SSB
modulator (Carlson, 1975). However, a
third approach in which SSB signals are
directly synthesized was chosen.
Figure 2 is a block diagram of the
MPSK TA/DVSB modulator. Consider the
processing of the in-phase (I) data
first. Impulses representing the I
data are input into two transversal
filters, one having an impulse response
equal to the root-Nyquist pulse shape
and a second having an impulse response
equal to the Hilbert Transform of the
root-Nyquist pulse shape. The outputs
from these filters, then, are the pulse
shaped I data, di(t) , and the Hilbert
Transform of the I pulse-shaped data,
di(t). These signals are then placed
on a carrier of _t, where _t is the
translation frequency, i.e _t=2Kft .
The resulting signal is the I portion
of the baseband TA/DVSB signal,
S I(t)=d I (t)cos0)tt + dI(t)sin_tt. (i)
The quadrature-phase (Q) data undergoes
identical processing producing
SQ(t)=dQ(t)cos0)tt + dQ(t)sin£0tt. (2)
Si(t) and SQ(t) are finally multiplied
by quadrature sinusoids at the carrier
frequency, _c, and summed to create
the modulator output. When a DC
component is added to Si(t), the tone
will appear in the output. The
composite output, then, is
S(t)=[d I(t)+dQ(t) ]cos[ (_c-_t)t]
+ [-dI (t) +dQ (t) ] sin [ (_c-_t) t]
+ [di (t) -dQ (t) ] cos [ (_c+_t) t ] (3)
+ [dI (t) +dQ (t) ] sin [ (_c+_t) t]
+DCcos(_ct) .
Figure 3 is a transmit spectrum
generated using the modulator of Figure
2. The shaping is 65% excess bandwidth
root-Nyquist, _t is 480Hz and _c is
6kHz. The shaping filters use i0
samples per symbol and extend over 8
symbols.
DEMODULATOR
The demodulator consists of a pilot
processor, sideband separation
processors, a phase-locked loop (PLL)
and pulse-shape matched filters.
Figure 4 shows this structure.
Pilot Processor
Figure 5 is a block diagram of the
pilot processor. The input to the
pilot processor is the modulator output
(3) after it has been passed through
the channel which introduces additive
white Gaussian noise (AWGN) along with
multipath fading and Doppler frequency
shift. The pilot processor assumes
that both the tone and the data
sidebands have undergone the same phase
distortions in the channel and,
consequently, uses the pilot as a
coherent reference to remove these
distortions.
The pilot processor first mixes the
input signal to baseband I and Q
signals using a local reference, _LO,
which is approximately equal to _c"
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international Mobile Satellite Conference, Ottawa, t990
Since the carrier of the received
signal will be offset from _c due to
Doppler frequency shifts and multipath
fading, the I and Q signals will be on
a carrier equal to the amount of this
frequency offset and the pilot tone
frequency will be equal to this offset
frequency. Each signal then is broken
into two paths, one which uses a
lowpass filter to recover the pilot
tone, Ip or Qp, and a second which
delays the corresponding I or Q signal
by an amount equal to the delay of the
lowpass filter.
The pilot tone components, Ip and
Qp, are then subtracted from the
corresponding I and Q signals,
effectively implementing highpass
filters. Ip and Qp are finally used as
coherent references to remove the
frequency offset from the I and Q
signals. In the absence of noise the
pilot processor outputs, I O and QO,
will equal the I and Q signals in the
modulator as given by (I) and (2).
Sideband Separation Processor
The sideband separation processors
act to isolate the upper and lower
sidebands from each other, delivering
each to the PLL. While this separation
can be performed using lowpass and
highpass filters if the signal is
placed on a carrier, this type of
bandpass processing is expensive in a
DSP format. Consequently it was
decided to perform the sideband
separation at baseband using a Hilbert
Transform filter. The resulting
baseband SSB signals will be complex
signals.
Figure 6 is a block diagram of the
sideband separation processor. An
input signal, x(t), is split into two
paths, one which uses a transversal
filter to perform a Hilbert Transform
and a second which introduces a delay
equal to the delay of the Hilbert
Transform filter. The Hilbert
Transform filter output, x(t) is used
to create the two SSB signals. The
upper sideband (USB) is
XUSB(t)=x(t) + j_(t) (4a)
and the lower sideband (LSB) is
XLSB(t)=x(t) - j_(t). (4b)
Figure 7 shows recovered USB and LSB
signals which have been placed on a
carrier of 6kHz for display. The
Hilbert Transform filter used was of
101st order.
Phase-Locked Loop
The PLL operates on the outputs of
the sideband separation processors and
locks a local oscillator to ft, which
is then used to recombine the data
sidebands. It is important to note,
however, that the PLL does not have to
track any channel perturbations since
the pilot processor removes these prior
to the PLL. The output of the PLL is
the recombined data sidebands which are
then processed by filters that are
matched to the root-Nyquist pulse
shape.
As shown in Figure 8, there are four
inputs to the PLL, the USB and LSB
signals derived from each of the pilot
processor outputs, I O and QO" At the
heart of the PLL is a voltage
controlled oscillator (VCO) which
generates a local approximation to _t
which will be indicated as _r- The
two VCO outputs are
rl(t)=exp[j(O}rt + 0 r) ] and (5a)
r2(t)=exp[-J(£°rt + @r +K/2)], (5b)
where 0 r is the phase of _r"
Consider the processing of the USB
and LSB signals which originate from
the I signal. These signals can be
expressed as
USB (t) = [d I (t) +jdI (t) ] exp [-j (0)tt+@t) ]
and (6a,b)
LSB (t)= [d I (t)-JdI (t) ]exp [+j (G)tt+0t) ],
International Mobile Satellite Conference, Ottawa, 1990
495
where the phase of £0t is 8 t. USB(t)
is multiplied by rl(t) while LSB(t) is
multiplied by r 2 (t) . The complex
conjugate of the LSB(t)r 2 (t) product is
then multiplied by the USB (t) r I (t)
product to obtain the in-phase error
signal
e I (t) = [di 2 (t) -di 2 (t) ] sin [2 (£0t-_ r) t
+2 (@t-Sr) ] +2dIdiCos [2 (£0t-0) r) t (7a)
+2 (St-_ r) ]+imaginary terms,
where the imaginary terms are not
listed since they are discarded.
Processing of the USB and LSB from the
Q signal results in the error signal
= [dQ 2 (t) __Q2 (t) ] sin [2 (_t-_r) teQ (t)
+2(St-Sr) ]+2dQdQCOS[2(_t-_r)t (7b)
+2(St-_r) ]+imaginary terms.
These error signals contain the
desired sin[2(_t-_r)t-2(St-Sr)] term
which, when minimized, would drive _r
and _t to the same frequency and
phase. The problem, however, is that
this term is multiplied by a
coefficient that is dependent on the
data and its Hilbert Transform. In
addition, there is a cos[2(_t-_r)t-
2(St-Sr) ] term which, if minimized,
would drive the VCO to a quadrature
lock.
Simulation of the PLL using either
ei(t) or eQ(t) as the error signal
indicated that the loop could not
acquire lock. Actually, for small loop
gains, the PLL would tend to maintain
its initial phase value, indicating
that ei(t) and eQ(t) have zero mean. A
later analysis showed that the
coefficients for both the sine and
cosine terms are zero mean.
Through simulation it was determined
that hardlimiting the error signal
makes the PLL acquire and maintain
phase lock. This hardlimiter is
indicated in Figure 8, where the error
signals from the I and Q channels,
ei(t) and eQ(t) are first summed and
then hardlimited before being sent to
the VCO. As will be shown in the next
section, this PLL has a considerable
amount of self noise which restricts
the maximum PLL bandwidth and thereby
limits its acquisition performance.
Finally, examination of (7)
indicates that the error signal will
not suffer from self noise if the data
signals, di(t) and dQ(t), do not make
any transitions, since the Hilbert
Transform terms will all be zero.
Consequently, transmitting the same
data symbol several times as a preamble
is viable approach for obtaining fast
acquisition. This property will be
demonstrated in the next section.
SIMULATION RESULTS
This section illustrates the
performance of the PLL by showing a
received eye diagram and a number of
PLL acquisition curves. The channel
was not included in these simulations
and, consequently, the pilot processor
was also omitted. The modulator
output, at baseband, was fed directly
to the sideband separation processors
which, in turn, fed the PLL. 8-PSK
signalling with root-Nyquist pulse
shaping (65% excess bandwidth) was used
in the simulations.
Figure 9 is a received eye diagram
obtained after the PLL has locked. The
loop gain for this diagram was 10 -4 , a
value that produces a tight lock point.
This eye diagram was generated after
the signal was match filtered. The
inter-symbol interference evident in
the eye diagram is most likely due to
distortion of the low frequency
components caused by the sideband
separation operation in the modulator.
Figure i0 is a set of PLL
acquisition curves for a random data
input using loop gain as a parameter.
The self noise caused by the presence
of the Hilbert Transform components in
the error signal is evident. Figure ii
496
International Mobile Satellite Conference, Ottawa, _990
is a set of PLL acquisition curves for
a constant data symbol input, i.e. no
data transitions. These acquisition
curves are linear due to the
hardlimiter in the error signal path.
Clearly, fast acquisition of the ft
signal is possible with a preamble
consisting of a constant data symbol.
CONCLUSIONS
This paper shows that the TA/DVSB
system is a viable approach to tone-
aided modulation. The baseband
implementation of the modulator and
demodulator is of relatively low
complexity. For burst-mode
communications, where fast acquistion
of ft is required, a preamble
consisting of the same symbol repeated
a number of times is required due to
the self-noise of the PLL.
REFERENCES
I. Carlson, A.B. 1975. Communication
Systems. McGraw-Hill.
2. Davarian, F. 1987. Mobile digital
communications via tone calibration.
IEEE Transactions on Vehicular
Technology. VT-36, pp.55-62.
3. Hladik, S.M., Saulnier, G.J. and
Rafferty, W. 1989. A tone-aided dual
vestigial sideband system for digital
communications on fading channels.
IEEE Military Communications
Conference. pp.709-714.
4. McGeehan, J.P., Bateman, A.J. 1984.
Phase-locked transparent tone-in-band
(TTIB) :A new spectrum configuration
particularly suited to the
transmission of data over SSB mobile
radio networks. IEEE Transactions on
Communications. COM-32, pp.81-87.
5. Rafferty, W., Anderson, J.B.,
Saulnier, G.J., and Holm, J.R. 1987
Measurements and a theoretical
analysis of the TCT fading channel
radio system. IEEE Transactions on
Communications. COM-35, pp. 172-180.
6. Simon, M.K. 1986. Dual-pilot tone
calibration technique. IEEE
Transactions on Vehicular Technology.
VT-35, pp.63-70.
Ft
Fig. i. TA/DVSB frequency spectrum
cos(re,t)
COS (_t)
Fig. 2. TA/DVSB modulator
5)
I
/"_< ,, / b--\
,(
/ ' /
\
I
Fig. 3. Modulator output spectrum
I i
I
PLL I I
=2
×
Fig. 4. TA/DVSB demodulator
International Mobile Satellite Conference, Ottawa, 1990
497
COS (_)
._(t.]_ c,.x,_,,,
SIN (o_ug)
Fig. 5. Pilot processor
d - USB
DELAY _1
la L.BEZ_T xln-j;.,
TRANSFORMER LS_
Fig. 6. Sideband separation processor
1
"'el. .,v.j"
e
_M
• r1[_
{
('
i
i
i
I
"2
Fig. 7. Recovered sideband
COMPtEX
CONJUGATE
7
VCO
COMPLEX ICONJUGATE
HARD
IIkUTKR
Fig. 8. TA/DVSB phase-locked loop
#
_,_ (A) 1 x i0 -? IB) 5 x I0 -] (_') I x 10 -3
_ . ( C)
_o
Fig. i0. Acquistion curves for random
data
IO
70
to
5o
40
30
_o
_o
o
-_o
(A) 1 x 10 -2 (B) 5 x 10 -3 (C) 1 x 10 -3
................ (C)
' \\
\\,\ \\
\,(A) "\_ (B)
'\
IBzl( S*,v eO,.5)
Fig. Ii. Acquistion curves for no data
transitions
498
International Mobile Satellite Conference. Ottawa, '1990


A Tone-Aided/Dual Vestigial Sideband (TA/DVSB) system for mobile satellite channels

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