Computer aided design of the new Crossing Current Resonant Converter (XCRC) is introduced. The XCRC enables zero voltage switching without excessive voltage and current stresses. This makes the converter operate with minimum dissipation. A systematic way of computer aided design of the XCRC is also introduced. Spreadsheet programme incorporates circuit equations for calculation of component values. These values are then used for circuit simulation which further probes into the waveforms before the hardware is actually built. Finally the circuit is built and experimental results are included.published_or_final_versio

Pong, MH

Poon, NK

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Title Computer aided design of a crossing current resonant converter(XCRC)Author(s) Poon, NK; Pong, MHCitation Iecon Proceedings (Industrial Electronics Conference), 1994, v.1, p. 135-140Issued Date 1994URL http://hdl.handle.net/10722/45961Rights Creative Commons: Attribution 3.0 Hong Kong LicenseComputer Aided Design of a Crossing Current Resonant Converter (XCRC) N.K.Poon* M.H.Pong** * BSC Electronics Division, Hong Kong ** Electrical & Electronic Enginering Department, Hong Kong University, Pokfulam Road, Hong Kong Abdmcf - Computer aided design of the new Crossing Current Resonant Converter (XCRC) b introduced. Tbe XCRC ea8bkr zero voltage mitcaimg without ertcuive vdhge d curreat rtrerrer, This makes tbe converter computer aided &sign of tbe XCRC & .Ira introduced, Spreadsbet progrrrmme incorporrter a d t  equrrtiou for dcuktion of component vduer 'IBere valuer a n  then used lor drcuit simuhtioa which further probes Finally tbe d d t  & built 8nd experiwntd results 8n induded. e p m t e  d t h  laidam diuip8th.  A SyStCEl8tiC W8y Of ht0 tbe W8VCfOcpDS bCfOrr tbe hdW8R b 8dU8Uy built. INTRODUCTION Resonant ConVQtQs are well-known for zero-voltage or z e " t  operations which enable them to aperate with low switching lorr#s. However, bacause ofthe resoMnt w a ~ ~ i n v o l ~ t h e s w i t c h i a g d e v i o e s  ar ubjectedto high voltage or-t stresses. In some cases the overall dissipation in the switching device is inaeased AlUW Crossing Cunmt ReSanant Converter (XCRC) is introduced which enables zero voltage switching without extra voltage of cumnt stresses. This rnllrra device dissipation and enables operstion without addit id  heat sinks, thus greatly reduxs the S k  of the converter. Like all atba am-, the XCRC switches between different states. In order to analysis and design the XCRC, a ?y&matk computef aided design approach is used. Firstly, the Circuit component values are calculated using a spresdshed computer prognun which is widely available in pe"l computers. This gives a quick and inexpensive solution to carry out a preliminary design. The results so OMaiOed are then applied to circuit simulation when waveforms can be simulated and checked. Onecancompareandexamine the results calculated by the spreadshaet and those OMaiOed f" simulations and d e  adjustments as required. This process can be itaatcd until the designet is satisfied with the results. Then the designer canbuild the hardwan w i t h a " e .  This paper starb with a descriptionofoperatioaof the new Crossing Current Resonant converter (XCRC). Analysis of the Circuit theal fdlows, the equetions so derived are then implearented to a spnadsheet program. Tbe d t s  so OMnimd are used fa circuit simulation. Finally a circuit is built and the wavefm OMaimd fKm the con- circuit is shown. 135 0-7803-1328-3/94$03.000 1994 IEEE CIRCUlT OPERATION OF THE XCRC The scbanatic diagnun of the XCRC is shown in Figure 1. T h e h u i t  is similar to a halfbridge topology butthe circuit aperation is quite difkent h m  collv"! reamant circuits. Them cepcreitaa Cl 8nd c2 have d values and theirvoltclgesare allowrd to swing up and down within thelinevoltage. ~diodesarecamctedinpsrallelwith tbcseerrpacitas.csldcgL~MosFETdrain-8ounx capacitance which are inhaent with the device. By d d e r i n g  the current thmugh the Primary inductor Lr, we can identify thee warking states, namely the A) Power transfa state, B) Current circulating state and C) Cmssover rrsaoant state. By Symmdric operation we shall CoaPidcT a halfcyck behaviour. A) Power tmn.$er state (Fig.2) While MI is turned on, current will flow through the transformer and charge up C2 until it reaches the supply voltage. Ofcoucpe. CI is being dischaqed at the same time. Dumg this state, the transf- will "sed' a voltage applied to it's taminals, so sacoodary voltage will build up and powerwill be tiansfare& hm the primary si& to the se€cduy si& 8nd t i l t e d  by the output tiltcr. This state will continue until the voltage Bcn#18 cz goes up to the supply voltage and turns on the diode DI. Voltage ~ccoss C2 will beclamped to the supply voltageand farce thecircuit to go to the "Current ckdating state". lkoughout the design, continuous cumnt through the output filter inductor is essumed. B) C u m t  cimthting state (Fig. 3) While MI is still on, the primary current will form a circulating current loop as illuscraed in Fig. 3. The voltage Bccoss the transformer end the series inductor Lr will drop to approximately m. In practice, the voltage BQDSS each component in the loop is nearly zar, ifthe diode voltage drop is neglected. Asa result, the di/dt in the loop is ucarly zem and cumnt circulates at a amitant magnihde. This state ends when MI is huned &. This state controls' the output power by varying its time p a i d  This malres the XCRC a frerluency modulated convater. I A 1 1  I Fig. 2 Power Td' State Fig. 3 Cumnt Circulating State I T  c 1 1  I Fig. 4 C-vm Resonant State C) Cmssover " a n t  state (Fig.4) At the begianing of this state, Mi and Mz m tumed off to allow for a dead time in order to avoid simultaneous umduction of Mi and Mz. The Primery curreat discharges csz ofM2 to -0.n and cumnt will be diverted through the body diode of MOSFET Mz (which is not illustrated). While the body diode ofM2 is amducting, it maintains a low drain source voltage of am diode forward drop. This facilitates the ZCIO voltage condition for Mz. Mz is then tumed OII which enables negative aad positive current to go thtough the MOSFET. The opedon cycle then rrpeats with the power transfer state. The Crossover resoaent state is less three difhtat modes in this state. More detailed straight faward than the f a m r  two states. In fact thae are dtsaiptioas can be f d  in the following d o n .  CIRCUlT DESIGN USING SPREADSHEET An arpaiarad power electronics enginea can tell that the design of a power electFollics circuit oftar involves a lot of engineaing judgment and compcomises among many factas. VeryoRen the eaginea has to choosc fnwn a range of values for a comparent, but yet he has to chedr his choice a@nst faetas lilreopaatiag voltage range. maximum and "um lod. cspabilities of the controlla. current or voltage sbrsscs upon the power devices, etc. In ader to make t h e e  choice forthe component c"ed while keeping the overall conwrterperfonnaace inhand, ahaody aad responsive design tool is necess~fy and the spresdsheet program is a good candidate. In this Section the XCRC is dysed  in detailed. The equatioasareapplicd to aspnadshedp". Resultsare pRxeated in graphical form. simuwed waveforms sccompenytheanalyticalequatioas. The XCRC circuit to be analysed is shown in Fig. 5. Throughout the analysis, continuous condudion is assumed and thus the filter and the load are rrpnsented by a c " t  current source. The waveforms c"ed (simulated) are shown in Fig. 6. The three operating states are indicated OD the diagram. V. Fig. 5 Continuous ogeration mode 136 Fig. 6 The three operating states A) Power tnmsfer state In this state the primary voltage across the t r a n s f m  start at a cettain voltage V e  which is ddemmed - bythe previous state. The amount of energy t r a n s f d  by the tnmsfclnnaisrepresentedby where vp is the traasfixmex primary and ip the trausfixmer pimarycurrent. As the semdary current is CoIlStpllt in the time period the primerycurrent is also amstant. Tbe primary cumnt cluugs up c2 and discherges c1. KCI and c2 have the same value and equal to c, the primary ament is in fnct charging up a total capacitance of2C. The capacitor voltage V a  ramps up linearly in time tA whae 2C.( Vin - V, ) t, = (2) 'P where Vw is the voltage a~oss C2 at the beginning of this state. It will be described in mote detailed when the ~ossova " n t  state is analysed As the cnpacitance voltage ramps down k l y  and ip is constant, the integral inequf3tion(I)canbe represented by the area under the triangutst waveshape of vp. 1 2 E = -t,.(Vin- VW).ip (3) = C.(Vin - v~., 1' with a switching fi-equency of fs, the power transferred in the two sy"e&ical half cycle is The output power is controlled by the s w i ~  iktquency pOWUiSinhaenllYLimitedbythCSWitChiIlgfi-eq~. fs and the charging of the capacitors. It is obvious that because of this powx transfer via the capacitom, output B) C u m t  cinrhting state Current now flows through one of the rectifier diode on the secondq side in this stak. Again, cumnt is assumed coascantinthetimepaiodconcermd.variationofthetime the ovaall switching fi-eqwncy E3 and period@- . regulates the output voltage. C) Crarsover " a n t  state Thae are three modes in this state. The simulated wavef- are show in Fig.  7. In the fitst mode (tz), the t h e c i "  voltage of the lower transistor to m. (Fig. 8) This facilitates the zero voltage tum on of the lower taptransisto€tunwof€. The drain sourcecapecitance ofthe twotransi~resooatt with the inductor Lr, which brings In order to ensure that the drain-source voltage swings to zero, the following aiteria must be met, This defines the limiting value for the inductor La. The time lapse tz f a  the Vds of the lower transistor to fall to zero is t, =-.sin-'[ 1 ,1 2VmCW W Wherc (6) 1 @ P P  JZZ 137 I 1 - I 113 :Sus Time Fig. 7 The simulated Crossover resonant state Once the Vds of the transistor reaches zero reverse current goes through the body diode of the transistor M2. In this mode the inductor Lf experiences the reverse voltage of vin whi&reduces thecumnt towards zero. The time @se for this period is Lr.ik, ti =- vm where ilr'=ip.coscDt, . (7) In orda to achieve z m  voltage switchin& M2 must be fundioasofthe pimerycunmt d voltage vis the insrant selacted turn 011 time of the transistor with respect to diEe"t values of tz and tf. switched 011 during the tf. AS both tz a d  tf a at whicaM2 switches an should be chosen with respect to the diffamt Opaatiog amdific~~. Fig 8 shows that the V a f W m d k & W  0.6 0.5 0.4 - 3 0.3 5 0.2 0.1 0 Q.tVW1ZOV . 0 0.5 1 1.5 0 Tbe UliFd mode in this state is tf during which positive current flows in the t"&tar. Now the inductor h a n d  the twocepscitas C1 andC2 fmm aU= " a u t  circuit. The current swiags up fiom ZQO until it takes up the WhelingState on the secoedsryside. This modeends with the reappeanuvce ofthe transfamr primary voltage. Hence the power transfa state daraibed eaclifx COMS in and the cyde continues. COmSPOPl(iiOg secoadary load Current Md stops tbe b- Tbe time tp ofthis mode is the solution of a Land 2C reacnrrt circuit with zero initial cumnt and current ip at the end ofthe time period bp. t, =- a a, L v i n J  1 This time period tf is important in the determination of the power t r a n s f d  in each period. This is because tzn determines vw in equation (2). vw=-'f-sina,t 1 v i  2 c  L.a, Itcanbeshownthat vin - vw = vin.coscD t , and hence 6ufn equation (2) 2 c  t, =-.vin.cosa, t, iP (9)  138 The presence of various time periods limit the opaating mexi" fhpeDcy the time period Q in the current circulating state is set to ZQO for "um power output. Themaximumfkqueacyisrcpresmtedby E-equency of the converter (Fig. 9). Ass\rme that at the 1 f- = t, +t,+t,.+tZ" the variation of finax is shown in Fig. 10. S M i  times atVil2ov 0 0.5 1 1 .S P Fig. 9 Variation of tz, tf, t y  and tA Maximum Frequency O.3 T Vm= 1 0.1 0.05 + 15ov 04 I 0 0.5 1 1.5 iP Fig. 10 Maxi" Switching Frecluen~y Theplesenceofmaxi" switclung fhquemyimposesa limit on the power output which is shown in Fig.11. Fmn tluchnchwb ' 'c curves shown, this convata is most suitable for constant CUtTent application although it can also be used as amstant voltage umverter. The optimum opaatiog primary current, ip, can be chosen which carespoadstothedesirsdapaating range. where Np and Ns are the primary and scmduy hans rtspedively. Po is the output power and Vo the output voltage. 40 35 f: 10 :I 5 04 I 0 100 200 300 w w  Fig. 11 Output power limit Also, Np has to have a minimum no. of hnns to avoid cofe Sahaation, by which 1 Np.B,.A, 2 -(Vin - Vaw). t, 2 2 (14) C.( Vin.cosa, t $.) i P .B,.A, and N p 2  where & is the maximum flux density of the core, and Ae isthemagneticaosssecLional area. A f h  confirmation of the design by SPICE circuit simulation, power of the canvater is 20W with voltage at 12V. Iaductor Lrhascminductance of 41uH, CI and C2 are 5.8nF, the transfonna aunS ratio is 1.7. The wavefonns caphm?d are shown in Fig. 12 which have been corre~tly prrdicted by sirnulatiom. M a e  detailed waveforms showing the Crossover current state are shown in Fig.13. The MOSEFTs used are m30 with To-220 heat sink and the tempeaature rise is merely 8°C above ambient kmpe". 811 off-line convater for IlOVac input is built. The ourput which can operate without any 139 l&Jc Running: 50.0MS/s Sample . . . . . . . . . . . . . . . . .  . .  r ; :  -.$ . . : . . . . .  Chl : Vgs(Mz), 20V/D ch3 : V(c2), 50vm Ch2 : Vds (M2X lOOV/D Ch4: id(M2). O.SA/D Fig. 12 Capbed waveforms 7&Jc Running: SOOMS/s ET Sample +---------f-r+------i [ " . "  , . . , . . . . /  . . / . .  i .  . i .  . ,  , , , . . , . .  , REFERwcEs [l]M.lLPoog "t3mpkr Aided Analysis of a Buck Zao- c m  switching Rnrarant calvatez", I k c e d q p  of Po- Essiaeering c4mf.kealcC IPEC, t h e h t a n a b d  - ~p.543-548, SingaPae, March 1993. [2] W.C.Ho. M.H.Pong "Power Loas and Efficiarcy Analysis o f Q u 0 s i ~ t c a l ~ " h ~ d l c o a f a a L c e o n  Idushid El-, Coatrol and Idnmea~tation W O N  93, Hawaii, Novanbtx 1993. Chl : V&M2), lOV/D Ch3 : Vp, SOV/D Ch2 : Vds (M2). H)V/D Ch4 : i-), O . S A / D  Fig. 13 Waveforms at the Crossover current state CONCLUSIONS A wvd Crossing Curwrit Resonant Converter (XCRC) is advantages of zero voltage switching, and mini" voltage andaarent streasesupon the switchgdevices. A syskodc d y s i s  aud desigp approrsch is adoptea 'Ihe circuit is analysed and the equations involved implemented to a spresdsbeet prognnn which is a versatile and low cost tool. in- This frequency modulated converter has the computed results aIz presented in graphical form and the opastiog points arc cboscn with rrspect to the operating limits. Design results tue a"ed by simulations and an -tal XCRC is built- Tbe experkntal results a" 140 

Computer aided design of a crossing current resonant converter (XCRC)

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Computer aided design of a crossing current resonant converter (XCRC)

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