Contains research objectives and reports on status of research projects.National Aeronautics and Space Administration (Grant NGR-22-009-163

Chen, A. Y. C.

Gerrath, K. H.

Majer, J. W.

Mohlere, R. D. V.

Peterson, D. F.

Rafuse, R. P.

Rudzki, J. E.

Saleh, A. A. M.

Steinbrecher, D. H.

Webb, R. G.

DSpace@MIT

VIII. SOLID-STATE MICROWAVE ELECTRONICS*Academic and Research StaffProf. R. P. RafuseDr. D. H. SteinbrecherGraduate StudentsW. G. Bartholomay D. F. Peterson R. W. SmithA. Y. Chen J. E. Rudzki R. E. SnyderR. D. Mohlere A. A. M. Saleh J. G. WebbRESEARCH OBJECTIVESOur major research objective is the same as it has been for many years, althoughit has not been previously stated explicitly. First, we attempt to discover fundamentalperformance limits to be placed on functions performed by solid-state devices, in termsof the technological and physical parameters of the devices. Second, we attempt toprovide the necessary circuit imbeddment procedures to realize the performance limits.These two areas combine to cover the analysis and synthesis of high-performance,solid-state circuits without regard to frequency, function or power level. Most of ouractivities are confined, however, to analogue circuits of low or medium power level.Specifically, we are interested in the development of low-noise, high-dynamic-rangesignal-processing circuits, low-noise high-efficiency solid-state receiver and trans-mitter technology for millimeter waves, and high-efficiency, extremely small trans-mitters for use in tracking of, and telemetry from, wildlife.R. P. Rafuse, D. H. SteinbrecherA. STATUS OF RESEARCH1. FREQUENCY DOUBLER 15. 2-30.4 GHzA frequency doubler from 15. 2 GHz to 30. 4 GHz has been designed (Fig. VIII-1)and put into operation. The structure consists of input and output step transformersand coaxial sections that couple the input port to the diode. The diode is mountedin the output waveguide.By adjusting the dimensions of the step transformers and the coaxial section, theoutput and input impedances of the doubler were made as in Fig. VIII-2. The cor-responding conversion loss (Pi/pout ) versus frequency for pin = 160 mW is givenin Fig. VIII-3. The bandwidth for 3 dB increase of conversion loss is ~1.4 GHz.The necessary change of the biasing voltage within this bandwidth is -3. 5-6. 5 Volts.The conversion loss can be decreased by further adjustment of output step trans-former (to increase the coupling of the output port) and by adding in the properThis work was supported by the National Aeronautics and Space Administration(Grant NGR-22-009-163).QPR No. 92... .............. ..... . _..... ........ ........ :, : -, _ ( ~........ ....... ... _ ~ ( ~. .... ..... ..... .... . .... ........( (.. .... .. ... .. ..... ... .. . ~~ ' ' '~ ';: :.................... ....... ........... ~ .............. ...... :,: ....... I .......... ~ ~ .................. I ... . . . .. ............... . ......... .... : ......... ..... ......... ~ . ......... .. ......... ........ . . ~ ............ ..................~ . ... ~ ... ... . :, * .... . .. .. ... -~ ~ ~ ... ~ ..................... . ...... ... .. .. .. .. . :.. . . ... ... : .. .. .... .. -................... .... .... . ............. ... -.... I . ~~ ................. : -........ .. I ........... .... -....... '': ... . .. .. - , '* .. . ............~ .. ... .. .... *:.... : ... - ~...... ...... ...... ........ .... ............... ..... :.- - - .., ............ ... ...... ........ i -. ..................... .. ...... -.--............... ..........-................. .. I ... ............~.................... ... ........ ..... ............. ...... ........... : ........ -.. . . . .. . . . .. . . ........... . . . . . . . . ..... .... ... .... .... ....... .........::: :::. . .. . .... ...... ... :.... : :............. .......... : : :: : :: ::.................... ::: : :.... . I......... ... :: : ::I ..... .  ... I ..... ..... ......i j.... : : : : : : :: : : : : : : : :............ ..........................w )Fig. VIII-2. Input and output impedances vs biasing voltageof the frequency doubler.QPR No. 92140 145 150 155F (GHz)Fig. VIII-3.Fig. VIII-4.Conversion loss vs frequency of the frequency doubler.Input and output impedances of the frequency doublerwith EH tuners at input and output ports.QPR No. 9210-160(VIII. SOLID-STATE MICROWAVE ELECTRONICS)place reactive elements (posts or irises) at input and output. An unwelcome consequenceof this would be a decrease in the bandwidth.By placing EH tuners at the input and output ports, the input and output impedanceswere modified according to Fig. VIII-4. The conversion loss was 5. 3 dB (efficiency29. 5%), the bandwidth 80 MHz.J. W. Majer2. L-BAND QUADRUPLERTwo years ago, theoretical solutions for all-idler punch-through varactor multiplierswere presented.1 An experimental verification of the theory2 has now been completed.Computer-aided circuit analysis was used to synthesize an imbedding network that pre-sented the punch-through varactor with the optimum impedance-frequency characteristicsFig. VIII-5. Experimental 0. 9-GHz punch-through varactor quadrupler.as predicted by the theory. The resulting circuit is shown in Fig. VIII-5. The multi-plier efficiency was 53% as compared with 52% calculated, and the power-handling capa-bility was within agreement with theory. The maximum L-band output power was 8 W.QPR No. 92(VIII. SOLID-STATE MICROWAVE ELECTRONICS)A sampling resistor was included in the circuit to monitor the varactor cur-rent waveform. A comparison of theoretical and experimental values for thee5 - MEASUREMENT FOR4 Pin= 6 Watts3-J S- MEASURED01r - I , 11-2 CALCULATEDoFig. VIII-6. Calculated and measured varactor current waveforms.varactor current waveform is displayed in Fig. VIII-6. Efficiency vs input poweris given in Fig. VIII-7, both with and without the 0. 5-Q sampling resistor. The504030200.2 0.5 1.0 2.0 5.0INPUT POWER (Watts)2.0304.0ULn0-J5.0zO06.0 crz7.0 010 20Fig. VIII-7. Efficiency vs input power for the quadrupler.0. 9-0 series resistance of the diode was obtained from the small-signal imped-ance vs bias contours at the multiplier output (Fig. VIII-8), with and without thesampling resistor.QPR No. 92A: QUADRUPLER WITHOUT SAMPLING RESISTORB: QUADRUPLER WITH SAMPLING RESISTOR-0- @-I-I I I I(VIII. SOLID-STATE MICROWAVE ELECTRONICS)fo = 900MHzFig. VIII-8. Small-signal output impedance vs bias voltage.The experimentally obtained current waveform was analyzed and the harmonic con-tent was found to be in excellent agreement with the results determined theoretically.K. H. GerrathReferences1. D. H. Steinbrecher, Ph. D. Thesis, Department of Electrical Engineering, M. I. T.,July 15, 1966.2. K. H. Gerrath, "The Design and Realization of Varactor Multipliers for the VHF-UHF Range," S. M. Thesis, Department of Electrical Engineering, M. I. T. , Septem-ber 1968.B. TASKS IN PROGRESS1. AVALANCHE-TRANSIT -TIME OSCILLATOR ANALYSISAn avalanche diode oscillator with separate tuning of both the fundamental andsecond-harmonic frequencies is being investigated. The frequencies were separatedin order to observe any variations in the noise and output power of the fundamen-tal which might be caused by higher harmonic tuning. The fundamental (9-11 GHz)QPR No. 92(VIII. SOLID-STATE MICROWAVE ELECTRONICS)was taken out through a coaxial lowpass filter and tuned appropriately, while the secondharmonic (18-22 GHz) was tuned by a K-band adjustable waveguide backshort. The low-pass filter essentially isolates the two frequencies.The experiment consisted of tuning the fundamental for maximum output power witha matched load on the high-frequency port, and then replacing this load with the adjust-able short and observing any changes in power and noise as the position of the short wasvaried. The power-level variations were not large (2-3 dB), but the noise performancewas rather marked. The spectrum could be greatly enhanced or degraded at certainpositions of the short. Apparently, both the second and third harmonics affect the output,S 0.3 dB . 80 mWE \ - BEST AVERAGES2.4dB I WORST 1 CAESWORST CASE NOISE PICTURE NO.3: ICASE WORST CASE 3BEST CASE 4X 2,g/2 AVERAGE 2POSITION OF SHORT, xZs =jZ o tan(B 2x +(0)1. MATCHED LOAD3. WORST CASE2. AVERAGE4. BEST CASEBANDWIDTH = 1 MHzDISPERSION = 10 MHzLINEAR SCALEFig. VIII-9. Illustrating changes in noise performance vs second-harmonic tuning as seen on a spectrum analyzer.QPR No. 92(VIII. SOLID-STATE MICROWAVE ELECTRONICS)since the output performance repeated itself at half-waveguide wavelengths of these fre-quencies. Also, there is a correlation between the power-level drop and the noise per-formance, as seen in Fig. VIII-9. The graph is the approximate effect for each of threeVarian Associates diodes used in the circuit. They all exhibited a power-level droop andchanges in noise performance, but some were worse (or better) than others. Detuningthe fundamental usually results in a poorer noise performance, but it can be improvedagain by retuning the second harmonic.D. F. Peterson2. 60-GHz MIXERPending completion of the machining of the balanced mixer diode mounts, a single-ended mixer was constructed and tested. With the image frequency terminated with thesource impedance and the sum frequency reactively terminated (less than ideal condi-tions), a double sideband noise figure of 25 dB was measured, by using a 5. 5-dB noisefigure IF amplifier. Significant improvements are expected in the balanced mixer withthe proper terminations.J. E. Rudzki3. MIXER ANALYSISFor any nonideal diode model both the single-ended Y mixer and the balancedG mixer show a dependence of the conversion loss on the source impedance. Thequestion is, "How sensitive is that dependence?" It has been shown by usinga simple diode model that for a given source impedance and the same diode theo mixer is superior to the Y mixer. The waveform driving the diode in theY-mixer circuit was a rectangular pulse with a nonzero width, which is a functionof the diode and the source impedance. This has proved to be the best drive wave-form from all classes of real waveforms. The G mixer was assumed to be drivenwith a square wave (pulse duty ratio one-half) which was the optimum drive for thismixer with any diode and any source impedance used; this is another advantage ofthe G mixer.The question of intermodulation distortion in a doubly balanced broadband mixer,caused by the switching of the diode by the signal, was investigated. We found that anonlinear memoryless transfer function followed by an ideal mixer is equivalent to theactual mixer circuit. This facilitates the calculations of the amount of the nonlineardistortion. Investigations are now taking place to compare the aforementioned mecha-nism of distortion with other mechanisms.A. A. M. SalehQPR No. 92(VIII. SOLID-STATE MICROWAVE ELECTRONICS)C. NEW TASKS1. INTERMODULATION DISTORTION IN MIXERSWe desired to build a spectrum analyzer whose principal function is the measurementof intermodulation distortion in mixers. The device should be capable of measuring theIM products accurately down to very low levels. Spectrum analyzers that are now avail-able have limited dynamic range and inaccurate readings, for several reasons that willbe discussed here.The most severe test of the IM distortion of a device is the measurement of the out-put with a two-tone signal applied to the input. Then, in the frequency spectrum, theIM products will appear at evenly spaced intervals on either side of the primary outputs.In this context the term "primary outputs" refers to the output of the device that wouldresult if the device were ideal. The actual generation of the two-tone signal can be mostcritical, as any coupling between the signal sources will result in IM products at theinput, and the output measurements of the device under test will suffer unwarranted deg-radation. Another problem in the measurement of IM distortion is the close proximityof the IM products to the primary output of the device under test. The measuring or dis-play instrument is then faced with the problem of accurately giving a readout of a smallsignal in the presence of a large signal, under the assumption that the device under testhas any efficiency at all. Because of dynamic range limitations, the accuracy of thismeasurement in most instruments is usually negligible when there is more than 70 dBdifference between the primary outputs and the IM products.These difficulties will be eliminated by the utilization of a bridge technique in theinsertion and extraction of test frequencies. The bridge will utilize sum and differenceproperties of hybrids to yield an output consisting of IM products only.The manner in which the inputs are divided and the outputs summed practically elim-inates any degradation in the distortion measurement because of imperfect isolationbetween the two signal sources. The elimination of the primary outputs in the displaymeans that IM products can be measured to much lower levels without increasing thedynamic range of the equipment.Thus far, 6 hybrids have been built with isolation between ports of greater than 35 dBto 100 MHz and 20 dB at 150 MHz. Insertion loss is less than 2 dB per branch, andVSWR is close to 1. 2.R. D. Mohlere2. HIGH-POWER 60-GHz SOLID-STATE SOURCEThe diagram in Fig. VIII-10 illustrates the last two stages of a 100-mW 60-GHz solid-state source. Multiplier designs have been completed and tested andQPR No. 92(VIII. SOLID-STATE MICROWAVE ELECTRONICS)QUADRATURE HYBRIDSFig. VIII-10. 60-GHz solid-state source.construction details are being worked out.A. Y. Chen3. NONLINEAR CIRCUIT THEORYThree network elements, R, L, and C,their derivatives and integrals, as follows:relate the terminal variables v and i, and2dvdt 2dvdt f vdt ff vdt2di R Ldt 2didi C R Ldti C R (i) L(i)f idt C(q) R(q) Lff idt C RIt can be shown, from energy methods, that many of these relationships cannot existphysically; for instance, an element relating d 2v/dt 2 to i is unrealizable, since whenit is connected in parallel with a linear resistor it becomes a source of infinite energy.In fact, it appears that only the four elements R(i), L(i), C(q), and R(q) are realizable.This suggests work in two areas.First, it should be possible to construct a proof that these four elements, and onlyQPR No. 92(VIII. SOLID-STATE MICROWAVE ELECTRONICS)these four, are realizable. Second, the element R(q), which has the relationshipdX vR(q) dq imay explain those phenomena that are not now handled by nonlinear network theory, suchas hysteresis. The justification for this statement is simply that if there are only 4realizable elements, and since three of them are not sufficient to describe hysteresis,then the fourth one must.R. G. WebbD. UNDERGRADUATE PROJECTSSeveral undergraduate projects are under way at the present time. One of these isthe design and construction of a FET bridge which will behave as a linear modulator at9 MHz. At the present time, the bridge works, and within reasonable limits the distor-tion is relatively low.Another project is associated with the log I vs V characteristics of Schottky-Barrierdiodes for use in two mixers, 50 0 and 200 S0. Included in this project is the constructionof an instrument to display these characteristics over a range of 6 or 7 decades.The other project is to construct a digital display system for the indication of windvelocity.D. H. SteinbrecherQPR No. 92

Solid-State Microwave Electronics

http://dspace.mit.edu/bitstream/1721.1/55831/1/RLE_QPR_092_VIII.pdf

Solid-State Microwave Electronics

Abstract

Similar works

Full text

Available Versions

DSpace@MIT