Contains research objectives and reports on two research projects

Adler, Richard B.

Gross, J.

Hurtig, C. R.

Mason, Samuel J.

Zimmermann, Henry J.

Contains reports on three research projects

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X. TRANSISTOR CIRCUITSProf. H. J. Zimmermann Prof. S. J. Mason J. GrossProf. R. B. Adler C. R. HurtigA. CONSTANT-RESISTANCE AGC ATTENUATORS FOR TRANSISTOR AMPLIFIERSFollowing the suggestion made in the Quarterly Progress Report of January 15, 1955,we have built a variable-gain bandpass amplifier, centered at 9. O0 Mc/sec and having anominal bandwidth of 1. 7 Mc/sec, with tetrode transistors. The amplifier in Fig. X-lconsists of four synchronously-tuned stages, two of which are gain-controlled by meansof constant-resistance agc attenuators. The pertinent data for three different designsare given in Table X-1.Table X- 1Characteristics of Tetrode AmplifiersDesignI II IIIMaximum power gain, db 29 17. 5 25Measured gain variation, db -42 -42 -42Center frequency (fc), Mc/sec 9.0 9. 0 9. 0Percentage of change in fc +1. 7 +0. 8 ±1. 4Nominal bandwidth (BW), Mc/sec 1.0 1.4 1.7Percentage of change in BW +15 +3.9 +3. O0Second base current (Ib ), ma 0. 3 0. 3 0. 3Maximum emitter bias current, ma 0. 3 0. 3 0.45Load resistor (RL), ohms o0 4700 4700The transformers are designed to provide an impedance match with a bandwidth ofapproximately 2. 3 Mc/sec per stage. Because of the dissymmetrical characteristicsof the external shunt diodes and the respective emitter diodes, and the diffusioncapacitance of the shunt diodes, the center frequency and bandwidth do vary somewhatwith changes in gain. The undesired variations may be reduced with a sacrifice in gain,as illustrated in case II. The tuned circuits, which are normally loaded by the variableattenuators, were shunted by additional 4700-ohm resistors. The bandwidth wasincreased, but the gain reduced by approximately 12 db. The variations in centerfrequency and bandwidth as functions of amplifier gain were correspondingly reduced. Incase III the maximum emitter current was increased by 50 per cent without redesigningthe transformers. For many critical applications the latter two amplifier designs mayprove satisfactory.-65-(X. TRANSISTOR CIRCUITS)CAPACITANCE IN fK =1000RL SEE TEXTALL TRANSISTORS RDX300Fig. X-lVariable-gain, constant-bandwidth tetrode amplifier.Several low-capacitance, point-contact, silicon diodes were tested in the attenuatorcircuits in an attempt to minimize the effect of diffusion capacitance variations with theagc voltage. The change in the center frequency with agc voltage was indeed reduced,but the bandwidth variation was increased by the increased dissymmetry of the diodes.It is probable that in all cases a significant portion of the variation in center fre-quency with agc voltage results from a corresponding variation in the transistor col-lector capacitance.B. THE OPERATION OF MULTIPLICATIONAs indicated in the Quarterly Progress Report of January 15, 1955, it is possible,by using a proper arrangement of the constant-resistance attenuator, to make the ampli-tude of a carrier signal a linear function of the control (agc) signal. By cascading twoof these stages, and applying the same signal at both control inputs, a square-law devicefor either a positive- or a negative-polarity control signal (but not both) may be built.This type of circuit is shown in Fig. X-2. The amplitude of the output voltage at thecarrier frequency of 1800 cps is shown as a function of a positive dc voltage at the inputfor two different pairs of transistors in Fig. X-3.When the input signal is equal to or greater than the fixed bias signal in the constant-resistance attenuator, the amplitude of the carrier signal remains constant. At verylow levels of input signal the carrier at the output becomes obscured in the transistornoise. The accuracy of the square-law action is better than ±10 per cent over at leasta 50-db dynamic range.-66-Fig. X-2Square-law device for a positive polarity input voltage.LAWIIDIRECT INPUT VOLTAGE (+ VOLTS)Fig. X-3Transfer characteristics of a transistor square-law device.-67-100I00(X. TRANSISTOR CIRCUITS)There are two basic limitations to this type of square-law device: the input isrestricted to a voltage of one polarity; the frequency response is severely limited. Ateither extreme of the dynamic range one of the diodes (either the shunt diode or theemitter diode of the transistor) is zero-biased, and the charge time constant is con-trolled by the large resistor used to simulate a current source. Thus, in the circuitshown in Fig. X-2 the charge time constant is approximately 0. 1 sec. The value ofthe capacitance may be reduced only if the carrier frequency is increased, sincethe emitter and shunt diodes must be kept at the same ac potential.Both of these limitations may be avoided by use of the circuit shown in Fig. X-4. Inthis circuit the quiescent operating point is set in the center of the linear region of theattenuator. The carrier signal at the output of the transistor at the quiescent point isbalanced out by the use of the phase-shifting transformer and the resistance-adding net-work. The amplitude of the output voltage is a linear function of the modulating voltage,with a phase shift of 1800 of the carrier signal occurring when the modulating voltagechanges polarity. This circuit acts as a balanced modulator whose output is a linearfunction of both the modulating input and carrier input. Preliminary tests indicate thata dynamic range of 60 db may be obtained with an accuracy of better than +10 per centover this range. The frequency response of the modulator may be as high as 10 percent of the carrier frequency. The stability of the balance of the modulator at thequiescent operating point has not been tested, but operation at room temperature hasbeen very satisfactory.To obtain the operation of multiplication, two such stages may be cascaded. Alter-natively, this type of stage may be preceded by a conventional balanced modulator. Toconvert the output from carrier waveform to direct current, a sensed amplitude detec-tor is required.C. THE PROCESS OF TIME INTEGRATIONIn many analog computers it is necessary to have a device capable of performing atime integration of a time-varying voltage. The term integration as commonly usedimplies a variety of desired objectives. Those of importance for our immediatepurpose are realized by a device that has a gain approaching infinity at zero frequencyand an ability to "remember" an integrated voltage after the input signal returns tozero. In applications where a memory accuracy of 0-10 per cent is desired for periodsapproaching one minute, the effective discharge time constant must be approximately10 minutes.The usual techniques for time integration require high-gain dc amplifiers. Unfor-tunately, our interest in transistor circuits for integration makes this method undesir-able because simple transistor dc amplifiers are severely limited by the temperaturedependence of the transistor parameters. Even though it is possible to minimize this-68-CARRIERINPUTesFig. X-4Balanced modulator useful for the process of multiplication.CURRENTSOURCE AMPLIFIER -- t LIMITER - MULTIPLIERINPUT VOLTAGEFig. X-5Block diagram of an integrator employing a pole on the j-o axis.Fig. X-6Transistor integrator employing a Q multiplier.-69-L___-_________________J(X. TRANSISTOR CIRCUITS)difficulty by using chopper stabilization or balanced-bridge techniques, the over-allamplifier performance is still limited by either time-varying parameters, excessivephysical size, power supply requirements, or component lifetime. In view of thesefacts, it appears that the technique of time integration at a carrier frequency may proveadvantageous for some applications.It can be shown fairly simply that a high-Q timed circuit acts as a lowpass filterto sideband modulation signals. As the Q approaches infinity, the memory time of thetuned circuit approaches infinity. That is, in the simplest case, a circuit having poleson the jw axis at values +00 will oscillate continuously at any prescribed amplitude. Bya proper arrangement, it is possible to control the total energy stored in the tunedcircuit as a function of an input signal. Since the Q of the tuned circuit is to approachinfinity, an external oscillator used to drive the tuned circuit would have to oscillate atexactly the circuit resonance, a condition difficult to satisfy. If, however, the signalinput is continuous, the energy stored in the tuned circuit may be used to drive theamplitude control circuit. One method of accomplishing this is shown in block diagramform in Fig. X-5. In this system the tuned circuit, composed of ideal components, isdriven by a current source. The voltage across the tuned circuit drives a limiter andmultiplier. The second input to the multiplier is the voltage to be integrated. The out-put from the multiplier is at carrier frequency (if the input is dc), but is either in phaseor 1800 out of phase with the carrier input. Thus, depending on the phase of themultiplier output, the total energy stored in the tuned circuit is either increased ordecreased.A preliminary transistor model is shown in Fig. X-6. The circuit consists of atransistor Q multiplier, an amplifier with a gain of 100, and a balanced modulator. Bycareful adjustment of the Q multiplier, the memory time may be made as long as 10minutes for signals in the range of 0-20 my rms across the tuned circuit. At the carrierfrequency of 250 cps the equivalent Q of the tuned circuit is 10 , representing aQ multiplication of approximately 10 . Although this multiplication factor can be main-tained satisfactorily at one temperature, it may change seriously with temperature.A less temperature sensitive and generally more stable system may be realized byusing a mechanically vibrating element to obtain a high initial Q. Bimorph quartzcrystals have Q's greater than 10, 000 at 1000 cps. However, the equivalent electriccircuit contains a zero and pole in close proximity. By using a bridge circuit it is pos-sible to obtain a transfer characteristic containing a simple pole, and then, by feed-back, to increase the Q.ERRATA: In the Quarterly Progress Report of January 15, 1955: Equation 3, page 72,should read: G 1 = Ie/K; line 2, page 74, should read: Thus the conductance G2 must ...Eq. 4, page 74, should read: G = G. - G = G.....2 n 1 InC. R. Hurtig-70-

Transistor Circuits

XII. TRANSISTOR CIRCUITSProf. H. J. Zimmermann Prof. S. J. Mason J. GrossProf. R. B. Adler C. R. HurtigRESEARCH OBJECTIVESThis research is aimed at problems of interest in communication systems. Prin-ciples of transistor circuit design and practical limitations imposed by available devicesare being investigated for circuits useful in a wide variety of applications. At thepresent time problems pertaining to gain control in transistor amplifiers are beinginvestigated.The study of noise in semiconductors is a closely related program.H. J. ZimmermannA. CONSTANT RESISTANCE AGC ATTENUATORS FOR TRANSISTOR AMPLIFIERSIt is well known that the variation of the transistor parameters with the dc biasconditions may be employed to control the gain of a transistor amplifier stage. Thereare, however, many applications that require a gain that may be varied by an externalvoltage, with the added restraint that the input and output impedances be maintainedapproximately constant. We have devised a unique circuit that satisfies these require-ments over a large range of gain control.The circuit employs a transistor in a grounded-base connection, having an ac loadresistance small compared to the collector resistance, and a dc bias supply resistorchosen as small as possible to insure operation at a constant collector voltage. Withthese restrictions, the output impedance of the amplifier stage is reasonably constant(or negligibly large compared to the load resistance) over a wide range of dc biascurrent. The incremental input resistance at 1000 cps of a typical p-n-p transistor,operated in the grounded-base connection with a low collector load impedance (less than1000 ohms), is shown in Fig. XII-1. Over a wide range of emitter current the inputresistance may be expressed asR. K kT (1)in Ie  qI ewhere k is Boltzmann's constant, T is the absolute temperature, and q is the elec-tronic charge. Thus if the input voltage to an amplifier stage is maintained constant,the output voltage will be a linear function of the dc emitter current.One possible method of maintaining the input resistance constant while varying theemitter current of the transistor is to place in parallel with the transistor input a resis-tance that is also controlled by a direct current (Fig. XII-2). The input conductance isGin = G 1 + G2  (2)butG2 = Ie/K (3)-72-UNIT NO.I10-4DC EMITTER CURRENT (AMP)Fig. XII-1Incremental input resistance of typical ck7l21's in the grounded-base connection.E GCI2 , VL I .Fig. XII-2Constant-resistance attenuator.Fig. XII-3Practical constant-resistance attenuator.e = I F -IFig. XII-4Alternate arrangement of a practical constant-resistance attenuator.-73-I10-6(XII. TRANSISTOR CIRCUITS)Gin = constantThus the conductance G1 must have the following characteristicI G.in K - IG2 = Gin - G2 = Gin - e e (4)2 in 2 in K KIt is seen that the required characteristic of the conductance G2 is identical with that ofthe transistor. That isI2G2 - K (5)where12 = Gin K - Ieor2 = IF - Ie  (6)Two practical arrangements of the circuit are shown in Figs. XII-3 and XII-4. Thecurrent sources are approximated by voltage sources in series with resistors large invalue compared to the desired input resistance. For many applications, resistors assmall as 20, 000 ohms are sufficient to satisfy this requirement. The experimental datashown in Fig. XII-5 apply to the circuit of Fig. XII-3. The output voltage is a linearfunction of the control bias within an accuracy of 10 per cent over a range of at least40 db, while the input impedance varies by less than ±10 per cent over the same range.The bandpass amplifier in Fig. XII-6 was constructed for a nominal center frequencyof 1. 7 Mc/sec. The data in Fig. XII-7 show that a bandwidth variation of approximately± 6 per cent accompanies a 54-db variation in gain. The same amplifier operated withoutthe diodes shows a bandwidth variation from 300 kc/sec to 70 kc/sec accompanying a48-db variation in gain. Thus the diodes improve the operation by a factor of approxi-mately 40. It should be noticed that the center frequency of the amplifier changed from1. 74 Mc/sec to 1. 64 Mc/sec along with the 54-db gain variation. Since this variationdoes not occur when the diodes are removed, the change in center frequency is attributedto the diffusion capacitance of the silicon diodes. By a substitution measurement tech-nique, the change in diode capacitance was found to be 250 4if from zero bias to themaximum bias of 100 pa. The effective hole lifetime of the silicon diodes was calculatedfrom this information to be approximately 0. 15 psec, which seems reasonable enoughto support the explanation proposed above.A bandpass amplifier, centered at 8. 0 Mc/sec, with a nominal bandwidth of 900kc/sec, has also been built and tested. A gain variation of 42 db produced a band-width variation of 11l per cent and a decrease of center frequency from 8. 4 Mc/sec to-74-0.1 1.0 10 100EAGC (VOLTSFig. XII-5Gain and input resistance of a practical constant-resistance attenuator.AGC/AI TCRa22KALL TRANSISTORSARE PHILCO SBTL 5106Fig. XII-6Bandpass amplifier incorporating two constant-resistance attenuator stages.-75-~ IVL ~I(XII. TRANSISTOR CIRCUITS)-3 -2AGC VOLTAGE (VOLTS)Fig. XII-7Characteristics of constant-resistance attenuator in an i-f amplifier.7. 83 Mc/sec. The change in bandwidth in this case is partially attributed to the fact thatthe input impedance of the transistor (even at an emitter current of 120 pa) is affected bythe base resistance and the collector load impedance. The change in center frequencyis caused by the diffusion capacitance of the silicon diode. Even with these limitations,the use of the constant-resistance attenuators reduces the bandwidth variation by a factorof 10, compared to the same amplifier with the diodes disconnected.Future work will include amplifier design employing tetrodes to reduce the feedbackacross the base resistance. A study of available diodes will be made, with the hope ofreducing the effects of diffusion capacitance.The possible variations of the age circuit are too numerous to illustrate. It is ofinterest to mention that the transistors used in the age stages may be connected to obtaindc gain for the age signal. If one of the transistors is connected with a large external-76-aIZ30oUjLOzImGLiiYr)'200zLiiDwa:LIiCrUt_TzLd0mzc)-JCr(XII. TRANSISTOR CIRCUITS)base resistance, by-passed for alternating current, it may be operated in the grounded-base connection for ac signals and in the grounded-emitter connection for the agc signal.The temperature stability of such a circuit is subject to all the limitations of a normalgrounded-emitter connection and is therefore less desirable than a grounded-base con-nection.B. THE OPERATIONS OF MULTIPLICATION AND DIVISIONThe data shown in Fig. XII-5 indicate that the constant-resistance attenuator may beused to vary the amplitude of a carrier signal as a linear function of a dc control signal.A diagram of a circuit that employs a constant-resistance attenuator to obtain the oper-ation of multiplication and/or division is shown in Fig. XII-8. From this figure, we havee = a i. RS in Lin = I (7)in s r d + r i  sIIe iR (8)Thus the amplitude of the carrier is a linear function of the current I 1 and inverselyproportional to the current I F . If two dc voltages are to be multiplied, two constant-resistance stages may be cascaded; or one stage may be driven from a balanced modu-lator, controlled by one of the dc signals.It is interesting to notice that by using a current source to inject the carrier signal(Fig. XII-8), the temperature-dependent term of the input resistance is eliminated fromthe expression for the output voltage. If the temperature range over which the devicemust operate is small, a voltage source may be employed.The accuracy of the multiplication circuit is difficult to evaluate. At the presenttime, preliminary experiments on devices employing resistors of less than 100 kilohmsas current sources indicate an over-all accuracy of better than ±10 per cent over a 40-db dynamic range. Higher accuracy resultsfrom a reduction of the dynamic range. If theamplitude of the carrier signal is to be variedby a dc voltage of variable polarity, the auxil-iary diode may be replaced by a transistorthat is the complement of the original. ByFig. XII-8 subtracting the outputs of the two transistors,Constant-resistance attenuator the desired function will be achieved.used to obtain the operations ofmultiplication and division. C. R. Hurtig-77-

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Transistor Circuits

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