Fan, Wei

Ji, Yilin

Pedersen, Gert Frølund

English

kyösti

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Wideband Radio Channel Emulation Using Band-stitching Schemes

In the fifth-generation (5G) new radio specifications, large signal bandwidth e.g. 400 MHz for frequency range 2 (FR2), is expected to be utilized. Therefore, there is a need to develop channel emulators that can support such large bandwidth. In this paper, we revisit the band-stitching scheme, i.e. combining multiple logic channels of small bandwidth to form a larger total bandwidth.We put a focus on the calibration stage of this scheme, and the effect of the phase jump at the junctions of adjacent subbands on the stitching performance is investigated through simulation

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    Aalborg UniversitetWideband Radio Channel Emulation Using Band-stitching SchemesJi, Yilin; Fan, Wei; Pedersen, Gert FrølundPublished in:2020 14th European Conference on Antennas and Propagation (EuCAP)DOI (link to publication from Publisher):10.23919/EuCAP48036.2020.9135788Publication date:2020Document VersionAccepted author manuscript, peer reviewed versionLink to publication from Aalborg UniversityCitation for published version (APA):Ji, Y., Fan, W., & Pedersen, G. F. (2020). Wideband Radio Channel Emulation Using Band-stitching Schemes.In 2020 14th European Conference on Antennas and Propagation (EuCAP) [9135788] IEEE.https://doi.org/10.23919/EuCAP48036.2020.9135788General rightsCopyright and moral rights for the publications made accessible in the public portal are retained by the authors and/or other copyright ownersand it is a condition of accessing publications that users recognise and abide by the legal requirements associated with these rights.            ? Users may download and print one copy of any publication from the public portal for the purpose of private study or research.            ? You may not further distribute the material or use it for any profit-making activity or commercial gain            ? You may freely distribute the URL identifying the publication in the public portal ?Take down policyIf you believe that this document breaches copyright please contact us at vbn@aub.aau.dk providing details, and we will remove access tothe work immediately and investigate your claim.Downloaded from vbn.aau.dk on: November 25, 2020Wideband Radio Channel Emulation UsingBand-stitching SchemesYilin Ji∗, Wei Fan∗, Gert F. Pedersen∗,∗(Affiliation): APMS section, Aalborg University, Aalborg, Denmark, e-mail: yilin@es.aau.dkAbstract—In the fifth-generation (5G) new radio specifications,large signal bandwidth e.g. 400 MHz for frequency range 2 (FR2),is expected to be utilized. Therefore, there is a need to developchannel emulators that can support such large bandwidth. Inthis paper, we revisit the band-stitching scheme, i.e. combiningmultiple logic channels of small bandwidth to form a larger totalbandwidth. We put a focus on the calibration stage of this scheme,and the effect of the phase jump at the junctions of adjacentsubbands on the stitching performance is investigated throughsimulation.Index Terms—Channel emulation, band stitching, testing,wideband.I. INTRODUCTIONChannel emulation techniques have been used extensivelyfor device testing in the industry [1]–[4]. Most of those tech-niques deploy channel emulators (CE) to generate target radiopropagation channels. For the fourth-generation (4G) and pre-vious generation communication technologies, the bandwidthis usually narrow, e.g. up to 20 MHz for singular-carrier LTEsignals. Some of the current commercially available channelemulators can support emulating channels with a bandwidthup to 160 MHz [5], which sufficiently cover the need forthose signals. However, with the development of the fifth-generation (5G), a much wider bandwidth, such as 400 MHzfor frequency range 2 (FR2) [6], is expected to be utilized.Therefore, methods to extend the current channel emulatorbandwidth are needed to meet the coming wideband signals.One of the methods, so called band stitching, is to combinemultiple logic channels in the channel emulator into one toform a wider bandwidth [7]. This method makes use of mostlythe existing resources in the channel emulator, so it is astraight-up approach compared to designing new hardware.In this paper, we go through the principle of band stitchingfor channel emulators, and put a focus on the calibration pro-cess of this method. Simulation analysis is done to investigatethe effect of calibration on the final performance of the formedwider logic channel.II. METHODSA. SchematicsThe schematics for stitching two logic channels for a single-input single-output (SISO) and a 2 × 2 MIMO system areshown in Fig. 1. The schematic for the SISO system is detailedin the following, and that for the MIMO system can be viewedas a simple step-up from the SISO case.In Fig. 1(a), a radio frequency (RF) signal xRF (t, f) is firstdown-converted by mixing with a local oscillator frequencyfLO. Here, t and f denote the time and frequency, respectively.The resulting intermediate frequency (IF) signal xIF (t, f) issplit to two identical signals via a splitter (Σ). The IF signalxIF (t, f) is further mixed with different subband IF frequen-cies fIFsubn in each branch. The resulting baseband signalxbn(t, f), with n = 1, ..., N being the index of subbands, issampled by the analog-to-digital converter (A/D). The targetchannel in baseband hbn(t, f) is realized via generating severaldelay taps (TDL) in the digital domain. The amplitude andphase of each tap is assigned according to the target channel.In addition, the filter shaping the frequency response of thechannel is also implemented digitally in this block, i.e. inthe finite impulse response (FIR) channel. This block can berealized via e.g. field programmable gate arrays (FPGA) [8].After multiplying the input signal xbn(t, f) with hbn(t, f),a up-converting process is followed similar to the down-converting phase, and the final faded signal yRF (t, f) isobtained. Note that there is a configurable attenuator and phaseshifter in each branch so that the power and phase of eachbranch can be tuned.This schematic can be easily extended for a 2 × 2 MIMOsystem as shown in Fig. 1(b). The target channel matrixH =[h11 h12h21 h22], (1)is emulated by stitching two logic channels for each entry toextend the bandwidth. The time and frequency dependency isomitted in (1) for brevity. A larger bandwidth can be realizedby combining more logic channels, and a higher order MIMOchannel can be emulated in the same principle similarly.B. Signal modelsFor brevity and without loss of generality, we focus on theSISO case in Fig. 1(a) hereafter. Given a radio propagationchannel consisting of L taps, the channel coefficients for thenth subband branch can be written ashbn(t, f) =L∑l=1αl(t) · e−j2πfIFsubnτl(t)· s(f − fIFsubn ),(2)where αl(t) and τl(t) are the complex amplitude and delay ofthe lth tap at time t, respectively. The term s(f − fIFsubn ) isthe filter response for the nth subband.FIR1Splitter CombinerCEAtten/PSA/D D/AFIR2A/D D/ATDL + Filter- : RF- : IF- : BB- : Digital(a)CEFIR1A/D D/AFIR2A/D D/AFIR7A/D D/AFIR8A/D D/AFIR3FIR4FIR5FIR6- : RF- : IF- : BB- : Digital(b)Fig. 1. Schematics for stitching two logic channels for (a) a single-input single-output (SISO) system and (b) a 2× 2 MIMO system. Note that the time andfrequency dependency of the signals is omitted for brevity.The raised cosine filter is considered in this work, whileother proper filters can also be used in general. The delay τdomain representation of the raised cosine filter s(f) can beexpressed ass(τ) = sinc(τ/T ) ·cos(πβτ/T )1− 4β2τ2/T 2, (3)where T is the reciprocal of the bandwidth of each logicchannel Bs, and in our case Bs is the same as the frequencyinterval between consecutive subband IF frequencies fIFsubn .The term β ∈ [0, 1] is the roll-off factor, and it is determinedbyβ =BstopBs− 1, (4)where Bstop is the stop bandwidth of the filter. The frequencyresponse of the filter s(f) can be obtained by Fourier transformover s(τ).The total response of the combined channel readshb(t, f) =N∑n=1hbn(t, f), (5)and the contribution of the lth tap taking (2) and (5) intoaccount, i.e.hlb(t, f) =N∑n=1αl(t) · e−j2πfIFsubnτl(t)· s(f − fIFsubn ), (6)are expected to comply with or at least approximate that of asingle tap channel, i.e. the magnitude of hlb(t, f) being thesame in the bands of interest and the slope of the phaseresponse of hlb(t, f) corresponding to the tap delay τl(t), aftersome tuning in β.C. CalibrationA vector network analyzer (VNA) or similar kinds canbe used to measure the frequency response of each branchsequentially while turning off the other branches.Magnitude responseMagnitude responseAfter tuning Atten/PSInitial measurementMagnitude responseAfter inserting with initial phase compensated(a)PhaseresponsePhaseresponseAfter tuning Atten/PSInitial measurementPhaseresponseAfter inserting with initial phase compensated(b)Fig. 2. Diagrams showing the effect of the calibration in different steps on the frequency response in (a) magnitude and (b) phase. An example of the 4-bandstitching with ideal filters is shown here for illustration. Different colors indicate different subbands.In the calibration phase, the FIR channel, if possible, canbe set to a “bypass” mode where only a single tap withzero time delay and unity amplitude is realized. When thisoption is not available, one can also set the FIR channelto a single tap with a small amount of delay, and subtractthat amount from the TDL model during testing. In fact,even when a zero delay is set in the FIR channel, it stilltakes some time for the signals to propagate in the branchbetween components. In this case, the propagation delay canbe estimated from the VNA measurement, and tuning theconfigurable attenuator and phase shifter can align the delayand gain in different branches. In addition, the initial phaseof each branch at fIFsubn can be read from the measurement.Finally, the baseband channel coefficients hbn(t, f) for eachsubband shall be calculated according to (2), and inserted tothe FIR channel as the complex amplitude of that tap with theinitial phase compensated.After the tuning and alignment across branches is done, aflat frequency response in magnitude and continuous phase re-sponse after unwrapping with its slope matching the estimateddelay is expected to see in the VNA measurement in the totalband of interest. With that, a wider bandwidth, i.e. (N · Bs),can be realized via stitching N logic channels. The diagramsshowing the effect of the calibration on the frequency responsein magnitude and phase after different operations are shownin Fig. 2.Nevertheless, as far as the authors are concerned, commer-cial channel emulators are built in high quality. Therefore, theinitial misalignment in amplitude and phase as shown in theleft most subplots in Fig. 2 might be small and negligible. Inthis case, a common way of calibrating different branches is totune the reconfigurable phase shifters exhaustively so that theunwrapped phase is approximately continuous at the transitionbetween adjacent subbands [7]. This method is intuitive buttime consuming. Compared to this time consuming method,the method detailed previously is more efficient and explana-tory from the signal model stand point.III. SIMULATION RESULTSIn the simulation, the raised cosine filter with parameterBs = 135 MHz and Bstop = 170 MHz is implementedaccording to (3) and (4). The resulting filter response s(τ)and s(f) in the delay and frequency domain are shown inFig. 3, respectively.Our objective in this work is to form a combined bandwidthof 500 MHz, so 4 logic channels, each of which has 135 MHzbandwidth, are stitched together.As shown in Fig. 2(b), the slopes of the phase responses ofdifferent branches need to be the same, which can be realizedby tuning the delay of each branch. Moreover, the phaseresponses between consecutive branches need to be connectedend to end. This is realized by compensating the initial phaseof each branch in hbn as mentioned in Section II-C. If theinitial phase of each branch is not compensated, phase jumpwould occur at the junctions between adjacent subbands. How-ever, the phase jump ranges in (−180◦, 180◦). Compared tothe large scale of the unwrapped phase response over the totalband of interest, this jump is probably unnoticeable, and henceoverlooked sometimes. Note that the stitching performancemight potentially deteriorate due to this. To investigate theeffect of the phase jump on the stitching performance, wedesigned two cases in the simulation, i.e. one with the initialphase compensation and one without the compensation.-10 -8 -6 -4 -2 0 2 4 6 8 10delay [1/Bs]00.51Amplitude [linear]-400 -300 -200 -100 0 100 200 300 400f [MHz]-150-100-50050Magnitude [dB]Fig. 3. The filter response in the (top) delay and (bottom) frequency domain.Bs = 135 MHz and Bstop = 170 MHz.-250 -200 -150 -100 -50 0 50 100 150 200 250Frequency [MHz]-4-3-2-1Phase [deg]104w/ compensationw/o compensation-250 -200 -150 -100 -50 0 50 100 150 200 250Frequency [MHz]-30-28-26-24Phase diffrence [deg]w/ compensationw/o compensationFig. 4. The phase response of the 1-tap TDL channel with the combined 500MHz bandwidth. (Top) the unwrapped phase against frequency, and (bottom)its first-order difference.The initial phase of each logic channel is generated ran-domly. A 1-tap TDL channel is generated for the calibrationprocess. The phase responses for the two cases are shown inFig. 4. As expected, the phase jump is more noticeable inthe first-order difference of the phase response for the casewithout the initial phase compensation.Fig. 5 shows the magnitude of the frequency response ofthe combined 500 MHz channel for both cases. We can seedeep drops in magnitude at the junctions between adjacentsubbands in the case without the initial phase compensation,which justifies the necessity of the compensation despite theinsignificance of the phase jump in the big picture over thetotal band as in Fig. 4(top).Fig. 6 shows the power delay profile of 1-tap TDL channelcalculated from the frequency response by the inverse Fouriertransform for both cases. The peak of the combined channel isout of shape in the case without the initial phase compensationcompared to that with the compensation. Nonetheless, a higher-400 -300 -200 -100 0 100 200 300 400Frequency [MHz]-150-100-500Magnitude [dB]w/ compensationsubband1subband2subband3subband4combined-400 -300 -200 -100 0 100 200 300 400Frequency [MHz]-150-100-500Magnitude [dB]w/o compensationsubband1subband2subband3subband4combinedFig. 5. The frequency response of the 1-tap TDL channel with the combined500 MHz bandwidth. (Top) the case with the initial phase compensation, and(bottom) the case without the compensation.25 30 35 40 45Delay [tap index]00.51Magnitude [linear]w/o compensationsubbandcombinedTDL25 30 35 40 45Delay [tap index]00.51Magnitude [linear]w/ compensationsubbandcombinedTDLFig. 6. The power delay profile of the 1-tap TDL channel with the combined500 MHz bandwidth calculated by the Fourier transform. (Top) the casewith the initial phase compensation, and (bottom) the case without thecompensation.delay resolution is achieved in the combined channel comparedto that in the single subband channel as shown in Fig. 6(top)due to the larger combined bandwidth.Finally, we show the frequency response and the powerdelay profile of a 4-tap TDL channel with the combinedbandwidth in Fig. 7. The 4-tap TDL channel is well generatedwith the combined 500 MHz bandwidth, and higher delayresolution is achieved compared to the single subband case.IV. CONCLUSIONThe principle of band-stitching for channel emulation isrevisited. We put a focus on the calibration phase of theprocess. It is found a wider bandwidth can be realized viacombining multiple logic channels into one. The effect ofphase jump at the junctions between adjacent subbands onthe stitching performance is investigated via simulation. Theresults show though the phase jump is small compared to the20 40 60 80 100 120 140Delay [tap index]-40-200Magnitude [dB]Power delay profilesubbandcombinedTDL-400 -300 -200 -100 0 100 200 300 400Frequency [MHz]-150-100-500Magnitude [dB]Frequency spectrumsubband1subband2subband3subband4combinedFig. 7. (Top) the frequency response and (bottom) the power delay profile ofthe 4-tap TDL channel with the combined 500 MHz bandwidth.total phase change over the band of interest, it can lead to deepdrops in the frequency response and out-of-shape peaks in thepower delay profile for the combined channel. Therefore, it isimportant to compensate that in the calibration phase.ACKNOWLEDGMENTREFERENCES[1] P. Kyösti, T. Jämsä, and J.-P. Nuutinen, “Channel modelling for multi-probe over-the-air MIMO testing,” International Journal of Antennas andPropagation, vol. 2012, 2012.[2] W. Fan, X. Carreño, J. Ø. Nielsen, K. Olesen, M. B. Knudsen, and G. F.Pedersen, “Measurement verification of plane wave synthesis techniquebased on multi-probe MIMO-OTA setup,” in IEEE Vehicular TechnologyConference (VTC Fall), 2012, pp. 1–5.[3] W. Yu, Y. Qi, S. Member, K. Liu, Y. Xu, and A. Two-stage, “RadiatedTwo-Stage Method for LTE MIMO User,” IEEE Transactions on Elec-tromagnetic Compatibility, vol. 56, no. 6, pp. 1691–1696, 2014.[4] 3GPP, “Verification of radiated multi-antenna reception performance ofUser Equipment (UE),” Technical Specification Group Radio AccessNetwork, Tech. Rep. 3GPP TR 37.977 V14.3.0, 2017.[5] Keysight, “Keysight Technologies Propsim F8Channel Emulator,” Tech. Rep. [Online]. Avail-able: https://literature.cdn.keysight.com/litweb/pdf/5992-1609EN.pdf?id=2754121[6] 3GPP, “5G; NR; Base Station (BS) radio transmission and reception,”Tech. Rep. 3GPP TS 38.104 version 15.2.0 Release 15, 2019. [Online].Available: https://portal.etsi.org/TB/ETSIDeliverableStatus.aspx[7] W. Fan, P. Kyosti, L. Hentila, and G. F. Pedersen, “A flexible millimeter-wave radio channel emulator design with experimental validations,” IEEETransactions on Antennas and Propagation, vol. 66, no. 11, pp. 6446–6451, 2018.[8] K. C. Borries, G. Judd, D. D. Stancil, and P. Steenkiste, “FPGA-basedchannel simulator for a wireless network emulator,” IEEE VehicularTechnology Conference, no. April 2009, 2009.

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Wideband Radio Channel Emulation Using Band-stitching Schemes

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